EXCLUSIVE INTERVIEW WITH CHAIRMAN - ELECTRONICS COMMISSION ELECTRONICS 5 | ;ndia OVER 100 EXCITING PROJECTS gin Pg r ^ ■3 P^Y ■ Radio & TV Power supplies A Supplement - Electronics India '88 Exhibition Volume-6 Number-9 ANNUAL NUMBER September-1988 CONTENTS Auufeafy rotative-voltage Source Dover lor bipolar stepce- moto-5 . Electron*: sand-g ass Production: C.N. Mithagari Address : ELEKTOR ELECTRONICS PVT. LTD. 52, C Proctor Road, Bombay-400 007 INDIA Telex: (011)76661 ELEK IN Fruit machine High-voltage BC547 Programmable switching sequence Quu timer Servo-pulse gene-ato- Stepper motor dnver Touch-sensitive . ght switch Universal SMO-to-D l adaptors Overseas editions: Up'Down control tor digital potentiometer Tram detecto- Maijai slide tador Computer -or-senso- -co Soft start lor halogen it Electronic mousetrap Two-wire remote conn 9-Chan nei touch -sensi Fox hunt Light-to-lrequency con Giant LED display Ight delay Deceptive car alar Car - IOC* delros-cr Computer-driven power controller Discrete + SV to -1 5V converter Flashing tignt Lead-aoc battery eha-get Over-voltage protection Secondary power-on relay Sett switcnirg power SJppy '■/t/BIl**' Strength and weaknesses ol Indian Electronics - An Exclusive Interview - Chairman - Electronics Commission NCST: MECCA ol soltware Computers in Air Deience °r PROTOTYPING BOARD FOR COMPUTER EXTENSIONS This printed circuit board is ideal for building and testing experimental ex- tension circuits for a wide range of com- puters. The double-sided, but not through-plated, board has contact fingers that enable it to be accepted in commonly used slot connectors for ex- tension circuits in many types of com- puter, including those in the MSX and IBM PC series. In addition, the board holds 3 general-purpose buffer chips which can be wired to requirement to ensure correct and safe interfacing be- tween the computer and the extension circuit being developed. Supply tracks are provided in the buffer and prototyp- ing area on the board for ease of wiring. When required, a number of contact fingers can be cut off to suit a particular slot width, or to prevent the board being fitted the wrong way around in the slot. Also, the contact fingers are relatively long so that a section of this PCB area can be cut off for use as an adaptor together with a purchased slot connec- tor. It is also possible to fit a slot connec- tor at right angles at either side of the PCB as shown by the printed markers. The' pin connections of the Type 74HCT245 octal transceiver, and the Type 74HCT541 octal three-state line buffer are given here for reference. These chips are suggested for use as databus and addressbus buffers re- spectively, because they have inputs and outputs arranged at opposite sides of the 20-way DIL enclosure. The user is, however, left completely free to choose his own bus buffers in accordance with the interfacing requirements. Remember to ground unused inputs on HCT chips! 9.66 elelctorindia September 1988 j T HREE-WAY TONE CONTROL Although tone control is not desirable in good-quality audio equipment, there are still instances, such as when playing well-used records, when it is. Such an add-on tone control should enable the frequency response to be altered to taste, have no detrimental effect on the audio equipment, and be fairly com- pact. The circuit proposed here meets these criteria. It is based on National Semiconduc- tors's LM833. This dual operational amplifier has a very low noise factor (4.S nV/|/f (Hz)), a high gain-bandwidth product (15 MHz), and a slew rate of 7V//s. The tone control circuit consists of three ranges, so that a presence control at around 1 kHz is possible. The opamp at the input, Ai, is connec- ted as an inverting buffer. Its non- inverting input is connected to a 10 k re- sistor to equalize the direct currents at both inputs (with respect to the bias cur- rents). This is necessary to keep the out- put of A i near enough at 0 V because of the d.c. coupling to A2. The second opamp has in its feedback loop a simple three-way tone control, whose cross-over points are determined by the value of the four capacitors. If desired, a capacitor may be added to the output of Aa, because the d.c. out- put of this opamp varies somewhat with the setting of the potentiometers. The cross-over points of the low- frequency and high-frequency controls lie at about 200 Hz and 2 kHz respect- ively. The presence control operates at around 1 kHz. Maximum attenuation is about 16 dB. With all potentiometers at the centre of their travel, the signal-to-noise ratio is better than 90 dB at a bandwidth of 1 MHz. The gain is 0 dB but can be altered by changing the value of R2. DMM AS FREQUENCY METER By providing a high-input-resistance multimeter (preferably of the digital type) with a frequency-to-voltage con- verter, it can be used to measure fre- quency. The range of the proposed device ex- tends from 10 Hz to 1 kHz on range A and from 1 kHz to 100 kHz on range B. The sensitivity for frequency measure- ments up to about 10 kHz is of the order of 35 mVpp, and for measurements from 10 kHz to 100 kHz about 350 mV PP . The input signal is applied to Schmitt trigger ICs via limiters Di and Dz Bistables FFi and FF2, and IC2 form a monostable. When the monostable is triggered, it generates a pulse whose width is accurately determined by a 12- MHz crystal. The number of times the monostable is triggered per unit time depends on the input signal. The pulse height depends on the supply of the monostable. The supply is provided by voltage regulator IC* and is about 5 V. At the output of the monoflop, ie., pin 13 of FF2, there will thus be a train of pulses, whose width and height are con- stant, but whose number and, therefore, the average voltage is directly pro- portional to the input frequency. The RC network at the output of FF2 forms a low -pass filter, so that the average voltage of the pulses will ap- pear across Ce. Potentiometers Pi and P 2 and resistors R7 and Ra form a potential divider which enables the frequency-to-voltage conversion factor to be adjusted. The voltage across C6 measured by the DMM is thus directly proportional to the frequency of the input signal. In range A, a voltage of 10 mV cor- responds to 10 Hz, and 1 V to 1 kHz. In range B, 10 mV corresponds to 1 kHz, and 1 V to 100 kHz. For adjusting the meter, temporarily connect the junction of R7 and Ra to pin 12 instead of to pin 13 of FFz There should be no input signal. Set the DMM to the 20 V range, and connect it across C& Set S2J0 position A, and adjust Pi un- til the meter reads 2.93 V. Then set the meter to the 2 V range, and S2 to position B. Now adjust P2 until, the meter reads 1.875 V. Finally, reconnect the junction of R7 and Rs to pin 13 of FFz The meter may be powered by a 9-V PP3 battery: the current consumption amounts to only 10 mA. °| R T RTTY FILTER FOR 170 HZ SHIFT Anyone interested in the reception of radio teletype traffic will appreciate the programmable audio filter described here, which may be fitted at the input of the RTTY converter. It improves the signal/-to- noise ratio, particularly of signals in the crowded short-wave bands. The circuit is based on programmable filter ICa-see Fig. 2. The special facet of this IC is that the resistors of the on- board RC filters are simulated by capacitors. This little-known technique is described in the January 1981, October 1982, and February 1983 issues of this magazine. The value of the capacitors, and therefore the pass-band frequency of the filter, is determined by the fre- quency of the clock at pin 8 of ICi The clock frequency is made variable by passing a 10-MHz signal through a pro- grammable divider, ICi. The divisor may be set between 1 and 256 with the aid of switches Sia-Sih. Monostable IC 2 converts the output pulses from the divider into a near- symmetrical signal, which is sub- sequently used as the clock for ICa (pin 8). fi(kHz) 884004-11 The filter characteristic is shaped by the resistors at the various pins of IC 3 to a very-narrow pass-band as is required for small RTTY shifts. The characteristic is shown in Fig. 1. The entire pass-band may be shifted with the aid of the switches. In narrow-band RTTY (70 - 170 Hz shift), one filter suffices, since both the high and low AFSK frequencies can be passed by the filter. For broadband RTTY signals (425-850 Hz shift), it is probably better to use separate filters for the high and low frequencies. The circuit draws a current not ex- ceeding 20 mA. 2 884004-10 °1 SECONDARY POWER-ON DELAY The circuit described here enables short-circuit protection and power-on delay to be added to a power supply. Power supplies with a large reservoir capacitor may draw such large currents on switch-on that problems occur, even at the primary of the mains transformer. Particularly when a toroidal mains trans- former is used, it may be necessary to fit a much heavier primary fuse than is desirable for normal protection. The current in the secondary is limited by a resistor, Ri, in series with the reser- voir capacitor, Ci. A few seconds after switch-on, Ri is short-circuited by a relay contact. Compared with switching at the primary side, this method has the advantage that no separate supply for the relay is necessary and that this does not have to switch the 240 V mains. Operation is fairly simple. After switch- mally the voltage regulator does not on. on, Ci is charged slowly via Ri. After a have to limit (less dissipation). The earth of the circuit is in a somewhat few seconds, the output voltage has Switch Si enables a choice to be made unusual place to enable ICi to be risen sufficiently for the relay to be en- between a fixed output of 12 V and one mounted on to the heat sink without an ergized, which causes Ri to be shorted. variable between 12 and IS V. insulating washer (IC ground is connec- When the output of the supply is short- With heavy loads it may occur that the ted to its case). For this reason, it is not circuited, the output voltage drops to a output voltage remains too low, because permissible to use the earth for external level where Rei is de-energized. of Ri, to energize the relay. In that case ground connection. Because Ri is then in circuit again, the it will be necessary to remove the load short-circuit current is limited and nor- from the supply before this can switch Many cars are fitted with some sort of The circuit described here is an add-on these switches to close, some additional alarm system as protection against petty to an existing alarm and energizes this circuitry is necessary, criminals and joy riders. Most of these when the position of the car is changed, The four D-type bistables in ICi deter- systems rely on a door switch and one for instance, by a jack being placed mine the output state of the mercury under the bonnet (to prevent inter- under it. switches. The outputs of ICi are con- ference with the battery connections to The position of the car is monitored by nected to gates Ni to Ni which function immobilize the alarm system). Such four mercury switches which are as inverters when the mercury switches systems afford no protection whatsoever placed in such a way that when the car initially are closed (so that there is a 1 at to another criminal pest: those who jack is horizontal they are open. Because a the output of the relevant bistable). This up the car and remove expensive alu- car is sometimes parked in an inclined results in the outputs of the four gates minium sports wheels. position, which causes one or more of remaining 0 as long as the mercury 9.70 da switches stay in that initial state. switches must be kept long enough to If only one of the mercury switches allow the switch to be slightly tilted with changes state, the output of Ns goes respect to the board. The side of the high and Ti switches on. This transistor switch in contact with the board may may, for instance, be connected in paral- then be fixed into position with araldite lei with the door switch. or a similar fixative. This arrangement The output state of the bistables may be ensures that all switches are open when stored via R1-C1 at the moment the the car is horizontal, supply is switched on. All car alarms have a certain delay after being switched on to give the occupants time to get out of the car. If a signal is available that becomes 1 after this delay, it may also be used to store the output states in the bistables. Resistor Ri and capacitor Ci must then be discon- nected. This second method has the ad- vantage that if a mercury switch is just about changing state, the closing of the car doors will render it stable. The mercury switches are mounted on the PCB together with the other compo- nents. One of the terminal wires of the Described is a digital panel meter— DPM— which is built around a special meter-IC, Type ADD3701, and may be used for the accurate measuring of voltage from a variety of sources. A highly stable reference voltage is pro- vided by an LM336. A ULN2003, IC< is used to buffer the outputs of the ADD3701, so that the common-cathode displays can be driven direct. The ADD3701 multiplexes the displays so that the number of control lines is kept down. The current through the display segments is limited by resistors Re to Ris incl. The oscillator that determines the con- version rate of the analogue-to-digital converter in ICi requires an extern RC network (R7-C6). Because of the need of adequate suppression of the mains fre- quency, the oscillator frequency must be exactly 400 Hz (it is very nearly equal to O.6R7C6). A preset potentiometer may be connected in series with R7 to adjust the frequency accurately. At this oscil- lator frequency, there are about 3 con- versions per second. Another possibility of avoiding inter- ference from the mains frequency is to use the DPM for measuring positive voltages only: LDs is then not required. The input voltage is applied to Vrur (pin 11) via a 100 kQ resistor. Input ter- minals V(+) and V(-) are not used in this case. Also, the oscillator frequency need not be exactly 400 Hz. The DPM is calibrated by short- circuiting the input and setting P2 to a position where the display reads 0.000. Then apply a voltage of 1.900 V to the in- put and adjust Pa till the display reads 3.800. An input voltage of 1.999 V will then result in a display reading of 3.999. Take this into account if an input at- tenuator is contemplated. The load presented by the input stage to a potential divider at the input is very small: typically, the input current is 1 nA (maximum 5 nA). The (unregulated) supply should be able to provide 8 to 12 V at a current of 250 mA. The circuit, including the dis- plays, draws about 150 mA. (National Semiconductor Application) I Repairs to the e.h.t. section of a monitor j or a television receiver always carry a certain amount of risk. It makes sense, therefore, particularly for the less ex- perienced technician, to seek a safe way of checking the extra high tension. ■ In all television receivers and monitors, the e.h.t. is generated in the deflection circuits. These circuits operate at about 1 16 kHz which generates a fairly strong magnetic field via the line transformer. ! It may be safely assumed that as long as the deflection circuits function cor- ! rectly, the e.h.t. will also be all right. Ad- | mittedly, there is a possibility that a j defect high-tension winding may be the culprit. But let’s not be pessimistic The proposed circuit enables ‘wireless’ monitoring of the e.h.t. section, since it picks up all signals between about 14 kHz and 45 kHz (and their harmonics) and converts them into audio signals. I The frequency of oscillator IC> may be | varied with the aid of a potentiometer. | The oscillator output is mixed with the 9.72 els detected deflection signal in Ti. Since IC2 is connected as a gyrator, filter Li -Ci at the drain of Ti removes an audio signal from the mixing product. The small audio signal is amplified in T2 to a level sufficient to drive a small loud- speaker. The detector ‘probe’ is best made from a short length of insulated equipment wire, preferably, but not necessarily, connected to a small insulated metal plate. To test whether the deflection cir- cuits operate correctly, the monitor or television receiver, as well as the test circuit, must be switched on. Then the probe should be placed in the vicinity of the line transformer and the poten- tiometer in the tester adjusted until a constant whistle is audible from the loudspeaker. When the monitor (TV re- ceiver) is switched off, this whistle should disappear. If this happens, the deflection, and therefore almost cer- tainly the e.h.t., will be all right. SIMPLE TRANSISTOR TESTER While experimenting with electronic circuits, it will often be necessary to rap- idly test bipolar transistors and FETs before they are fitted in the circuit, or when they have been removed from the circuit when a malfunction is suspected. More specifically, constructors will need to know whether a transistor of known type and make is sound or not, and also whether an unknown device is a particular type of FET, or a bipolar transistor (PNP or NPN). This tester can be built from parts found in the junk-box. When the transistor under test (TUT) is OK. and correctly connected, the circuit will oscillate dur- ing half the period time of the alter- nating supply voltage (50 or 60 Hz). Red LED D2 lights when the TUT is OK and of the NPN type. The function of green LED Di is similar for PNP TUTs. The TUT OK/not OK indication is obtained with S2 set to the centre position, and Si opened as shown in the circuit diagram. The LEDs will indicate that the oscilator amplitude is significantly reduced, or nought, when Si is closed with a bipolar TUT mounted. Correctly operating FETs produce oscillation ir- respective of the position of Si. Only J- FETs and dual-gate MOSFETb produce oscillation when S2 is set to positions A and C. The accompanying table should speak for itself. Note that S3 must be opened and closed after each change in the position of S2. Finally, the tester is preferably fed from a 6 VAC mains adapter. 1 i9.73 HIGH-VOLTAGE BC547 It is sometimes desired to use a BC547 at rather higher voltages than permitted according to the data book. Yet, it can be done by connecting a number of them in series as shown in the ac- companying diagram. The set-up has a few, small, disadvan- tages: there is a constant leakage cur- rent through the series resistors and the saturation voltage is rather higher. Where these disadvantages are of little or no consequence, the circuit shown here can be used with voltages up to about 100 V. Assume that a voltage of 100 V has to be switched and that the maximum current is 2 mfl. If the current amplification is 200, the base current will be 10 /jR. Transistor Tj will then switch on as soon as the p.d. across R« is 0.68 V. The base current of T? also flows through R4, so that the drop across this resistor rises to 1.36 V. The current that switches Ti flows through Ri, so that it does not cause an additional p.d. across the potential div- ider. There is, of course, the usual saturation voltage of about 0.2 V across Ti. The total drop across the divider is then 3(10' 5 x 68 x 10 3 ) + 0.2 = 2.2 V. Increasing the resistor values to 270 k raises the saturation voltage to 8.3 V. The leakage current is then much smaller. n SERVO-PULSE GENERATOR Circuits for the generation of control pulses for servo apparatus remain popular, which seems a good enough reason to present another one. The popularity of servo control is en- hanced by the low price of servo motors, and the fact that they can be used for a variety of applications. The present design is geared to stand-alone use of the servo. Simplicity of the circuit was the first design consideration, and it seemed reasonable, therefore, to base it on the well-known SS5 IC. Unfortunately, this chip has the property, in its standard configuration, of producting pulse trains with a duty factor* of 50% or greater. This is so, because the charging time constant is always greater than the discharge one, since during charging the discharge resistance is in series with the charge resistance. Servos, on the other hand, require pulse trains with duty factors well below 50%. Ideally, the pulses should have a width of 1-2 ms, and the pulse repetition frequency -prf- should be about 50 Hz. This gives a duty factor of 5 - 10%. This problem may be resolved by invert- ing the output signal of the 555 with the aid of a transistor and two resistors, but this was considered extravagant. All it needs is an extra diode and relocating the discharge resistance. The charging time, and therefore the length of time that the output is logic high, is now de- termined by Pi, Ri, and the discharge time through R2. The component values in the circuit have been chosen in a manner that causes the pulse width to change from 1 ms to 2 ms when the resistance be- tween the positive line and the anode of Di is increased from 2k7 to 5k4. This re- duction in resistance is brought about by a 75° shift of Pi (normal joystick travel), if this potentiometer has a value of 10k. This potentiometer must be set to a position where its resistance is 4kl when the joystick is at centre position. Resistor Ri should then be replaced by a wire link. It is possible to use the normal 270° travel of the potentiometer, which should then have a value of 2k7. Resistor Ri must then be used as shown. 0 1 SELF-SWITCHING POWER SUPPLY The proposed power supply switches itself off when no current is drawn by the load. How this is done is shown in the circuit diagram, Fig. 1. When a load current flows, the p.d. across Di is sufficient to cause D2 and T2 to conduct. Ti is then switched on and the relay is energized. When the load current ceases, T2 switches off. The base current of Ti will then charge C2 so that after a few seconds the relay is de- energized. The relay contact, rei, will then switch off the mains at the primary of the transformer. The supply is switched on again by reconnecting the load and pressing Si briefly. The output voltage depends on the re- sistance between A and B. A wire link there results in an output voltage of about 3.5 V. For each 100R increase, the output voltage will rise by about 1 V (the current from the regulator to ground is a nearly constant 10 mA). This makes it possible to obtain a variable output voltage with the aid of some resistors and a rotary switch as shown in Fig. 2. The relay, Rei, should be of a type that is suitable for switching mains voltages. The a.c. rating of the secondary of Tri must be about 1.5 times as high as the desired do. output current. The output current should not exceed 1A; if that magnitude of current is drawn regularly, it is recommended to increase Ci to 1500 11F. The delay in switch-off may be ex- tended by increasing the value of Cz The heat sink of ICi should be in ac- cordance with the output current. 1 9.75 m LCD FOR Z80-DRIVEN COMPUTERS There is a growing tendency to use liquid-crystal displays (LCD) as the screen of computer monitors. Such dis- plays may also be used wherS the nor- mal monitor is too large or draws too much current; they are readily available. An LCD is normally driven by a micro- processor: in the proposed circuit by a Z80. The display in the proposed circuit is a Sharp Type LM16251: a full description of this appeared in the May 1986 issue of this magazine. It is located in the I/O region, addresses 0 to 3, of the pro- cessor. This arrangement enables the circuit also to be used in combination with the 32-bit I/O and timer cartridge described in the January 1987 issue of this magazine. This cartridge does not use the lowest four addresses (choose address 0 for the cartridge so that an ad- ditional I/O region of 0 to 15 is ob- tained). The address coding is effected by gates Ni to N4. When Az to Ai are, and IO REQ becomes, low, the output of N3 goes low. If Ml is high (no interrupt demanded), N4 outputs a 1 and an en- able signal is given to the display. Depending on the logic levels at inputs R/W and RS, data is transmitted or re- ceived. The RD and WR outputs of the Z80 are not used, because the R/W and RS signals of the LM16251 must be stable not later than 140 ns before the E input goes high. If the RD or WR signals of the processor were used, the E input of the display would be accessed together with the other signals and that is not per- mitted. By using an address line, the timing is arranged by that of the Z80, because the address bus must be stable not later than 320 ns (180 ns for Z80A) before an IO REQ signal is generated. Owners of a Z80B-driven computer might have some problems here because the time delay is then only 110 ns. Note that MSX computers invariably use a Z80A. The negative voltage for the contrast setting (Pi) of the display is provided by Ne. Note that some types of display need a positive voltage for the contrast setting. Wire link ‘a' provides a negative supply, and ‘b' a positive one. Link ‘a’ is required for the LM16251. If another type of display is used, make sure that the pin numbering is the same as shown in the diagram. Gate Ns serves to render the BUSDIR line low at an I/O read command in MSX systems. In other systems, this gate is not required. CAR INTERIOR LIGHT DELAY It's dark and it’s raining cats and dogs. You rush to your car, open the door and quickly close it behind you again. Then you sit there fumbling for the ignition lock. Solution? Add the following cir- cuit, which will keep your car’s interior light on for a little while after the door is closed. The circuit is connected across the switch in the door post. These switches are removed quite easily. In the circuit diagram, Si is the switch in the car's doorpost and Li is the interior light. As long as the door is open, Si is closed and the light is on. When the door is closed, Si opens and the light goes out. The full 12 V from the car bat- tery is then present across the switch. The circuit detects when the voltage across Si begins to rise. Transistor T3, and consequently Ti and T2, is then switched on. This results in the voltage across Si rising to about 1 V, after which it can increase only very slowly. This means that the interior light stays on, although its brightness will slowly decrease. At a certain level of potential across Si, transistor T4 switches on, which results in the drive to T3 becoming zero, and T3, 9.76. 1-2, and Ti switch off. The interior light will then go out very quickly. The delay in the light going out after the car door is closed is preset by Pi, although it is also affected by the value of Ci. The larger this value, the longer the delay and the smaller the variation in the brightness of Li. After the light has gone out, the circuit no longer draws current. POLAROTOR CONTROL The polarization of satellite TV signals is defined as horizontal (H) or vertical (V) with respect to the equator below the subsatellite point, and not, as is often wrongly assumed, with respect to the horizon on earth. Depending on the lo- cation of the receiving system on earth and the satellite’s geosynchronous pos- ition, a horizontally polarized signal may have some offset with respect to the horizon. As a rule of thumb, the lower the dish elevation for a particular satel- lite, the greater this polarization offset angle. The difference between horizon- tal and vertical is, however, always 90°. Most commercially available polariza- tion rotation units (polarotors ) used for selecting between horizontally and ver- tically polarized transponders on board a TV satellite incorporate a small servo motor whose direction of travel is con- trolled automatically by the channel selection circuitry in the indoor unit, or simply by a switch. The servo motor rotates an angled probe fitted in a PTFE bush in the waveguide flange that is secured onto the feed horn. This probe can be rotated over 90°, and re-transmits the received 11 GHz satellite signal by means of a VtX probe fitted vertically in the waveguide that connects to the LNB. The polarotor assembly is fitted perma- nently between the feed horn and the LNB input, and is connected to the in- door unit via a length of 3-wire cable, which runs in parallel with the downlead coax. A polarization selection switch, S3, is provided on the Indoor Unit for Satellite TV Reception (1) , but not the accompanying driver circuit, which is given here. The polarotor control is an astable multivibrator that determines the direc- tion of travel of the servo motor by sup- plying output pulses with a duration of 1 ms (V) or 2 ms (H) (typical values). When horizontal/vertical (H/V) switch S3 is closed, Pi is short-circuited, so that ICi supplies pulses with a duration of 1 ms. In the polarotor assembly, a com- bination of a potentiometer coupled to the motor spindle and an electronic cir- cuit is used for comparing the duration of the received control pulses with that of the internally generated spindle posi- tioning pulses, and actuates the motor until the pulses are of equal duration. The microwave probe in the feed horn waveguide is then positioned vertically. Similarly, when S3 is opened, Pi is in- cluded in the R-C timing circuit of ICi. Due to the higher total resistance, ICi supplies pulses with a duration of about 2 ms, so that the waveguide probe is rotated over 90° for reception of horizontally polarized signals. The control circuit and the servo motor are powered from a regulated 5 V supply, which is simple to construct around a Type 7805 3-pin integrated regulator. In the case of the above men- tioned Indoor Unit, the input of the 7805 can be connected to the input of IC7 (Type 7812 on the vision/sound/PSU board). Due care should be taken, how- ever, not to overload the mains trans- former, Tri, or optional series resistor Rx, since the maximum current con- sumption of a blocked polarotor motor is typically about 300 mA. In some cases, it may be necessary to fit a rela- tively large electrolytic decoupling ca- pacitor direct across the supply ter- minals of the servo motor. The value of this capacitor depends on the actual current consumption of the motor, but 470 11F should work satisfactorily in most cases. It is recommended to use fairly stout wire for connecting the polarotor to the control circuit. The circuit is simple to set up: connect an oscilloscope to the pulse output line, and adjust P: and P2 for correct dur- ation of the rectangular output pulses (note that the settings interact). Open the available polarotor to check that the travel of the probe covers the full range of 90°. In the absence of an oscilloscope, Pi and P2 are adjusted until the servo motor works reliably over the full range in both directions of travel. Polarization offset correction can be achieved by ad- justing tl\p presets accordingly. Con- tinuous adjustment of the probe pos- ition (skew) for satellite reception ex- periments can be achieved by using potentiometers instead of presets in positions P: and P2. Current consump- tion of the control circuit is about 7 mA. b, 1988 9.77 A number of appliances, such as an EPROM programmer, require a supply voltage that can be switched tb£ variety of levels. The proposed circuit enables the user to do so between S V and 21 V. As soon as the switching transistor con- ducts, R3 is connected in parallel with R2. This lowers the total resistance be- tween the ‘adj’ pin of the LM317 and earth, and consequently the output voltage. It is possible to add a number of switching transistors and associated resistors and capacitor to the circuit to increase the number of available output voltage levels. The level of the output voltage depends on the ratio between Ri and the resulting value of R2 in parallel with R3. The p.d. across Ri is always 1.2 V. Thus, U 0 = [1+5^2] volts Capacitors Ci and C3 serve to optimize the switching behaviour of the circuit. The value of these components has to be established with the aid of a square- wave generator and an oscilloscope. The effect of these capacitors on the graph. An additional advantage of the use of an integrated voltage regulator is that this affords a means of current limiting. If, for instance, the ‘L’ type of this IC is used, current limiting starts at about 100 mA. This magnitude of current will be more than adequate for most EPROMs. Finally, it is possible to replace Ti and Ra by a high-voltage open-collector TTL gate, such as provided by the 7407. The proposed AVC gives weaker com- 4 ponents of the input signal extra amplifi- cation while ensuring that this dynamic compression is not disconcerting. It therefore eliminates those annoying dif- ferences in loudness between speech and music on radio and television. The principle of the circuit is fairly simple. Field-effect transistor Ti is used as a variable resistance. The value of this resistance, rus(on), can vary from infinity to about 1S0R. It is in parallel with R3 and, in conjunction with Ra, determines the gain of Ai. Without the effect of the FET, the gain of Ai is about 20 dB. Opamp A2 is connected as a straightfor- ward amplifier, whose gain may be varied by Pi. The negative part of the output signal of A2 is connected to the gate of Ti via a rectifier formed by Di, Ci, R7, and Ra. Resistor Ra ensures that the switching of Ti happens gradually. This means that it takes a short time before Ti operates; in other words, momentary the signal level has to be kept as low as be processed with a distortion of not differences in input level do not affect possible (thanks to the use of opamps, greater than 0.6%. With an input of 1 V the overall gain. The reduction in gain there is no direct voltage across the r.ms., the signal-to-noise ratio is about also takes place gradually, because Ci drain-source junction). An attenuator, 70 dB. has to discharge via R7. R1-R2, which gives an attenuation of The amplification in Ai and A2 compen- Because the resistance of Ti is influ- 40 dB, is therefore provided at the input, sates the losses in the attenuator: the enced by the drain-source voltage, Uds, This enables signals of up to 1 V r.mi. to total gain of the circuit, with Ti switched 9.78. off, is 0 dR Network R9-C4 is a high-pass filter which ensures that strong bass signals do not affect the control function to much ex- tent. The cross-over point may be altered to personal taste. Signals at a level below that set by Pi are amplified by a factor of up to 6.9 (gain=17dB). Fig. 2 shows the relation between input and output levels. The circuit needs a supply voltage of ±15 V and draws a current of about 6 mA. SAMPLE & HOLD FOR ANALOGUE SIGNALS Conventional analogue sample and hold circuits are notorious for their tendency to drift, a phenomenon unknown in digi- tal memories. It is, therefore, interesting to study the use of a digital memory el- ement for storing an analogue signal. The present circuit is based on in- termediate storage of digitized analogue information, and therefore re- quires an analogue-to-digital converter (ADC) at the input, and a digital-to- analogue converter (DAC) at the output. Unfortunately, DACs and ADCs are typically expensive components, and the present circuit is therefore set up with a DAC only, driven by an up/down counter— see Fig. 1. The counter is es- sentially an ADC, since the output voltage of the R-2R based DAC is con- tinuously compared to the input voltage with the aid of a window comparator. The error signal produced by the com- parator arranges for the counter to count up or down, depending on the magnitude of the difference between the input and output voltage. The up/down counter is corrected until the input and output voltage are equal. The digitized result of the A-D conversion is available at the counter outputs. The extensions for converting the basic set-up into a sample & hold circuit are relatively simple. The current count is retained by activating the HOLD input, which enables halting the U/D counter. Evidently, the counter state is not sub- ject to drift, so that the analogue output >9.79 signal is available unaffected for as long as the circuit is powered. The converter used here is the Type ZN43S ADC/DAC from Ferranti. This chip contains everything shown in the dashed box of Fig. 1. With reference to the practical circuit diagram, Fig. 2, the internal voltage reference and the oscillator are adjusted with R1-C1 and R2-C2 respect- ively. The latter are dimensioned for 400 kHz, ie., nearly the maximum oscil- lator operating frequency. The internal counte r is controlled via inputs up, down and mode. The logic level applied to the mode input determines whether the counter continues or halts upon reaching state 0 or the maximum value, 255. In the present application, the counter is halted. Gates Ni and N2 are added to enable blocking the U/D counter. Opamps Ai-A 2 form the win- dow comparator. Current source Ti-R? and Re arrange for the toggle threshold of Ai to be 20 mV higher than that of A2. This off-set creates the window, or inac- tive span, needed to suppress oscil- lation of the counter’s LS bit, and to pre- vent unwanted effects arising from the comparators’ offset voltages. Decoup- ling capacitor Cs is fitted for sup- pressing spikes that occur during state changes on the counter outputs. The 2 conversion time of this design is about slew rate of 4 rriW/is at the input. Finally, 640 /js, as determined by the oscillator bear in mind that the output impedance frequency (400 kHz), the resolution (8 (ICi, pin 11) is relatively high at about bits) and the input voltage change 4 kS. (2.55 Vpp max.). This corresponds to a LOW-FREQUENCY LC OSCILLATOR It is not always appreciated that LC cir- cuits may be used for generating low frequencies. The proposed circuit, pro- vided it uses good-quality components, can be used for frequencies down to 150 Hz, and possibly even slightly lower. The oscillator proper consists of Ti and T2 with the LC circuit connected in the collector circuit of T2. The amplification is set with the aid of the current source around Te. The voltage across the tuned circuit is tapped at high impedance and amplified by Ts. The output of this FET is buffered by T3 and then rectified by Di— D2. The resulting direct voltage is used for driving the current source. Since the rectified voltage still contains a ripple, a further buffer, Ti, is added at the output of the circuit. The circuit draws a current of about 20 mA, which can rise to about 25 mA at higher frequencies. Its output im- pedance has been kept as low as poss- ible to render the bandwidth of the os- cillator as broad as possible. Fairly high values of inductance may be used, provided the Q is of a reasonable value. Capacitor values may go up to 10 ^F, but note that electrolytic types can not be used. In the prototype, Li had a value of 150 mH and Ci was 6/18: the resulting frequency was 150 Hz. The oscillator generates pure sine waves up to 7 to 8 MHz and operates well up to about 30 MHz; the waveshape is then no longer a pure sine wave, however. Oper- ation at still higher frequencies is poss- ible, but the output level then drops from the nominal 250 mV. The circuit may be used to measure unknown capacitors or inductors, pro- vided the other component in the LC network is known, with the aid of the formula f=l/2nl/LC. 0 SINGLE-CHIP 150 W AF . POWER AMPLIFIER The Type LM12 operational amplifier from National Semiconductor has at least one remarkable characteristic: its huge output current capability of about 10 A. The chip is housed in a 4-pin TO-3 enclosure; can handle peak powers up to 800 W, and has extensive internal pro- tective circuits to prevent damage caus- ed by current and voltage overloading, or by overheating. Peak operating tem- perature of the on-chip power output transistors is measured for controlling a limiter that forms part of a so-called dynamic safe area protection circuit. The power output stage is not connec- ted to the relevant pin until the supply voltage exceeds 14 V (±7 V). Output disconnection is automatic when the chip temperature rises above 150 °C. It is possible to connect LM12s in parallel, or in a bridge configuration, for very high power applications (voltage regulators, automotive drivers, stepper motor or power servo controllers, etc.). The present application discusses the use of the LM12 in a high-power AF amplifier. The circuit diagram shows two clamp- ing diodes at the chip output. These prevent the output voltage swing ex- ceeding the supply voltage when the push-pull output stage in the chip is overdriven, and the output load is mainly inductive. The diodes also pro- tect the chip when the output is short- circuited to the positive or negative supply rail. The Type LM12CL or LM12C may be used with supplies up to +30 V or ±40 V respectively. Parts list Resistors (±5%): Rt;R3-10K R2-220K R4 = 2R2; 4 W Capacitors: Cl = 1(i0; MKT C2 = 6p8 C3:C4= 100,u; 40 V Semiconductors: Dl;D2 = BY229 ICt = LM12 (National Semiconductor! Miscellaneous: Large heat-sink for ICt CS1.5 °C|. Insulating material for ICi. PCS Type 884080 Input bias currents are compensated because the circuit is laid out for virtu- ally equal impedance at the inverting and non-inverting input of the opamp. Input offset is 20 mV maximum. If this is considered too high, it can be cancell- ed completely by applying an appro- priate offset compensation voltage to one of the inputs (use a well-decoupled potential divider). Output offset voltage in a number of prototypes without com- pensation circuitry was between 100 and 200 mV. Half-power (-3 dB) bandwidth of the amplifier is 16 Hz to 40 kHz; distortion is approximately 0.02% at Po = 1 W and J?i=2Q or 4 Q. At full drive, distortion increases to 0.05% (K>= +30 V; i?L=4 Q). Maximum current is supplied to a 2 Q load, but distortion then increases to 0.1%. Quiescent current of the amplifier is be- tween 65 and 100 mA. Inductor Li is wound as 40 turns of 1 mm dia. enam- elled copper wire on power resistor R4. It serves mainly to ensure correct oper- ation of the feedback amplifier with capacitive loads, such as large voice coils and loudspeaker cross-over filters. It will be clear that the supply for the amplifier must be capable of handling the peak current requirement of the LM12. For the LM12CL, it is rec- ommended to use a toroid mains trans- former with a 2x22 V secondary winding (150 W can then be supplied to a 2 S load only). Depending on the ap- plication and the output power re- quired, the transformer's secondary 12 A. Smoothing capacitors in the sym- metrical supply should be not smaller than 20,000 /iF on each rail. Finally, ICi should be bolted on to a large heat-sink, from which it is elec- trically insulated. >9.81 It AUXILIARY NEGATIVE- VOLTAGE SOURCE Many circuits require, apart from the usual positive voltage source, a negative supply from which only a smalkcurrent is drawn. In such cases, a mains trans- former with twin secondary winding would be a rather too costly solution. The circuit proposed here generates a negative potential from a positive supply. This supply may provide be- tween 5 and 15 V. If the current drawn from this supply is smaller than 1 mA, the level of the negative voltage generated lies about 1.8 V below that of the supply voltage. Thus, if the supply is 5 V, the negative potential is - 3.5 V. When a current of 2 mA is drawn from the supply, the difference between the two voltages increases to about 2.5 V. Operation of the circuit is fairly simple. Gate Ni, in conjunction with parallel- connected gates N2 to Ns incl., functions as a square-wave generator with buf- fered output. The peak -to -peak value of the square-wave voltage is, due to the use of CMOS gates, very nearly equal to the supply voltage. Rectifier Di-Da en- sures that the alternating voltage is con- verted into a steady negative one. If a clock frequency between 10 and 50 kHz is available, this can be applied to the input of N>. Capacitor C> and Ri are then not required. (Intersil application) 2 H SINGLE-CHIP SOLID-STATE RELAY Light-duty (25 to 600 W) solid-state relays -J have recently been introduced on the market by Sharp. These small and com- pact devices switch accurately at the zero-crossing and provide the required electrical separation. The photograph shows clearly that switching occurs exactly at the zero-crossing. This prevents switch-on currents of lamps becoming large and so extends the life of the lamp. The breakdown voltage of the triac sec- tion is 2 kV and the pins are on a 0.1 in grid. The relay requires an energizing cur- rent of 10 mA at 1.4 V, but with inductive loads about 25 mA is necessary. The additional components shown in the diagram make the relay more universally usable. Diode Di prevents the IC being damaged if the input is connected incorrectly. Transistor Ti sets the trigger current to precisely 10 mA. The RC network at the output protects the triac from sharp voltage peaks. The IC may be used without heat sink to switch currents up to 1 A. For switching larger currents, up to a maximum of 3 A, a 2mm thick 100x100 mm heat sink should be used. 9.82 This timer can be set to count a maxi- mum of 60 hours. It also allows an inter- val to be set. When this interval is reached, a buzzer sounds. The larger part of the circuit is con- tained in an Intersil Type ICM7217 four- digit CMOS up/down counter and dis- play. Circuit ICa is the clock that generates a square wave with a period of 1 s. The clock signal is available at pin 3 (013). The clock signal may be divided by 60 in ICa if it is required to time more than 1 hour. When Se is closed, the supply is switched on and ICi is reset via Rs and Cs. The position of Sa determines whether minutes or seconds are counted: maximum 59 h 59 min (pos 2) or 59 min 59 sec (pos 1). If, for instance, a total time of 35 min with an interval at 20 min is to be counted, Sa is set to position 1. Thumb wheel switches S? to S10 are then set for a dis- play reading of 20.00. Briefly pressing Sj stores this setting in the memory of ICi. Then S7 to S10 are set for a display reading of 35.00. During these settings, Si should be open. Pressing S2 causes the ICM7217C to count down from 35.00. When display reading 18.00 is reached, the buzzer briefly sounds (energized via Nj and Na). The timer may then be stopped by closing Si. When Si is opened again, the timer restarts the down count to 00.00. When that reading is reached, the buzzer sounds briefly again. Note that at any time during the count down the timer may be stopped by closing Si. The timer is reset with Ss; when that hap- pens, the buzzer sounds briefly and the display reads 00.00. The set count down period of 35 min is, however, retained in the memory until a new period is pro- grammed. The current drawn by the timer, includ- ing the displays, is about 100 mA. If a battery supply is used, it is possible to switch off the displays when the timer is counting by adding a switch (with single break contact) between points A and B. This switch enables the display to be read briefly. With the displays switched off, the current drawn is of the order of only 4 mA. Do not set the thumb wheel switches to readings greater than 59.59, because the timer will then no longer count cor- rectly. FLASHING LIGHT This is a rather unusual application of the Type 317 voltage regulator. With only a handful of external components, it can be used for flashing a small 12 V lamp. The output voltage is not stabiliz- ed by the circuit: it is simply a few volts lower than the input voltage. The 317 is capable of delivering more than 1 A. The circuit automatically limits the switch-on current, so that lamp life is considerably extended. The waveforms at the four major points in the circuit are shown in the accompanying photo- graph. The component values given result in a flash frequency of about 4 Hz. Flashing can be stopped by driving Ti with a voltage of more than + 1 V. Source: Lambda Power Supply Hand- book IC1 LM317 0 AMPLITUDE-MODULATED CALIBRATION GENERATOR A calibration generator is used for quickly checking receiver operation. The design shown here generates RF signals (markers) at 1 MHz intervals over a frequency extending up to about 2 GHz. These signals can be amplitude- modulated by driving T4 with a sine wave generator. A stable 2 MHz oscillator is set up around Xi and Ti. MOSFET T2 functions as a digital buffer for clocking bistable/divider FFi. Pulses at the out- put of FF2 have a frequency of 1 MHz and a width of only 12 ns, which is ob- tained by FF2 clearing itself after output Q has gone low. The pulses drive T3 into saturation. This SHF transistor consequently produces a wide spec- trum of harmonics, and its class C set- ting causes it to function as a frequency multiplier. The collector current can be modulated via series transistor Ti. Since the two sidebands generated in the pro- cess of amplitude modulation are offset from the carrier by the modulation fre- quency, AM can be used to generate signals at frequencies in between the markers. Example: modulating the cali- bration generator with a 204 kHz sine wave gives two additional frequencies adjacent to the marker at, say, 1120 MHz: 1120-0.204 = 1119.796 MHz and 1120 + 0.204 = 1120.204 MHz. Hence, a con- tinuous tuning range from 1 MHz to 2 GHz is obtained when the sine wave generator output frequency is adjust- able between 500 kHz and 1 MHz. The measured amplitudes of four markers produced by the calibration generator show that available output levels fall with increasing frequency: /=100 MHz: Po= -25 dBm 7=400 MHz: Po=-45dBm 7=1.0 GHz: Po= -5S dBm 7=1.8 GHz: Po= -70 dBm Note: 0 dBm = 1 mW in 50 S. Construction of the calibration gener- ator is straightforward even for those with limited experience in building RF circuits. It is essential that close- tolerance (2.5 or 5%) polystyrene capacitors be used in positions Ci, C2 and C<. Inductor Li is wound as 3 turns 0.2 or 0.3 mm dia. enamelled copper wire through a small (3 to 5 mm long) fer- rite bead. Be careful to avoid short- circuits between the windings as the enamel coating may be damaged when the wire is pulled through the hole in the bead. The calibration generator is powered from a 6 V battery pack so that it can be used as a portable test instrument. Cur- rent consumption is less than 20 mA. B R2 = 33K R3 = 47K R4 = 1K0 Rs;R6;R7 = 22K Re = 56R Capacitors: * = 470p' C2>22p C3 = 40p toil trimmer C7 = 4p7 C8 = 390p C9 = %7; 16 V Semiconductors: Di = 1N4148 ICl=74HCT74 Ti=BF494 T2 = BF981 or BF982 T3 = BFG65 (Philips/Mullard) T4 = BC550B Miscellaneous: Xi = 2 MHz quartz crystal; 30pF parallel PCB Type 884054: SIMPLE PHONO PREAMPLIFIER This circuit shows that a preamplifier for magneto-dynamic cartridges can be relatively simple without seriously com- promising compliance to the IEC stan- dard in respect of frequency response. Compared to the RIAA standard, the IEC frequency curve has an additional roll-off point at 20 Hz— see Fig. 1. The circuit diagram of Fig. 3 shows that input and output of the preamplifier based around the Type TIi071 oper- ational amplifier are direct coupled, making it possible to accurately define the previously mentioned roll-off by means of network R2-C3. Output offset of the preamplifier is about 3 mV. Output capacitor C< can be fined if this offset voltage can not be handled by the input of the line or power amplifier. , For optimum compliance with the IEC frequency curve it is recommended to use close tolerance polystyrene (Siemens Styroflex) capacitors in pos- itions Ci and C2, and an MKT capacitor in position C3. Resistors are preferably high-stability metal film types from the E48 or E96 series, although less expens- ive and commonly available types from the E12 series may also be used with reasonable results when selected for the required resistance with the aid of a digital ohmmeter. It was with this in mind that Ra has been dimensioned at 5K62 (E12: 5K6). This value gives a roll- off at 18.9 Hz instead of the required 20.0 Hz,» so that the low-frequency response (up to 50 Hz) of the preampli- fier deviates slightly from the IEC curve. The deviation, A, of the amplification with respect to the values set by the IEC is shown as a function of frequency in Fig. 2. A prototype of the preamplifier built with the component values given in the circuit diagram gave the following test B( 1988 9.85 results: voltage gain 39 dB at 1 kHz; signal-to-noise ratio greater than 70 dB at 1 kHz and 100 mV output signal (up to 80 Hz: greater than 60 dB). The input was connected to a test generator which supplied 1 mVrms at an output im- pedance of 1 kQ. The circuit should be fed from a well- regulated symmetrical supply (preferably ±1SV, but ±12V or +8V should also work). A suitable supply is simple to build around two integrated regulators such as the 78Lxx and 79Lxx types, which can step down supply voltages already available in the line or power amplifier. Current consumption of the preamplifier is only 2 mA. ALTERNATING CURRENT SOURCE One of the less known properties of no voltage difference between the gate -J field effect transistors is that some of and the source, so that the FET functions these are electrically symmetrical, as a current source as shown above, which means that the drain and source The constant alternating current sup- may be interchanged under certain con- plied by the circuit can be defined by ditions. This circuit is based on this fitting small resistors in the drain and phenonemenon, and feeds a constant source lines, so that Vcs is set to values alternating current through P2 when other than 0 V. The input voltage range connected to an alternating voltage of the current source is 6 Vims to 18 Vrms. voltage on it equals Vq there is, again, A VOLTAGE-CONTROLLED SHF OSCILLATOR This oscillator supplies an output level between -10 dBm and +3 dBm, and can be tuned between 1250 MHz and 1800 MHz simply by varying the supply voltage. Operation of the circuit is based on the fact that the transition fre- quency, fr, of the BFG65 is reduced when the collector current rises above 10 mA. The oscillation frequency is also determined by the physical layout of in- ductor L3, which is a strip line made from two parallel running lengths of 1 mm dia. silver plated wire. The length is established experimentally, starting from 13 mm. Chokes Li and Lz are 3 turns of thin enamelled copper wire (dia. 0.2 or 0.3 mm) through a small (3 mm) ferrite bead. Capacitors Cz and C3 are leadless ceramic types (rec- tangular or disc). The SHF test oscillator is ideal for quickly finding the maximum usable in- put frequency of, for instance, a fre- quency meter specified to reach up to 1.2 GHz. In addition, it can be used for testing RF input sections in indoor units for satellite TV reception. COMPUTER-DRIVEN POWER CONTROLLER This circuit enables a computer to con- trol the power supplied to a mains oper- ated device (lamp, heater, drill, etc.) in 255 steps. Variation of power is achieved by controlling the voltage supplied to the load (Rl in the circuit diagram of Fig. 2). A conventional power regulator is used here, composed of a triac and a simple associated circuit to control the phase angle at which the triac is trig- The power supply and mains trigger cir- cuitry are shown in Fig. 1. The circuit around Ti. . .T4 incl. and ICi is a zero- crossing detector which produces an active high pulse every time the mains voltage is zero. Opto-coupler ICi in- sulates the rest of the circuit from the With reference to Fig. 2, Schmitt-trigger Ni inverts the zero-crossing pulses, causing 8-bit binary down counter IC2 to load the 8-bit word applied to counter preset (jam) inputs J0. . ,J7. The counter is decremented one count by each clock pulse supplied by oscillator Nz. When counter state nought is reached, output ZD goes low, and N3 inhibits further clocking of ICz. Simultaneously, N4 produces an output pulse, so that Ts conducts and fires the triac. As the triac is only fired when ICz counts to zero, the instant at which this happens depends on the value of the 8- bit control word received from a com- puter. Hence, the time that lapses be- tween the zero crossing instant and the triac firing instant is a function of the magnitude of the control word. The greater the 8-bit word, the greater the phase angle, and the less power is delivered to the load. Inductor Li suppresses RF interference caused by the triac, and should be able to carry at least S A. The triac in this cir- cuit can be a TIC206D (4 A) or a*TIC216D (8 A). Other types may be used if these are known to trigger at a gate current of less than 10 mA. The value of R12 is de- termined empirically, and should be as high as possible without causing the disappearance, on point A, of pulses with an amplitude of 6 Vp. The only adjustment required is that of Pi. If complete switching off of the load is required, this preset is adjusted for 0 V indicated by an AC voltmeter con- nected instead of the load, with data FFh (285 10) written to the power con- troller. If regulation from 0 V onwards is not desired, Pi is adjusted so that the meter reads the required minimum voltage. When writing programmes for the power controller, it should be remembered that the power delivered to the load is an inverse function of the value written into the computer’s output port. Safety precautions: The shaded parts in the circuit diagrams are operated at mains potential, and must never be touched while the unit is being powered. Great attention should be paid to proper insulation in the selecting and mounting of the parts within the shaded areas. It is strongly recommended to bend the pins of the optocoupler away from the package to ensure an insulation distance of at least Finally, it should be noted that the cir- same result as FFh, namely minimum cuit may not operate correctly with voltage applied to the load. Regulation loads below about 40 W, and that efectively starts with data 01h. writing 00h to the data input has the FIVE-BAND STEREO GRAPHIC EQUALIZER I other four This design of a stereo equalizer is fairly unusual because it is based on induc- tive feedback. In theory, the feedback circuit around opamp Ai would provide 15 dB amplification or attenuation of each frequency range, but in practice only about 13 dB is attainable owing to losses in the inductors. A virtually flat frequency response is obtained when all five potentiometers Pi to Ps are set to the centre position (0 dB). Total control range of the unit is about 33 dB. The TL072 dual opamp in each channel is a trade-off between cost and perform- ance in respect of noise and distortion. Set to 0 dB gain, a prototype of the equalizer produced 0.04% distortion at an input signal of 1 kHz; 1 V, and 0.13% at 5 and 10 kHz. Distortion is highest when the test frequency lies within one ] band that is fully attenuated while the ' . — maximum gain. In this condition, test measurements resulted in a maximum distortion of 1 1.5%, which is certainly tolerable given the simplicity of the circuit. Signal-to- noise ratio is greater than 90 dB at an in- put amplitude of 1 V. The frequency response curves were obtained with the following settings: curve 1: all controls set to maximum; curve 2: 4 controls set to 0 dB, and 1 to 3: 4 controls curve 4: all controls set to minimum. Due attention should be paid to the DC resistance of the inductors. The total re- sistance of the inductor and series resistors in each feedback network should remain 680 S, so that R3 to R12 ,incl. may have to be dimensioned dif- (Cirkit stock no. 34-10513). ' L2.U' = 680 mH. e.g. Toko Type 293LY-684 'Cirkit stock no. 34-68413). -3;L3' = 150 mH. e.g. Toko Type 293LY-154 'Cirkit stock no. 34-154131. U;U' = 68 mH, e.g. Toko Type 181LY-683 - iCirkit stock no. 34-68302). i-S:L5’ = 10 mH, e.g. Toko Type 181LY-103 iCirkit stock no. 34-10302). ferently than shown in the circuit diagram. Always measure the resistance of the inductors used, and then calculate the value of the resistor re- quired to obtain a total of 680 Q. Example: a Type 239LY-1S4 ISO mH in- ductor from Toko was found to have a DC resistance of 37 Q, requiring a series resistor of 680 - 37 = 643 Q. This value is approximated with the aid of a 680 Q and 12 kQ resistor in parallel (R7-R8 in the circuit diagram). Ferrite- encapsulated inductors are rec- ommended to reduce magnetic coup- ling, and to keep crosstalk at relatively high frequencies down to an acceptable level (<-60 dB at 10 kHz). *C::ICr=TL072 =CS Type 884049 BP QUIZ TIMER Here is a simple ‘who’s the first’ circuit that can be used in quiz games with up to eight participants or groups of par- ticipants. The circuit indicates the first one to press his key by a glowing LED against his number or any other identifi- cation used in the quiz or game. At the same time, the circuit gives an audible indication that some key has been pressed. The reset key enables the quiz master to restore the circuit’s original state before updating the score and pro- ceeding with the next question or assignment. After reset has been pressed, the eight R-S bistables in ICi and IC2 are reset. The 0 outputs all go logic low and, consequently, the output of IC3 goes logic high. The circuit is now ready to be operated. For example, if S\ is pressed first, the first bistable is set and output 01 goes high. The output of IC3 pulls the common line of keys Si. . .Sa incl. logic low to prevent more bistables being set. Hence, Q1 is the only output that is logic high. This condition is indi- cated by LED Di. Simultaneously, Ti is biased and switches on the buzzer to at- tract the attention of the quiz master. Capacitors Ci to Cs incl. prevent the bistables being set permanently if a key is kept pressed for a long time. Finally, Sa is pressed to reset the circuit. This causes all Q outputs to be made logic low, and the common key line high, returning the circuit to its original con- The circuit is not critical in respect of supply voltage, which is preferably the working voltage of the active piezo- buzzer (6 V or 12 V). Current consump- tion in the de-activated state is less than 1 mA, while less than 28 mA is drawn when one of the LEDs is illuminated. The supply voltage need not be regulated, making it possible to use an inexpensive mains adapter of the DC type. HEADLIGHTS INDICATOR It is never advisable to leave a car’s headlights on for long periods when the engine is not running. Yet, especially during the winter months, many of us in- advertently do this. The indicator de- scribed here helps to prevent you suf- fering the consequent and inevitable flat battery. In its simplest form, the indicator con- sists of a d.c. buzzer and a diode as shown in Fig. la. With the headlights on (Sa closed), the interior light (Li) does not come on until one of the front doors is opened (when either Si or Sj closes). At the same time, the buzzer is ener- gized and sounds. Either closing the door or switching off the headlights (S3) turns the buzzer off. The diode in series with the buzzer is necessary because B is normally at + 12 V via Li and A at ground via the headlights (L2— only one shown here). The buzzer should then not sound, of course. A slight modification to the circuit, en- abling it to operate only on the switch in the driver's door, is shown in Fig. lb (where it is assumed that Si is the rel- evant switch). The buzzer draws a current of only about 10 mA when energized. 9.90 . POWER MULTIVIBRATOR This simple multivibrator circuit is remarkable for its high efficiency and ability to drive relatively heavy loads. The circuit supplies a symmetrical rec- tangular signal that floats with respect to the supply voltage. An astable multivibrator is formed by Ts, T6, Ri, R2, C) and C2. The collector currents of Ts and T6 drive Ti and T2 respectively, while the emitter currents drive T3 and T4 respectively. Current limiting may be dimensioned to requirement by chang- ing Rt. It should be noted, however, that the transistors may carry relatively high currents. Their current amplification, hFE, is, therefore, fairly low, so that the current limit point can be approximated with hFE(max)(£/b — 1.4)/Ri. With Ri=68 Q as shown in the circuit diagram, the multivibrator can be used for switching loads up to about 3 A. Output frequency of the oscillator is approximated by 0.7/(R2C), and is about 53 Hz with R2 = 68kQ, C = Ci=C 2 =220nF and £/b=12 V (14 V: 50 Hz). One of the many applications of the power multivibrator is a battery- operated mains converter. Its outputs are then connected to the low-voltage secondary winding of a mains trans- former. A prototype of the multivibrator was dimensioned for relatively high out- put current at 50 Hz by fitting Ri=33 Q; R2 = 2x68kQ in parallel, and C = 2 x 220 nF in parallel. Connected to a 9.5 V; 5 A mains transformer, it powered a 40 W mains bulb with a rectangular voltage of nearly 240 Vims. Supply voltage and current consumption were 14 V and 6 A respectively, yielding an acceptable efficiency of about 40%. Quiescent current consumption of the circuit is determined by Ri, and was 0.3 A in the test set-up. When the multivibrator is used for driv- ing an inductive load, as in the above ap- plication, each output transistor must be protected from inductive voltage peaks by two fast high-current diodes fitted in reverse across the collector and emitter terminals. | TOUCH SENSITIVE LIGHT SWITCH This low-cost circuit enables turning -\ room lights on and off simply by touching a round metal sensor. The light is turned on by briefly touching the sensor, and off again by touching it slightly longer. With reference to the circuit diagram, when the sensor is briefly touched, hum and noise induced on the body are amplified by cascaded gates Ni, N2 and N3. A pulse train with a swing of nearly the supply voltage (4.7 V) and a frequency equal to that of the mains voltage (50 or 60 Hz) is ap- plied to a bistable set up around Ni and Ns. C2 is charged via D2, and the bistable latches in a high output state. Triac Trii is triggered via driver Ti, so that the lamp lights. When the sensor is touched for about 2 seconds or longer, the pulse train charges Ci via Rs and Di. Inverter N6 pulls the input of Nt low when the voltage on Cl is sufficiently high. Bistable N4-Ns toggles and Ti breaks the gate current for the triac, so that the lamp is turned off. The circuit also works in a relatively noise-free environ- ment. When the user forms a relatively low resistance to ground, the input of Ni is effectively pulled low by R1-R2, whose total resistance is low relative to R3-R4. The effect of this on the bistable and triac circuit is similar to that outlin- ed above. A suggested construction of the sensor and LED is shown in the accompanying drawing. The LED is fitted in a plastic holder, and in the dark indicates the lo- cation of the light switch. The LED holder (C) is secured in the side or top panel (A) of the ABS enclosure that houses the light switch circuit. The LED is located in a thin aluminium or brass washer (B), which is connected to Ri, and glued onto the outside surface of the plate. In the interest of safety, it is recommended to observe a minimum distance of 7 mm between the LED and Ri. In this context, constructors are urgently advised not to use a metal or metallized LED holder as the sensor. Also, never replace Ri and R2 with a single 4M7 resistor. Since this circuit is connected direct to the mains, it must be fitted in a safe and sound ABS enclosure that is impossible to open without it being deliberately damaged. Once more we advise that the presence of the mains voltage is a serious source of danger, so that the first and foremost concern of every con- structor should be absolute safety. 1 9.91 N1...N6 = IC1 = 4069BE || PRINTER SHARING BOX This simple circuit makes it possible to connect two computers to a single printer. Toggle switch S2 selects the rel- evant computer by applying the appro- priate logic level to the enable inputs, G, of octal bus transceivers Type 74LS641 (ICi. . .IC4 incl.). The direction input, DIR, of these is hard-wired to +5 V, so that the data direction is from An to Bn. When 5 is logic high, the buffers are switched to the high impedance state, so that chip outputs can be con- nected to form a bus structure. With this in mind it is relatively simple to see that the circuit is the electronic equivalent of a 16-way toggle switch. Input BUSY of the non-used computer is held logic high to prevent this machine attempting to send data when the other computer is accessing the printer. The 74LS641 was chosen because it has open-collector outputs — the reason for this should be clear when it is remembered that the Centronics stan- dard dictates the presence of pull-up resistors in the printer. The 74LS641s, of course, need pull-up resistors at the computer side also, and these are formed by resistor networks R3, R4, Re and R7. A RESET switch, Si, is provided to clear the printer buffer by means of an INPUT-PRIME pulse should the user find out that the wrong file is being printed. This reset option is definitely neater than switching off the printer completely to correct the error. The circuit is conveniently powered from the 5 V supply in the printer. In most cases, this supply voltage is available on pin 18 of the 36-way Cen- tronics input connector, but this would have to be ascertained by measuring and reference to the printer manual. It is recommended to connect +5 V to non- used pins 15 and 34 also to distribute the current over several wires in the Cen- tronics cable. Once again, check the o isjo 00 OOO OOO 000000000)36 P JL P P 9 P MooooooooooooooooooI -'b ' — < 1 «o 0B0 printer manual to see whether these pins are actually available for this pur- pose. The printer sharing box will generally be located close to the printer. PCB con- nectors Ei, K2 and K3 are 36-way straight headers. Three cables are required for connecting the completed PCB be- tween the computers and the printer. Two 10— 15 cm long adaptor cables are made from flatcable with IDC (press-on) connectors at either end. One end is ter- minated in a 36-way IDC socket for plug- ging onto the PCB header, the other in a female IDC Centronics socket (blue rib- bon type) for receiving the printer The short printer output cable is com- posed of a female 36-way IDC socket as above, and a 36-way male Centronics plug. Current consumption of the printer switch is about 200 mA. 0 3 I OMA-2500 TIME STANDARD RECEIVER OMA-2500 is a 1 kW time standard trans- mitter on 2500 kHz. The station is located in Liblice, Czechoslovakia, and is operated by the Astronomical Insti- tute of the Czechoslovak Academy of Sciences. Contrary to time standard transmitters in the VLF band (DCF77, HBF), modulation is pure AM instead of a combination of AM and PSK or FSK. This means that the seconds pips transmitted by OMA-2500 are free from phase noise, which is a must for some types of PLL, particularly in communications equipment, where the 2500 kHz signal supplied by a time standard receiver is used for generating or deriving other frequencies of equal stability. Transistor Ti is configured as a regenerative buffer which acts as an ac- tive filter with an effective Q (quality) factor of about 1,000 at a 3 dB bandwidth of 2.5 kHz. The received signal is further raised in amplifier T2-T3 before it is ap- plied to active crystal filter T4-X1 which ensures- a 3 dB bandwidth of about 500 Hz. Output amplitude of the re- ceiver is sufficient for driving almost any type of simple PLL. The receiver is powered via the downlead cable at the output to enable it to be mounted in a noise-free environment. Inductor Li is wound as 2 turns (primary) and 50 turns (secondary) of 0.3 mm dia. enamelled copper wire on a Type T50-2 core. Quartz crystal Xi is a 2500 kHz type for series resonance. 2.5 MHz. Reduce the signal amplitude reception, when OMA-2500 is received Construction of the receiver should and redo the adjustment of the trimmer, with high field strength throughout follow the standard rules for RF circuits: Disconnect the function generator, and Europe. Daytime reception in western keep all connections as short as poss- connect the aerial. Connect the scope and northern Europe will mostly range ible, and use ample screening and to the output of the circuit. Peak Cs for from poor to just usable, depending on decoupling. optimum amplitude of the AM signal, propagation conditions and location of Adjustment: set a function generator to but make sure that this is not greater the receiver. 2.5 MHz at Uo = 10 mV. Connect the out- than 500 mV. Remember that Ti is a The circuit is fed from 12 V, and con- put to Ci. Connect an AC-coupled os- regenerative stage, so that the settings sumes about 10 mA. Finally, bear in cilloscope to the source of Ti, and peak of C2 and Cs interact. If necessary, re- mind that a good aerial Gong wire or Ca. It may be necessary to reduce or in- adjust the trimmers to ensure that the rhombic quad) is imperative for reliable crease the number of windings of the signal at the collector of T3 is stable, reception, secondary of Li to obtain resonance at and not clipped during night-time PROGRAMMABLE SWITCHING SEQUENCE The proposed control circuit is shown in the diagram as containing two relays, but this number may be increased if necessary. The switching sequence is determined by the time delay of an RC network at the input of a gate that is used to energize a relay via a darlington transistor. When Si is connected to the supply voltage (as shown in the diagram), the input capacitor, Ci, C2, .... begins to charge via a resistor, Rii, Ri2, .... and di- ode Di. After a given time, depending on the time constant of the relevant RC combination, the voltage across the ca- pacitor has reached a value sufficient to toggle the gate. The relevant transistor is then switched on, and the relay is ener- gized. By giving the input of each gate a differ- ent time constant, the sequence of switching is determined. When Si is switched to ground, the op- posite happens. Diode Di is reverse- biased and the capacitors, Ci, C2 9.94 olektorindia September 1988 N1 , N2 = 4584 " ^ ipi 55 4584 rfe^| |j>n rt>*] L Hltiiiikl}lMkl^ discharge via resistors Roi, R02,. . ., and diode D2. The discharge time constant determines how fast the capacitors can discharge and retoggle the gates. So, here again the switching sequence is determined by time constants. The gate with the shortest time constant will always toggle first. The supply voltage may lie between 5 V and 15 V, but must, of course, be equal to the operating voltage of the relays. Furthermore, the BC516s must not switch more than 400 mA, and this again influences the choice of relay. A good, practical energizing current for the relays is 200 mA. The values of resistors Ri and Ro may lie between 1 kQ and 10 MS; the value of capacitors Ci, C2,..., between 10 pF and 100 /;F. Time constants exceeding 1,000 seconds create problems in prac- tice, because the leakage current of the electrolytic capacitors then becomes comparable with the charging current. In general, choose the time constants so that two consecutive ones always differ by at least 0.1 s. BURST GENERATOR A burst generator is indispensable for testing the dynamic response of loudspeakers, and, in some cases, AF amplifiers. The fact that a number of cycles of a sinewave are applied to the loudspeaker under test, and not a con- tinuous signal, eliminates the adverse effects of reverberation, reflection and echoes which are otherwise caused by the test room, and are almost inevitably picked up by the test microphone. In addition, the burst provides a good indi- cation of the loudspeaker's perform- ance in respect of voice coil transient response, resonance, and ringing. The test signal provided by an external sinewave generator is switched on and off at or around the zero crossing, de- pending on the setting of phase control Pi. The pause amplitude can be set by P2, while controls Pa and Pi are used for adjusting the duration of the pause and the burst respectively. It should be noted that the settings of these poten- tiometers interact, so that an oscillo- scope is required for correct alignment. The duration of pause and burst is not related to the input signal. This means that the number of cycles supplied by the generator increases with the fre- quency of the sinewave applied to the input, unless, of course, P3 and P4 are re-adjusted. Comparator ICi converts the sinewave at the input into a rectangular signal. The switching takes place at a specific instantaneous amplitude of the sinewave, set by Pi. The timing of the switching instant is arranged by astable mulitivibrator IC2, and is copied in bistable IC3 on the first positive edge of the sinewave, since this corresponds to the rising edge of the clock signal. Out- put 0 goes high, so that the pole of elec- tronic toggle switch ICi is connected to pin 12, and hence carries the attenuated sinewave burst. The burst generator is not critical in respect of supply voltage, as long as this remains between +5 V and +9 V. Do not exceed ± 9 V on penalty of damag- 0 3 I/O EXTENSION FOR AMIGA 500 The Commodore Amiga is claimed to be a computer with plenty of facilities for extension circuits. The model 500, for instance, comes with no fewer than twelve connectors and sockets. There are, however, awkward constraints to the practical use of all these extension facilities. The serial connector is cumbersome to use with TTL circuits because of the ± 12 V logic levels on it. The use of the 86-pin connector on the machine is complex and risky because of the unbuffered connection to many internal signals. The one remaining op- tion is the PARALLEL CONNECTOR, Which can be extended to a maximum of 56 I/O lines as shown here, with the possi- bility to realize a bidirectional port. The circuit was designed and built for the Amiga 500 computer. It is likely to work equally well on models 1000 and 2000, but this has not been tested in practice. Output lines BUSY, P-OUT and SEL on the parallel connector can be pro- grammed to supply a 3-bit address selection code which is applied to binary decoder ICi. Octal bus buffer IC2 is the input port at address 2, latch IC4 the output port at address 3, and transceiver IC3 the bidirectional port at addresses 0 (read) and 1 (write). Th e re- maining 3 addresses (lines E4, E5 and E6) can be used for 3x8=24 additional I/O lines. Line 7 on ICi may not be used for selecting an input ot output port, and is used instead for driving ready LED Di when none of the ports on the I/O extension is being selected. It should be noted that IC3 is not a latch, which means that it can only output data for as long as it is written to by the micropro- cessor. Output port IC4 does have a lat- ching function, so that datawords are kept stable on the outputs until overwrit- ten by the microprocessor. The accompanying listing is intended as a guide to writing software for the I/O extension. As an example of the practi- cal use of the subroutines, instruction a=l:n=123:GOSUB Wr Init: POKE 125711365,199 POKE 125706245,255 POKE 125754895,0 RETURN 'call once after power-on ■BUSY, P-OUT and SEL = output bits 'select address 7 (light READY LED) 'set port to input sends dataword 123io to IC3, which then functions as an output port. Conversely, instruction a=2:GOSUB RdrPRINT n reads the dataword applied to IC3, and prints it on screen. Subroutine Init need only be called once at the beginning of the program- ming session. Input ports must not be written to. The I/O extension should be fed from a separate 5 V supply. Rd: POKE 125754895,0 POKE 125706245, 248+a n=PEEK( 125749775) POKE 125706245,255 RETURN Wr: POKE 125706245, 248+a POKE 125754895,255 POKE 125749775, n POKE 125706245,255 ' load contents of address 'set port to input 'select address a 'read value 'light READY LED in variable n 'store variable n in address 'select address a 'set port to output 'write value ' light READY LED RETURN UNIVERSAL SMD-TO-DIL ADAPTORS An increasing number of electronic components, and in particular inte- grated circuits, is now only available as surface-mount devices (SMDs). Circuit design on the basis of these leadless, tiny, components invariably poses prob- lems to many because there is no way to go round making a printed circuit board for building and testing prototypes. Making PCBs for SMD based designs is cumbersome and time-consuming. In many cases it will, therefore, be desirable to develop the circuit as it would have been done using ICs and components of standard size. The PCB adaptors introduced here make this possible. With the exception of the general-purpose type, they are slightly larger than ICs of normal size, but still fit in the generally adopted 0.1 in. raster. The adaptor PCBs effectively enable a range of SMD ICs to be handled just as their normal-size equivalents, and so alleviate the plight of designing and et- ching a new PCB for every experiment or minor change to the circuit. SMD ICs with 8, 14 or 16 pins are usually housed in a 'narrow' enclosure, and 16, 20, 24 and 28 pin types in a ‘wide’ en- closure. The printed circuit board shown here allows making multiple adaptors that can be used for fitting: • Narrow SMD ICs with a maximum of 16 pins. For 8 and 14 pin types, the PCB can be cut to the required length. • Wide SMD ICs with a maximum of 28 pins. PCB sections are cut off to the required length as above. • SMA transistors, capacitors and resistors. These are arranged in a DIL configuration on a general-purpose adaptor to enable fitting networks and circuit sections as complete modules on a standard prototyping board. The size of this adaptor does not exceed that of a standard 16-pin integrated circuit. Suitable lengths of terminal strip are pushed through the holes at the under- side of the boards to create pins for fit- ting the modules in standard IC sockets. TW WIPER DELAY This two-key wiper delay circuit is remarkable for its simplicity and ease of use. The wipe is started by pressing the set switch, which also serves to adjust the length of the wipe interval. The cir- cuit is turned off by pressing the reset button. The wiper delay shown in Fig. 1 consists of three opamps and a monostable multivibrator (MMV). Opamp Ai is set up as a triangular wave generator, con- trolled by the output of the MMV. When this is low, a slowly rising sawtooth voltage appears at the output of At. The rise time of the sawtooth depends on R2-C3. Opamp A3 compares the voltage across C4 to the instantaneous sawtooth amplitude. The output of A3 drops from 8 V to 0 V when the sawtooth voltage ex- ceeds U(C4i. This change in the output voltage of As is delayed by R 6 -Ce and passed to A2, so that the MMV is trig- gered somewhat later. The wipers are switched on via Ti and Re when pin 3 on the 555 goes high. Also, C3 is rapidly discharged via Di and Ri, while D2 prevents the voltage across C3 becom- ing positive. When the MMV output goes low, Ai generates a new sawtooth period. 1 9.97 When the circuit is first switched on, C4 2 is discharged, and the output of Ai is slightly higher than 0 V due to Vic-ei of the internal output transistor. This causes the outputs of A3 and As to re- main low, so that the wiper relay re- mains energized initially. When reset is pressed, C4 is charged via Rs, causing the the output of As to go high, and the MMV to be stopped. The delay circuit around A2 is necessary to prevent C4 being discharged completely after pressing the set button. The relay contacts should be wired such that the dashboard switch is by- passed when the relay is energized, and that the hold switch, H, for the wiper motor is opened— see Fig. 2. Due atten- tion should be given to the correct con- nection of the hold switch on penalty of short-circuiting the car battery. WIRELESS HEADPHONES LUB (TRAN SMITTER) A circuit for the transmission, with good quality, of the sound output of a TV re- ceiver over a couple of metres. The input signal for the circuit is taken from the headphone or video recorder output of the TV receiver. If these are not available, NEVER ATTEMPT TO FIT ONE YOURSELF: THE CHASSIS OF THE TV SET MAY BE AT A LETHAL HIGH VOLTAGE. The audio signal is amplified by ICi, which has been given some extra ‘body' by the addition of an output buffer, Ti. Capacitor C4 has a potential that is equal to half the supply voltage (via R3-R4), on to which the amplified audio signal is superimposed. The resulting varying direct voltage is used as the supply voltage for emit transistor T2 via the primary of L2. The carrier os- cillator, also formed by T2, can oscillate between 1,750 kHz and 3,500 kHz. The consequent amplitude-modulated signal across the secondary of L2 is strong enough to span a few metres. A ferrite rod is used as transmit antenna. Diode Di serves two functions: it in- dicates that the transmitter is ‘on’ and it stabilizes the direct voltage (about 1.5 V) for the oscillator. The supply voltage for the oscillator is thus independent of the 12—18 V line. The inductors are easily made. A T50-2 toroid with 80 turns of 0.2 mm dia. enam- elled copper wire is used for Li. A fer- rite rod of 10 to 20 cm is needed for L2: 0.5 mm dia. enamelled copper wire. Lza consists of three turns of 0.6 mm It is recommended to power the circuit dia. enamelled copper wire and should from a mains adaptor, because a current be wound at the ground side of L2B. This of up to 150 mA may be drawn, secondary winding consists of 30 turns WIRELESS HEADPHONES (RECEIVER) To arrive at a suitable headphone re- ceiver that meets the requirements of being light, battery-powered, and offer- ing good-quality reproduction, a Ferran- ti ZN41S was chosen. This IC contains a complete AM detec- tor, an output amplifier, and operates from a single 1.5 V battery. The circuit shows the ZN415 in its stan- dard application as a medium wave re- ceiver. Circuit Ci— Li is, however, tuned to a frequency above the medium wave band. The output stage drives a high- impedance headphone without any problems. The circuit draws a current not greater than 5 mA, which ensures a good battery life. The tuned circuit, Ci— Li, receives the signed from the transmitter described in the preceding article. The inductor con- sists of 40 turps 0.2 mm dia. enamelled copper wire close-wound on a 20 mm dia. ferrite rod. For optimum reception, Ci must be adjusted with a non-metal screwdriver. Note that the transmit fre- quency lies somewhere between 1,700 and 3,400 kHz. LEAD-ACID-BATTERY CHARGER Modern sealed lead-acid batteries are simplicity itself in use. In contrast to NiCd batteries, they may be charged by connecting them to a constant voltage (at the correct level). The charging cur- rent then gives a pretty good indication of the state of charge. These batteries may also be charged at a rapid rate, as long as the charging cur- rent is limited at the onset of the charg- ing process. Dependent on the make, a charging current of several times one tenth of the capacity in Ah is permiss- ible. For instance, a 5 Ah battery may be charged with an onset charging current of 1 A. The charging voltage may then be 2.45 V per cell. At such a (relatively) high voltage, the current has to be limited, otherwise the onset charging current through a flat battery may be as high as 10 A. The proposed charger, whose circuit is shown in Fig. 1, incorporates a 'stan- dard' voltage regulator, ICi, and a vari- able current limiter consisting of Ti, Ri, and R4. As soon as the current through Ri becomes too large, Ti switches on and the output voltage drops. The out- put voltage is given by: 1.2 (Pi+R 2+R3)/ R3 | volts |. The current limiter becomes operative The charging voltage for a 6-V battery than 3 V higher than the output voltage, that is required to be charged rapidly is The LM317 needs a heat sink, not 3x2.45=7.35 V. The total effective value because it is easily damaged, but of R2+P1 should then be 585 ohms. In because it cannot deliver its full output practice, this value may be slightly dif- current at high temperatures, ferent. It is, of course, possible to use the pro- For charging 12-V batteries, the value of posed circuit as a common supply unit. R2+P1 needs to be about 1290 ohms. Maxim Integrated Products have re- cently introduced a series of integrated step-up switching regulators designed for simple, minimum component count DC-DC converters. All control and stabilization functions are contained in an 8-pin DIP package: a bandgap volt- age reference, oscillator, voltage com- parator, catch diode, and an N-channel medium power MOSFET. In addition, the ICs have a built-in low-battery (LB) detection circuit. One of these new chips is the Type MAX641, which is of particular interest for no-break 5 V supplies in computers. In the application shown here, the out- put current of the step-up regulator is boosted by an external bipolar power transistor, Ti. The low-battery detector compares the voltage at input LB1 with the internal + 1.31 V bandgap reference. Output LBO goes low when the voltage at pin 1 drops below 1.31V. The low- battery threshold voltage, Ul b, is deter- mined by potential divider R1-R2 as £/lb=1.31(Ri/R2 + 1) (VI R2 is typically 100 kQ. In the application circuit shown here, LED Di at the LBO output lights when the input voltage drops below 2.62 V. It is possible to make the output voltage adjustable by connecting input Vre to a potential divider R3-R4 instead of ground. This option is shown inset in the circuit diagram. The output voltage, Uo, then becomes £/o=1.31(R3/R4+1) [V] R4 is, again, typically 100 kS. Cx is 100 pF. Remember to observe the voltage rating of C3. Maximum output current of the circuit is 1 A. The input voltage should remain below 5 V. Maximum conversion ef- ficiency is about 80%. As to components: the minimum value for Li, Lmin, is expressed by 9.100 alektor india September 1988 3V Li 884086-10 Lmm= t/ln/(2A/max) ing the input voltage or lowering the in- ductance. This causes the current to rise /max depends on the current rating of at a faster rate, and results in a higher the inductor and external power. transis- peak current at the end of each cycle, tor. Factor /o is the converter oscillation The available output power increases frequency, 45 kHz. The available output since it is proportional to the square of power can be increased by either rais- the inductor current. The calculation of the maximum inductance of Li is, unfor- tunately, relatively complex, and falls outside the scope of this introduction to the MAX641. The inductor should be able to handle the required peak cur- rents whilst having acceptable series re- sistance and core losses. The inductor in this application circuit should be rated at 2.5 A minimum. Due account should be taken of the rela- tively high ripple amplitude at the out- put of the converter. The ripple voltage is composed of high (45 kHz) and low- frequency components, and is practi- cally impossible to suppress further. Finally, D2 should be a fast Schottky di- ode. Alternatives to the type shown in the circuit diagram are the Types 1N5817 (1 A), 1N5821 (3 A), or the BYV27 (2 A). General purpose rectifiers from the lN400x series are not rec- ommended because their slow turn-on time results in excessive losses and poor efficiency. Source: Fixed Output 10 Watt CMOS Step-Up Switching Regulators. Maxim Integrated Products. | FISHING AID This circuit provides audible and visible 1 warning when a fish is nibbling the bait. Although this event is fairly easy to signal with electronic means, the circuit is relatively extensive to ensure that it can be powered from a 9 V battery. The circuit is based on a slotted opto- coupler Type CNY37, and a home made notched wheel. Unfortunately, the cur- rent amplification of slotted opto- couplers is very low (0.02 min.), requir- ing considerable current to be fed through the LED before a usable collec- tor current flows in the phototransistor. To avoid rapidly exhausting the battery, MMVi pulses the LED at about 250 Hz and a duty factor of 0.05. MMV2 detects the presence of these pulses. When a fish pulls at the bait, the notched wheel revolves in the slot, and intermittent pulse bursts are received at the trigger input of MMV2. Green LED Di lights, buzzer Bz sounds, and bistable N3-N4 is set, so that red LED D2 flashes at a 1.5 Hz rate. Di and -the buzzer are turned off when the fish gets off after nibbling the bait, but D2 continues to flash. The cir- cuit around Ni, T2 and Ci then serves to keep the current consumption as low as possible. The circuit can be reset by pressing Si. Preset Pi enables adjusting the fre- quency of the buzzer oscillator between 600 and 2500 Hz. When several fishing- rods are being used, each can be as- signed a particular signal tone. The buzzer can be switched off by means of S2. A suggested construction of the light barrier and the notched wheel is shown in Fig. 2. A small shaft is used in combi- nation with a reel around which the fishing line revolves. The slots cut into the detection wheel should not be too wide: 1 mm is a good starting value. The detection sensitivity is determined by the number of slots in combination with the reel diameter. The light barrier should be screened from daylight. In the stand-by condition, the circuit consumes no more than 4 mA, which goes mainly on account of the LED in the opto-coupler. In the actuated state, the current consumption rises to about 12 mA. 1 9.101 m Wideband RF signal tracer This simple and versatile circuit can aid in troubleshooting defective RF ampli- fier circuits. The usable frequency range of the tracer is about l6'0 kHz to 30 MHz. Measured signals (0.5 mV to 500 mV) are amplified, detected and made audible with the aid of a small loudspeaker. MOSFET Ti functions as an amplifier with a high input impedance to avoid loading the signal source. Transistors Tz, T3 and Ti form a high-gain logarithmic amplifier that drives AM demodulator T5-D5. A single chip AF power amplifier, ICi, is included to make detected signals audible. Testing of RF equipment is carried out simply by “probing around" at suitable lo- cations in the circuit and listening to the detected signal, whose relative ampli- tude can provide an indication of poss- ible sources of malfunction. The tracer's logarithmic amplifier obviates the need for frequent re-adjustment of the volume control, Pi. The unit is so sensitive that it produces audible output when the in- put is only held near the circuit section under test. As to construction of the tracer, this is best fitted in a short length of ABS tub- ing to make a probe with three connect- ing wires for the supply voltage and the loudspeaker. Constructors are advised to strive for ample RF decoupling and short connections in view of the rela- tively large bandwidth. Current con- sumption of the tracer is about 100 mA from a regulated 6 V supply. DRIVER FOR BIPOLAR STEPPER MOTORS For some applications, the Universal control for stepper motors (see |11 ) may be considered too extensive a circuit. Many small motors with limited speed range can be equally well controlled by a relatively simple circuit, based on, for instance, the Type SAA1027 or TEA1012 |2) . Most commercially available con- trollers are, however, intended for driv- ing unipolar stepper motors, which are now gradually superseded by bipolar types of similar size. In practice, the lat- ter can provide a larger torque, but re- quire a different type of controller. The recently introduced Type MC3479P from Motorola requires a minimum of external components for controlling a bipolar stepper motor. The maximum quiescent stator current, Is, depends on the value of resistor R between pin 6 and ground: Is=(Ub-0.7)/0.86R [mA] where R is given in kQ. The above rela- tion between Is and R is valid as long as the output transistors are not operated in the saturated area. The saturation point is reached sooner at low levels of the supply voltage, or when the ohmic resistance of the stator winding is fairly high. The manufacturers state a maxi- mum current of 350 mA per stator. The supply voltage for the motor (pin 16) depends on the ohmic resistance of the stator windings, and is allowed to vary between 7.2 and 16.5 V. When a high supply voltage is used, it must be remembered that the output transistors will not operate in the saturated area to prevent exceeding the set stator cur- rent, Is. The current control used here allows a fairly high step rate at the cost of an increase in the dissipation of the driver IC, particularly when the motor is held stationary. If necessary, the MC3479P can be cooled by connecting the 4 central ground terminals to a rela- tively large copper surface on the PCB. The integrated controller has 4 TTL and CMOS compatible inputs (see Fig. 1): CLK (pin 7): every rising edge of the clock signal causes the motor to revolve one full or one half step, depending on the level at pin 9. The maximum step rate and the minimum pulse width are 50 k Hz and 10 /is respectively. CW/CCW (pin 10): the logic level ap- plied here determines the motor’s direction of travel. F/H step (pin 9): this input allows selec- tion between full (0) or half step (1) operation — see Fig. 3. OI (pin 8): this output impedance selec- 3 tion input is only effective in the half step mode. It determines whether the stator winding is effectively discon- nected from the driver (0), or connected to the positive supply at both ends (1). The latter option improves the damping of the motor in the half step mode, and will prove useful at relatively low step Pin 11 of the driver IC is an open- collector output with a current capacity of 8 mA, activated during period A in Fig. 3. A LED connected to this output will flash rhytmically when the motor is running. Transistor Ti was added to obtain a reset function. No stator current flows, and the logic circuitry in the driver is reset, when the stand by input is driven low. When a logic 1 is applied, the motor is energized starting from state A. The addition of R2 makes it possible to switch the driver to the power-down state, rather than the reset state. The stator current is reduced to the value set with R2, as shown in the above formula. The motor driver is probably best con- trolled by a computer output port. The circuit in Fig. 2 is intended for stand- alone applications. It is composed of a supply, Rs-Da, an oscillator, N1-C3-R9-P2, and a re-triggerable monostable multivi- brator, N2-C2-R10-D2. When Si is opened, the oscillator is enabled, and the motor OUTPUT SEQ will start running. The clock frequency, is., the step rate, is adjustable with P2. The monostable will remain set via D2, and Ti will conduct, as long as clock pulses are applied to the motor driver. The amount of ever reversing stator cur- rent is limited by the stator inductance, but can still be increased with the aid of Pi. When the motor stops, Ti is turned off, and the stationary stator current is reduced to the value set with R2. The above arangement keeps the dissi- pation of the motor and the driver within reasonable limits. The current consumption of the com- plete circuit is practically that of the motor alone (700 mA max.). The motor driver IC consumes about 70 mA. NON-INTERLACED PICTURE FOR ELECTRON Owners of the Acorn Electron home computer may well object to its interlac- ed, and therefore slightly instable, pic- ture. There is a trace of display flicker in non-moving areas on the screen, and this is mainly due to the internal video procesing circuitry operating on the basis of interlacing, a technique used in conventional TV transmission for smoothing the appearance of moving picture areas. Arguably, interlacing is not very useful in computers, since these work with text in most appli- cations. Special displays with a rela- tively long afterglow time are no remedy for this awkward problem, and that is why the present circuit was de- signed. It effectively switches off the in- terlace function, and so ensures a restful display, albeit that the individual lines that make up the characters become slightly more prominent. Figure 1 shows that a TV picture is com- posed of 62S lines divided between 2 rasters of 312.S lines each. In an inter- laced picture, these rasters are vertical- ly shifted by one line. This is done by starting the second raster x and a half time later than the first raster. Inter- lacing can thus be rendered ineffective by starting the second raster half a line period earlier (ie., after 312 lines rather than 312.6). To retain the normal number of lines (626), the second raster is ar- ranged to comprise 313 lines. The ULA chip (Uncommitted Logic Ar- ray) in the Electron computer provides a horizontal and a composite synchroniz- ation signal, which are shown in Figs. 3a (HS) and 3b (CSYNC) respectively. With reference to Fig. 3c, and the circuit diagram in Fig. 2, MMVi forms a new vertical synchronization pulse, VS, with the aid of the CSYNC signal. The period of pulse VS is different for the first and second raster, so that MMV2 is needed to make VSYNC equally long in both. MMV2 is triggered on the first line pulse (HS) that occurs when VS is active, and is retriggered when VS goes low- see Fig. 3d. The length of the VSYNC pulse so made is about 160 tis, or about 2.6 times the line time (64 //s). The HS and the new VS signal are combined in XOR gate N2 for driving the video modulator. Gate Ni serves to buffer the HS output of the ULA. The final results obtained with the cir- cuit depend mainly on the type of TV 9 * 1 04 elektor india September 1988 Z first :== second raster == raster _ _ _ flyback = = = (blanked) set or display used, and may not be opti- mum when the TV is driven via its RF in- put. On an older type monochrome set, the central area of the picture was stable, but the upper and lower areas gave a less favourable look. Good results were obtained, however, from the use of Type TX chassis, which are currently the basis of TV sets sold under many different names and licenses. Even better performance can be ex- pected from a video monitor, whose (TTL compatible) H and V synchroniz- ation inputs can be driven by N< and N3 respectively. The polarity of the sync signals can be selected with the aid of wire jump ers. Co nnecti ons c a nd c’ result in VSYNC and HSYNC. The choice between jumper a or b depends on the type of display used. Preset Pi is adjusted until the picture appears ver- tically synchronized: the adjustment is fairly critical when jumper a is used. The final results obtained with the cir- cuit can be judged from looking at a few characters in the upper and lower area of the screen. The modest current con- sumption of the circuit, 10 mA, makes it possible to power it direct from the Electron computer. 13889.105 This is one of the very few ‘one-armed output state of the counters is not bandits’ to which the maxim the sole predictable because of the inconstant way to win is not to gamble is not ap- delay between the disable instants, plicable. In other words, this circuit NAND gates N13-N15 detect the winning does not have a slot for inserting coins: combinations, ie., LED D2 lights, and Bzi every play is free. is sounded, when 3 identical counter Actuation and release of the 'PLAY' but- outputs are activated. Note that diodes ton, Si, causes the circuit to become Ds-Ds form a 3-input OR gate, and that operative. Series regulator Ti is driven the buzzer also produces sound when into saturation by T2, which is con- the LEDs are Hashing, since the pulses trolled by N2-N7. The outputs of Na, Nil, at output Qz of IC3 enable the oscillator and N10 go high in succession, and intermittently. disable counters ICs, ICi and IC3, The play is ended when the voltage which are all clocked by oscillator N12- across C3 is high enough for gate N7 to Ns, and reset by the pulse at their Q3 change state. T2 is turned off, and Ti no output. The 3 LEDs driven by each of the longer powers the circuit. An on/off counters, therefore, lights cyclically, switch is not required for the fruit When a counter is disabled by the high machine, thanks to its very low current level at its CE input, one of the LEDs in consumption in the de-activated state, the 3 groups remains illuminated. The 1 9.106 2 Semiconductors: Di;D3;D4;Ds = 1N4148 D2:De;D7:Da= LED (red) Ds;Dl0;Di 1 = LED (yellow) Di2;Di3;Di4 = LED (green) ICt;IC2 = 40106 IC3;IC4;IC5 = 4017 IC6 = 4073 Ti=BC557 T2 = BC547 Si = momentary action push button. Bz = PB2720 buzzer (Cirkit stock no. 43-272011. PCB Type 87476 0 5 WIDEBAND LEVEL-INDEPENDENT TRIGGER PREAMPLIFIER This circuit eliminates the difficulty in re-adjusting the trigger level of an os- cilloscope or frequency meter any time the amplitude of the input signal changes. The block diagram shows that the trigger pulses are supplied by a fast comparator that compares the instan- taneous input signal amplitude with a reference voltage deduced from the dif- ference between the peak amplitude of the positive and negative half cycles of the rectified input signal. The circuit is fast enough to handle input signals with a frequency of up to 100 MHz, and has a sensitivity of 100 mV PP . With reference to the circuit diagram, the input signal is raised in a wideband preamplifier based around a UHF dual- gate MOSFET, Ti, fed by constant cur- rent source Ta. Presets Pi and P2 define the potential at the source of Ti, and hence form the fine and coarse offset compensation adjustments for the direct-coupled chain of opamps ICi- IC2-IC3. The signal rectifier and direct voltage amplifier are formed by D1-D2- R4-C7 and IC2. The relatively weak signal is raised further in direct-coupled opamps IC3 and IC4 for comparison with the amplified measuring signal in opamp ICs. Schmitt-trigger/inverter IC 6 cleans the trigger signal before it is ap- plied to the test instrument. The trigger sensitivity is set by potentiometer P4. Choke Li is wound as 4 turns of 0.2 mm dia. enamelled copper wire through a small ferrite bead. MOSFET Ti may be replaced by a Type BF991 or BF966 if either of these is easier to obtain locally. The circuit should be constructed with due attention paid to the relatively high frequencies it can handle. In this con- text, it is recommended to use a large copper area as an effective ground plane onto which the parts are fitted. The shortest possible connections, am- ple screening, and effective decoupling of the supply voltage at various point in the circuit are also a must to ensure cor- rect operation. Optimum sensitivity is achieved by ad- justing Pi, P2 and P3 for lowest offset measured at the output of IC3. These adjustments are carried out after a warming-up period of a few minutes. and with the input of the preamplifier temporarily short-circuited. FAST STARTING WIPER DELAY A wiper delay is essentially a bistable multivibrator whose off-time is adjust- able with a potentiometer. Many wiper delay circuits are based on the Type 555 timer in its standard application circuit, which has the disadvantage of introduc- ing a delay of about 1.6 times the set in- terval before the first wiper action takes place. This is especially annoying when an interval of, say, ten or more seconds has been set. This circuit is also 555 based, but is unique in that it arranges for the wipers to be activated immedi- ately at power-on. The circuit diagram of Fig. 1 shows the internal organization of the 555 timer to aid in clarifying the operation of the present circuit. When SI is closed, pin 6 is immediately pulled to +12 V because Ci is discharged as yet (see also Fig. 2b). The bistable in the 555 is reset, the output goes low, and Rei is energized. This forms the basic difference with the standard application of the 555, where Ci, connected as shown in Fig. 2a, delays the relay action until charged to % of the supply voltage. Returning to Fig. 1, Ci is charged via R2 and the 555's internal transistor when the output is ac- tivated. The bistable is reset when the voltage at pin 2 drops below ViVcc, causing the relay to be de-energized, and Ci to be discharged via R1-P1. The discharge time, and hence the wipe in- terval, is defined by the setting of Pi. When this is set to the shortest delay, the wiper motor is constantly powered via Rei, since Ci is not charged via P1-R2 only, but effectively via voltage divider P1-R1-R2 also. The wiper delay is fed from the 12 V car battery, and its current consumption is practically that of the relay alone. Note that the coil current may not exceed 200 mA. TEST-VOLTAGE SUPPLY For testing zener diodes, base-emitter breakdown, diacs, and so on, a fairly high voltage is needed. The usual type of laboratory power supply is not suitable, because its output is normally of the order of only about 30 V. If the re- quired current does not exceed 10 to 15 mA, it is possible to make a short- circuit-proof power supply with vari- able output voltage from 0 to 50 V from a handful of components as shown in the accompanying diagram. Circuit ICi amplifies a direct voltage set by P2 by a factor of about 6. Its output voltage should be about 25 V with respect to junction C1-C2. This voltage is inverted by IC2, whose output is thus -25 V. There is then available either a symmetrical ±25V potential with respect to junction C1-C2, or 50 V asym- metrical across the outputs of the ICs. The actual value of the voltage is set with Pi. The maximum current is limited by the ICs to about 20 mA, so that the likelihood of damage to a component under test is very small. The output is short-circuit-proof for an indefinite To avoid common-mode problems, and also to make it possible to vary the out- put voltage to 0, the supply voltages to 1C i and IC2 overlap to some extent, which is arranged by D6 and D7. Zener D6 also functions as the voltage refer- ence. The supply to ICi must be decoupled separately by a 100 n capaci- tor; that to IC2 is decoupled adequately by Cz and C3. The mains transformer may con- veniently (and inexpensively) consist of two 18 V types, otherwise a single 36 V unit is required. The secondary must be able to provide a current of 20 to 30 mA. If two transformers in series are used, make sure that they are in phase. Before inserting the ICs into their sockets, check the voltage at pins 4 and 7: this should be not higher than 36 V if a 741C is used, or 44 V for other types (741A, 741E, and 741). If the voltage is too high, a transformer with a lower rated secondary (2 x 15 V or 30 V) should be used. If, however, the voltage at pins 4 and 7 becomes lower than 27 V, it may be impossible to obtain an output voltage of 50 V. Motorist are generally well aware that car fuses do not blow just like that. None the less, when something appears to be amiss in the electrical circuit, a new fuse is nearly always fitted prior to in- vestigating the possible cause for the malfunction, which then, of course, costs two fuses. The circuits shown here are short circuit proof power switches, or electronic fuses with switch control dimensioned for relatively heavy (lamp) loads in a car. Both circuits are com- posed of a power switch, Ti, and a cur- rent limiter, T2. The circuit is fully short- circuit and overload resistant, provided Ti is adequately cooled, and the whole unit is constructed in a sturdy enclos- ure. The circuit in Fig. la has the lower voltage drop of the two, while that in Fig. lb is used when a TO-218 style Type MJE2955T or TIP2955 is not obtainable. It is interesting to note that the plastic TO-218 package is mechanically inter- changeable with the wellknown TO-3 outline, and enables ready mounting of the transistor onto a flat surface using an insulating washer— see Fig. 2. The use of a die-cast enclosure and TO-3 style transistors is illustrated in Fig. 3. This unit houses two power switches, one of which has its contacts at the rear side. Pay great attention to the correct elec- trical insulation between the transistors and the enclosure, and, if required, that between the enclosure and the car body. Switch Si is the existing control for the relevant lamp in or on the ve- hicle. Note the difference in respect of the connection of Si in Fig. la and lb. Table 1 shows how Ri and R ? are dimen- sioned in accordance with the current p«n list Resistors <±6%): Ri;Ri= see text T| = MJE2955T or TIP2955 (Fig. la) Ti = MJE3055T or TIP3055 (Fig. lb) Tj = BD136 or BD140 (Fig. la). Ti = BD135 or BD139 (Fig. 1b). Miscellaneous: Si = see text PCB Type 87467 requirement of the load, and also gives a suggested area of the cooling surface. Finally, when the printed circuit board is used, Tt should be a TIP29S5 or a MJE29S5T, not a MJE2955, since this has its outer terminals (B-E) reversed. MHz CLOCK GENERATOR Currently, 48 MHz quartz crystals are widely available at relatively low cost thanks to their use in computer systems. In these, there is often a need for several clock frequencies that can be derived from a central oscillator. When this sup- plies a buffered 48 MHz signal, it is rela- tively simple to add a divider circuit that provides lower, phase-synchronous, clock signals of, say, 6, 8, 12, 16 or 24 MHz. Obviously, this obviates the need for separate quartz crystals and as- sociated oscillators, and so economizes on hardware expenses. A reliable 48 MHz oscillator is fairly dif- ficult to make with HC or HCT gates. The oscillator shown here is, therefore, built around discrete RF transistors. It operates with inexpensive, third over- tone series resonance quartz crystals in the range between 44 MHz and about 52 MHz. A parallel L-C network may be 9.110 alelctor India septembeM988 connected in series with the crystal as quency to 48.000 MHz, but also for ‘pull- frequency multiplier (local oscillator shown in the circuit diagram for ac- ing’ the oscillator a few kilohertz around chains in 2 m, 70 cm, or 23 cm amateur curate setting of the oscillation fre- this frequency if it is used for driving a radio equipment). Digitally controlled attenuators almost invariably use some kind of tapped re- sistor network to simulate a poten- tiometer. This solution is fine as long as the number of steps required is small. When finer control is required, how- ever, the normal tapped resistor net- work is hardly ever used because of the large number of components that would be required. The circuit shown here of- fers relatively high resolution (attenu- ation range: 48 dB) whilst requiring few components only. The technique used is similar to that of multiplying DACs (digital-to-analogue converters). In a conventional R-2R lad- der DAC, the output voltage is given by (£7rof/384)W, , where N is the binary number applied to the inputs. The direct dependance of the output volt- age on Urei makes it easy to obtain a variable attenuator by substituting the input for Kef. The output will then be (Kn/384)At The R-2R ladder network used here is composed of resistors Ri to R17 incL, while electronic switches ESi to ESs incl. form the switching elements. These are of the two-way type (SPDT), connecting either the input voltage or ground to the inputs of the ladder net- work. Buffer ICi presents a constant im- pedance to the source. Pin 7 of ICi, IC2 and IC3 should be grounded unless the circuit is operated with bipolar signals. In that case, pin 7 of all three ICs is con- nected to - 8 V. The circuit can handle signals of up to 400 kHz with a maximum amplitude of about 4 Vpp. With signals of lower level, higher frequency response should be obtainable. The high frequency limit is due to the buffer at the input— the elec- tronic switches by themselves can handle signals up to 10 MHz. The fixed attenuation of the circuit is about - 3.5 dB. Signal-to-noise ratio is more than 100 dB at an input signal of 1 Vims. The output offset voltage is com- pensated by adjusting Pi. Current con- sumption of the circuit is about 6 mA at K>= +8 V. Finally, it should be noted that TTL circuits can not drive the circuit direct, unless 47 kQ pull-up resistors axe fitted at control inputs D0 to D7 incl. IC4=LF356 ES1...ES3 =IC1 ES4...ES6 = IC2 ES7, ES8 = IC3 R1...R10=22ls,1% R11...R17=11k,1% 4053 ELECTRONIC SAND-GLASS This electronic version of the reversible sand-glass uses a set of LEDs to simulate the passing of sand grains from the up- per to the lower bulb. The simple to build circuit is accurate enough for most domestic timing applications. The circuit diagram appears in Fig. 1. On power-up, shift registers IC3 and IC4 are reset by the low pulse from network Ras-Cr. A few seconds later, the sand- glass is started. The oscillator in IC2 generates a clock signal for the shift registers. The clock frequency is adjust- able with Pi. Switch S2 enables selecting one of the three timing periods stated in the circuit diagram. Si is a small mer- cury or ball changeover switch mounted inside the sand-glass. When this is reversed, the switch toggles and so selects the odd or even numbered LEDs. Assuming that Si is set as shown in the circuit diagram, every clock pulse causes a logic high level to be shifted into ICs , for as long as pin 13 of IC4 re- mains logic low. The MS bit of IC3 (out- put Q7) is shifted into the second shift register, IC4. Controlled by the shift register outputs, transistors Ti to Tie incl. switch off the odd numbered LEDs, and light the even numbered ones Ci = 100/i; 16 V; axial Cj;Ci;Ci;C» = 1 00n Ci = 2/i2; 16 V; radial Cr;C«= 1/r C. = 220n Semiconductors: Di . . . Du incl. = red LED D». . Dia incl. = 1N4148 TV . . Tir incl. -BC547 ICi = 4093 ICa = 4060 ICi;IC4 = 74HCT1 64 ICb«7805 B 21 = PB2720 (Toko; Cirkit stock no. 43-27201). Si = SPOT mercury, ball or tilt switch, e.a. 9.112, sequentially. When pin 13 of ICa goes high, counter IC2 is reset via N1-N2, while oscillator Na is started. Buzzer Bzi is actuated and sounds for about 2 seconds (Ca-Rai). The pitch of the tone can be set with P2 . When the sand-glass is reversed, Si toggles, ending the reset state of IC2. Logic low levels are shifted into IC3 because pin 13 of ICa is logic high. The even numbered LEDs go out one by one, and the odd numbered ones light, until pin 13 of ICa goes low again. IC2 is reset, Bzi produces a short beep, and the sand-glass can be reversed for a new timing period. LED D33 indicates that the sand-glass is operative. The cir- cuit is fed from a small mains adaptor capable of supplying about 200 mA at an output voltage between 7.5 and 12 VDC. Construction of the sand-glass is straight-forward using PCB Type 87406— see Fig. 2. The position of the LEDs on the front panel of the enclosure is shown in Fig. 3. Make sure that each LED is connected to the corresponding soldering island on the PCB. SPDT Switch Si is made from two SPST mer- cury or ball switches, fitted together but mutually reversed at a suitable position in the enclosure. The action of the switches is tested by reversing the sand- glass and measuring the switch con- figuration with the aid of a continuity tester or an ohm meter. All parts in the sand-glass enclosure should be fitted securely in view of the reversibility of the enclosure. The socket for connect- ing the adaptor, and rotary switch S2, are fitted in one of the side panels. A proto- type of the electronic sand-glass is shown in Fig. 4. The detachable front panel that holds the LEDs was cut from perspex sheet. ing the reset input logic low causes the motor to remain halted in the home pos- ition (LED Ds is quenched). Power driver Type L298 supports con- stant current drive of the stator win- dings. Current .drive gives good results because it allows stepper motors to be connected to a voltage that is higher than specified for voltage drive. Current drive considerably improves the motor's dynamic characteristics (start frequency and maximum step-rate). An internal oscillator sets a bistable at the start of each period, when the stator windings are connected to the supply voltage. Due to the stator inductance, output current will initially rise linearly, resulting in a linear voltage on current sensing resistors Ri and Ra. When the measured voltage reaches a certain user-defined peak value, Vrot, two inter- nal comparators reset the bistables, and the stator current is interrupted. Free- wheeling diodes then reduce the induc- ed stator field. From the above it is clear that current drive works by peak detec- tion. The resultant avarage current depends on Viet (adjustable with Pi), the oscillator frequency (adjustable with P2) and the values of the sensing resistors. Ripple amplitude on the stator current depends on stator self-inductance and the logic level at the MODE input. When this is high, the outputs of IC2 are switched to high impedance during the free-wheeling period. The stator field is reduced fairly rapidly via the free- wheeling diodes which conduct because the instantaneous voltage on the stator winding is slightly higher than the supply voltage. When MODE is held logic low, one transistor in the bridge circuit internal to the L298 remains on during the free-wheeling period. This causes the free-wheeling voltage on the stator winding to remain relatively low, resulting in slower reduction of the stator field strength and, therefore, re- duced ripple (phase chopping, see Fig. 3). This option is offered to enable ef- ficient current control of motors with a relatively low stator self-inductance. Synchronization of the oscillators in the L297s is required when multiple drivers and motors are used in a single system. This is simple to accomplish by fitting parts P2, R11 and Ci on one driver board only, and feeding the signal available at the SYNC output to the SYNC terminal on the other boards. An on-board divider, IC3, is provided to supply the clock signal when the rel- evant computer output line cannot be programmed to toggle at the required step-rate. The divider is clocked with the SYNC signal of the L297, and jumper block Ki allows selecting 1 of 7 available clock frequencies (step-rates). On-board clocking via IC3 can be disabled by driving input GATE logic low. The CLOCK input then functions as an output, enabling the computer to keep track of the number of steps per- formed. When external clock pulses are applied to the board, IC3 is simply omit- ted. The 5-40 V supply rail need not be regulated — smoothing is adequate here. The maximum attainable step-rate increase with supply voltage, but 40 V should not be exceeded. The chopper frequency (refer to Fig. 3), and hence the step-rate in stand-alone applications, is set with P2. Stator cur- rent is set with Pi. Lisping sounds pro- duced by the motor point to instability of the current drive. This effect can be remedied by either re-adjusting the chopper frequency, or by selecting the other logic level at the MODE input of ICi. When this still fails to stabilize the current drive, the supply voltage must be reduced until the motor operates with voltage instead of current drive. Stand-alone use of the driver is simple to accomplish by connnecting three exter- nal switches as shown in Fig. 5. Figure 6 shows how to connect the driver board to a unipolar motor. The oscillator inside ICi is used only for generating the clock signal required in stand-alone ap- plications of the driver. When it is used, the step-rate can be set by fitting a jumper in the appropriate position on Ki, and adjusting P2. Finally, IC2 is purposely located at the edge of the printed circuit board to en- able it to be bolted on a metal surface for cooling. .9.115 The Type TEAS114 from Thomson-CSF comprises three electronic switches fol- lowed by a buffer/amplifier. Normally the voltage amplification is '2 (6dB). When the input voltage exceeds 1.2 Vpp, or when the output voltage ex- ceeds 1.5 Vpp, an internal selector reduces the amplification to unity (0 dB). The threshold of 1.2 Vpp is created with the aid of voltage divider R4-R5, which also forms the input termination of 75 0. Series resistors Ri-Rs ensure 75 Q output impedance for driving video equip- ment via standard coax cable. The TEA5114 can be used as a video source selector also, provided each input has its own 75 Q termination network. The non-connected inputs should then be fitted with a coupling capacitor. Chan- nel selection is effected by controlling the logic level at pins 10, 12 and 15. Note that the logic 1 (high) level corresponds to +2.5 V here. DISCRETE +5 to -15 V CONVERTER This negative voltage converter differs from a host of other designs in not being set up around the latest integrated cir- cuit. The circuit diagram shows that only a handful of commonly available parts are required to build an efficient +5 to - 15 V converter. 1C i functions as a self-oscillating multi- vibrator that supplies an output signal with a relatively high duty factor. The LM311 is designed to operate from a single 5 V supply, and has a high output current capability for driving switching transistor Ti. Duty factor of the output signal is determined mainly by voltage divider R2-R3, and frequency of oscil- lation by C2-R4. Transistor T2 forms part of a regulation loop that modifies the os- cillator duty factor to maintain - 15 V at the output of the converter. The output voltage, Uo, is calculated from Uo= -(VD1+ Ub.E(T1)XR8/R9 + 1) [V] The component values shown give the following design data: Efficiency (Po/Pi): max. 75% Oscillator frequency: 6 kHz Duty factor: approx. 0.8 Output ripple voltage: 100 mV at Il= 200 mA Maximum load current: 200 mA Ti should be fitted with a small heat- sink. It happens from time to time that very large voltage spikes (lightning; switching of large loads) are superim- posed on the mains. Although these spikes are of very short duration, they may have disastrous consequences for mains-operated equipment. A mains power supply can be effectively pro- tected from such spikes with the aid of varistors. These components can handle, but only for a few microsec- onds, currents of thousands of amperes. In the proposed protection circuit, three varistors are used: one between L(ive) and N(eutral); one between L en E(arth); and one between N and E. The varistors are preceded by fuses, so that only the equipment connected via the circuit is protected. If these fuses were omitted, the entire household supply would be protected with the risk that one of the main fuses blows during an over-voltage. The circuit is best built into a small man- made-fibre enclosure with integral plug and socket. The mains-carrying bare wires should be kept separated by at least 3 mm. i9.117 FROM ALTIMETER TO VARIOMETER The altimeter published some 18 months ago (,) can be adapted to func- tion as a variometer by the following cir- cuit. The difficulty in the design of the circuit is, of course, that it has to work with very small input voltages. It is based on the fact that differentiating the absolute height gives as result the rate of change of altitude. In the diagram, ICi is the differentiator that operates with a time constant, R1C1, of 1 s. Since this type of differentiator in- verts, it is followed by an inverter. If the amplification is arranged at 60 (Pi set to 60 k), the eventual read-out shows the rate of change of altitude in m/min, assuming, of course, that the altimeter has been calibrated as prescribed in Ref. 1. Because of the very low levels of signal input, the choice of components is critical. For instance, Ci must be an MKT, not an electrolytic, type. The dif- ferentiator is a CMOS opamp that not only has a very high input impedance, but also extremely small drift of offset voltage with temperature. This drift is so small only if the opamp is used in the low-bias mode (pin 8 connected to +). This has the additional benefit of very low current (typically 10 //A). It also has a disadvantage in that the slew rate is only 0.04 V//s, but that does not matter here, since for all practical purposes the stage functions as a d.c. amplifier. Offset voltages are also undesirable in IC2, because they are added to those of ICi and appear amplified at the output. Therefore, P2 has been incorporated to compensate all offset voltages. Preferably, IC2 should also operate in B84084-10 the low-bias mode, but it may be found PCB, and should be well screened, necessary to connect pin 8 to pin 3 When the unit is used as variometer, the (medium bias) or even pin 4 (high bias) multiturn potentiometer (P7) in the to obtain full offset compensation. This altimeter must not be turned. If the unit has to be tried out in each and every in- is used as barometer, the switch should dividual unit. Simply adjust P2 for a dis- be set to the altimeter position, play reading of 000 when the unit is at Readers should note that the circuit has rest. been tested in laboratorium conditions The terminals should be connected to only and NOT in practical use. the corresponding ones in the altimeter. The switch at terminal F allows selection between altimeter and variometer use. The add-on circuit may conveniently be mounted above or under the altimeter BACKGROUND-NOISE SUPPRESSOR Hiss, crackling, and other discordant sounds are disconcerting and frequent sources of annoyance to most music lovers. Unfortunately, the sources of this background noise are not easy to eliminate, but the circuit proposed here will be of help. It should be ap- preciated, however, that the suppres- sion of noise is always a last resort: the best way of getting rid of it at source. The circuit is based on the fact that background noise is always at its most annoying during quiet music passages. It attenuates the output signal by some 45 dB when there is no or very low music signal input. When the input rises, the attenuation decreases propor- tionally, becoming 0 dB with normal to loud passages. The input signal is taken direct to the output terminals via R11 and R12 respect- ively. At the same time, they are summed via Ri and R2 and applied to non-inverting amplifier ICi via poten- tiometer Pi. The cross-over point in the gain characteristic of ICi is determined by Rs and Ci. Frequencies above the cross-over point are not amplified, and so do not contribute to the suppression. The output of ICi is rectified by Di— D2 and used to switch off Ti. This enables T2 and T3 to short-circuit the output and thus suppress the noise signal. When Ti begins to conduct, the base voltage of both T2 and T3 decreases and the output attenuation is reduced: noise signals are thus suppressed to a lesser The sensitivity of the circuit may be varied by Pi: the higher the sensitivity, the sooner the suppression lessens. This allows the sensitivity to be matched to different music sources. The peak signal level the circuit can handle is about 210 mV. Distortion at that level is not greater thaj. 0.01%. The delay before the circuit operates is determined by time constant R7C4. With values as shown, it is about 1 s but can, of course, be altered to individual taste. The circuit operates from a 12—30 V supply and draws a current of 2 to 3 mA. K 9.118 ifj DC. DETECTOR The d.c. component of a signal can only be detected by separating it from the a.c. component. This is most con- veniently done by filtering the a.c. com- ponent. In the proposed circuit this is effected with the aid of the common- mode rejection ratio (CMRR) of an opamp. (The CMRR is a measure of the ability of the opamp to produce a zero output for like inputs). The complete signal is applied to the in- verting input of opamp Ai, and only the a.c. component, via Ci, to the non- inverting input. The lowest frequency that can be detected is determined by time constant (R3+ROC1. With values as shown, a.c. suppression amounts to about 50 dB at 20 Hz. The output of Ai is fed to a low-pass filter to further attenuate high fre- quencies. This is necessary because the CMRR of an opamp decreases at higher frequencies. The difference signal is then applied to comparator A2. Diodes Di and D2 ensure that A2 reacts only to voltages greater than +300 mV. A negative direct voltage at the input of the circuit results in a positive potential at the inverting input of A2, which causes the relay to be deactuated (it is normally energized as long as the 12 V supply is on). A positive direct voltage at the input results in a negative poten- tial at the non-inverting input of A2, so that, again, the relay is deactuated. In normal operation, the voltage at the non-inverting input of A2 is arranged by potential divider R7-P1-R8-R9, so that the relay is energized. Because of Cs, the relay is energized a few seconds after the supply has been switched on. Capacitors C3 and C< serve to smooth low-frequency signals so as to prevent clattering of the relay. The relay is driven by a BC547B which can switch currents up to 100 mA. The supply to the relay should be not higher than 18 V. If the circuit is powered by a not entirely symmetrical supply, it may happen that the travel of Pi is insufficient: the value of R? should then be altered as re- quired. When the circuit is used in an active loudspeaker system, each output stage should have its own detector, consisting of Ai and associated components up to points A and B in the diagram. The out- puts of these detectors are then connec- ted in parallel to A and B. For mid- and high-frequency sections of the loudspeaker system the time con- stant of the input to Ai may be made smaller to obtain a faster reaction to d.c. components. Finally, the current drawn by the circuit is determined primarily by the relay. STEPPED VOLUME CONTROL The circuit consists of three distinct sec- tions. The first consists of a straightfor- ward amplifier, ICia and ICib. The sec- ond is a digital counter, IC3, which con- verts a binary code into a resistance value via ICz. That value is used to con- trol the degree of amplification. Finally, there is a pulse shaper, ICt, which enables IC3 to count up or down. Amplifier ICib has a switch-controlled gain of 0 dB or 24 da The control switch, S3, is an electronic type driven from output Qd of IC3. The gain of ICu can be set between 0 dB and 21 dB in steps of 3 dR The total gain of the two amplifiers can thus be set between 0 dB and 45 dB. The bandwidth of the amplifier extends from 10 Hz to 40 kHz. The peak value of the amplified signal should not rise above 8 V PP with a supply voltage of 5 V. The pulse shaper, formed by bistable N1-N2 and network C5-R16, indicates to IC3 whether it should count up or down. The RC networks suppress spurious pulses. The delay introduced by the RC net- works before N3 ensures that the clock pulse can not appear at the clock input of IC3 before the direction of counting has been set. The count position can, therefore, be increased or reduced by switches Si or S2 respectively. Inputs Ck and D/U of IC3 may be used to connect a software potentiometer: a two-wire connection per control is suf- ficient. An 8-bit user port can thus ac- commodate four of such digital poten- tiometers. A standard CD4051 must be used for IC2, because HC or HCT types do not allow the use of a negative supply voltage on pin 1. The other ICs may be HC or HCT types. If an LS type is used for ICs, 4k7 pull-up resistors are necessary at the outputs of this circuit to match the voltage levels of the two logic families. Note that C2, C6, and C7 are bipolar electrolytic types. The total current drawn by the circuit is about 10 mA. NOSTALGIC SINE WAVE GENERATOR As far as young engineers and techni- cians are concerned, a sine wave gener- ator is something you make from an XR2206. In the pre-IC era, sine wave generators were designed around discrete components. The generator described here has, however, more than just nostalgic value: it is also educational (and perhaps suitable for writers of the history of electronics). The (fixed) output frequency is fairly stable at 1 kHz, and the distortion, after proper adjustment, below 1%. The gen- erator is suitable for use as an audio test generator or as a morse code trainer and costs only a couple of pounds to make. The generator is of the so-called double-T type, which has the advantage of not needing any inductors. The oscil- lator proper, Ti, is followed by an emit- terfollower, T2, which ensures a suffi- ciently low output impedance. The frequency is set to 1 kHz by Pi, and P2 minimizes the distortion of the wave- form. With P2 set for minimum resist- ance, the amplitude of the output signal will be maximum, but the distortion will be quite appreciable. Increasing the re- sistance will reduce the distortion, but it may happen that when P2 is nearing its maximum value oscillations stop. Setting P2 is thus finding a compromise be- tween acceptable distortion and re- liable oscillations. The output level also depends on the setting of P2: it lies be- tween 1.8 Vp P and 3 V PP . The circuit may be powered by a 6 to 12 V supply: a PP3 battery (9 V) is perfect. Power consumption is about 48 mW. FOUR-CHANNEL STEREO SWITCH The circuit described here enables a choice to be made from four different stereo channels with only one switch. Internal switching is effected by CMOS devices to obviate crackling;, bounce, and other annoyances associated with mechanical switches. The two D-type bistables in IC2 are con- nected as binary dividers by linking their Q output to the D input. The 0 out- put of FFi is also linked to the clock in- put of FF2, which results in a kind of four-bit counter. The push-button is connected to the clock input of FFi. The four OR gates, N i to N4, decode the output states of the bistables, so that at all times only one gate has a high output. The outputs of the gates drive the CMOS switches in IC3 and IC4. The out- puts of the four electronic switches in these ICs are strapped together. The input of each switch incorporates a potential divider, ensuring that the switches operate in their linear regions. This arrangement ensures minimum dis- tortion of the audio signals: the negative parts of these signals would otherwise be distorted, since the switches work from an asymmetrical supply. The circuit draws a current of only about 1 mA at a supply voltage of 5 V. The supply voltage may be increased to about IS V. i9.121 PULSE RELAY An alternative to a two-way (or three- way) switch is the so-called pulse relay. This has the advantage that fewer wires are required and simpler switches may be used. Such a relay functions as a bistable: each input pulse changes its state. The relevant bistable in the proposed circuit is FFa, which functions as a J-K type. Every time the logic state at pin 13 changes from 0 to 1, the state at pin IS changes and causes the Ti-driven relay to be energized or deactuated. Bistable FFi serves as contact de- bouncer. When any one of switches Si to Sn is pressed, the reset input of FFi goes high, and so does the 0 output. After a short time (=time constant R3C3), FFi is set again. Since the bistable then has a set and a reset signal, the Q_output goes high and clocks FF2. The 0 output will go low again only when the relevant switch is released. The circuit is powered by a simple power supply that must provide not less than 14 to IS V and a current of at least 100 mA. Resistor Ri limits the supply to the bistables to 12 V. Since Ti is connected as an emitter follower, the operating voltage of the relay will stabilize at some 11 V. The BC107 then operates in its linear region and its dissipation will depend on the voltage across Ci and the current drawn by the relay. In some cases, it may be necessary to mount the transistor on a small heat sink. The relay will switch the mains voltages as required; a type should thus be chosen that can switch 240 V a.c. at about 1 A. The separation of coil and switch contacts should be at least 6 mm. Because the operating switches are completely isolated from the mains, light-duty, low-voltage types may be Readers should note that it is not al- lowed to place the connecting wires to the switches in the same conduit as mains-carrying cables. 0 6 I CRYSTAL FILTER FOR RTTY Excellent, small band-pass filters may be built with the aid of inexpensive 2,457.6 kHz crystals. A number of pro- totypes has proved the excellent reproducibility of these filters. The input and output impedance of the filter lies between 470 and 520 ohms. The 6-dB bandwidth is 150 Hz, and the bandwidth at -60 dB is 500 Hz. The filter is eminently suitable for use on CW, RTTY, and (AM) TOR. Since the insertion loss of the filter is only 3 dB, it is possible to cascade a number of them. The 6-dB bandwidth is then 120 Hz and the -60-dB bandwidth is about 240 Hz. Operating with RTTY, it is then attractive to work with 85 Hz shift. 9.122, 01 UP/DOWN CONTROL FOR DIGITAL POTENTIOMETER ub This circuit enables a digital poten- tiometer to be controlled manually with the aid of up/down buttons. The clock signal and count up/down selection for cascaded counters IC3-IC4 are supplied by R-C oscillators N1-N3 and the gate network at their outputs. The 8-bit control word at the output of the circuit is applied to the correspond- ing inputs of the analogue switches in the digital potentiometer. When all eight bits are used, the control circuit gives a resolution of l/2S5th part of the maximum attenuation of the digital po- tentiometer. Carry outputs (CO) of the counters are connected to an OR network D1-R7-N6 to prevent the digital word at the output jumping from 0 to 255 or from 255 to 0 when the lowest or highest volume set- ting is reached. When either IC4 or IC3 reaches its highest or lowest count, N6 inhibits further counting by blocking the clock pulses from Ns. Counting down is, of course, still possible from the highest count, 255, and counting up from the lowest count, 0. The volume is set to nought by pressing RESET button S3. Evidently, the circuit can have more ap- plications than that discussed here. It is, for instance, also suitable as a word gen- erator in computer systems. >9.123 jjjP EIGI EIGHT-BIT ANALOGUE I/O SYSTEM Analog Devices have in their catalogue a complete 8-bit analogue I/O system on a single chip, the Type AD7S69. This IC comprises an analogue-to-digital con- verter with 2 /is conversion, time; a digital-to-analogue converter with 1 /is conversion time; a reference voltage; and a bus interface for direct coupling to a microprocessor system. With an asymmetrical supply voltage of 8 V, the input and output voltage range from 0 to 1.25 V ot 0 to 2.5 V respectively (depen- dent on the logic level at the RANGE in- put; widest range available when RANGE = 1). When a symmetrical supply is used, voltages of + 1.25 V or + 2.5 V respectively can be processed. All that needs to be added is an address decoder as shown in the diagram. Here, the AD7569 is connected to output port 0 of a Z80 microprocessor. Gates Ni to Ni decode I/O address 0 to I/O at a read or write instruction. When this happens, the output of N4 goes low and IC3 is selected. At a write instruction, the data are read into the data bus and converted to an analogue output voltage. At a read instruction, the con- version is started and the pro cessor placed in the wait state via the BUSY output of the AD7569. When the BUSY pin goes high again, the data can be read and stored by the processor. A simple example in MSX BASIC; 10 OUT 0; INP (0):GOTO 10 This program immediately retransmits (via the DAC) the signal that is being written via the ADC. This shows how easy it is to work with the IC. The AD7569 is manufactured in CMOS and it therefore draws a current of only 12 mA. TRAIN DETECTOR In model railways, reed switches are in- Either way, the complete circuit is small is triggered by the current through R4. variably used for the detection of ap- and simple enough to be built in quan- The engine current is measured by the proaching trains. These glass tubes do tity so that all sections of the track can p.d. across the ai and a2 pins of the not look very natural and the corre- be supplied with one. triac. When the input goes high, T3 is sponding magnets are also relatively The economy version consists of the ex- switched on and the gate of the triac is large, particularly when N- or Z-scale treme lefthand part of the diagram. In- at ground potential. The triac will then trains are used. The detection system dependent of its polarity, the engine switch off. If a locomotive is present in proposed here offers a much more current flows through D3 or D4 and the relevant section of the track, it wil elegant solution. Trains are detected by causes a p.d. of about 1 V across the di- stop. It is interesting that the presence ascertaining from which part of the ode. Either Ti will then be switched on of this stationary engine can also be railway track current is drawn. via Ri, or T2 via R2. This results in the detected, because either Ti or T2 will The circuit is suitable for a.c. and d.c. output going logic low and D2 lighting, get a base current via the engine, systems, as well as for digitally- Diode Di serves to prevent the output The circuit needs an auxiliary supply for controlled model railways. It can be voltage dropping below 0 V when T2 the logic section. If this is chosen at 5 V, built in two versions. The first one is a conducts. the signals generated are TTL and simple design with LED indicators and a The more sophisticated version is CMOS compatible and may, therefore, digital output. The second offers the ad- shown at the right-hand side of the be processed by a computer, ditional facility of powering a given sec- diagram. In this, the two 1N4001S are ra- tion of track via a digital control input, placed by a triac. Normally, this device 9.124, An excellent instrument amplifier with differential input may be built from a single, inexpensive Type NES514 quad- opamp. The circuit shown is a develop- ment of the well-known arrangement in the data books of, among others, PMI, Burr Brown, and Analog Devices. The input stages, Ai and hi, amplify the difference signal from inputs U2 and Ui, while the comtnon-mode signal, Ucm, is not amplified. If all components are as- sumed ideal, the output voltages of the input amplifiers are Ua = U i (1 + 2R2/R1) + Ucm Ub=U 2 ( 1 + 2 R 2 /Ri')+Ucm The difference voltage is then Uo = U a— Ub =(Ui— U2XI + 2R2/R1) This voltage is amplified x 1 in A3. If a symmetrical output is required, inverter hi should be added. Unfortunately, this symmetrical output is not of good qual- ity at higher frequencies, since Ai in- troduces appreciable phase shifts. To obtain good common-mode suppres- sion, it is essential that R2, R21 and R3 to R8 are 0.1% types. It is also possible to use a small preset potentiometer be- tween A, B, and C (as shown inset) with which the common-mode suppression can be optimized. The amplification may be controlled within certain limits by potentiometer Pi. The supply voltage should not exceed 16 V at which the current drawn amounts to about 6 mA. s9.1 25 MANUAL SLIDE FADER Readers who have built the computer- controlled slide fader published a few months ago* 1 ', may add a manual fader at the cost of a few extra components. It is assumed that the projector has been fitted with a new DIN chassis-mounting socket. If a 7- or 8-way socket is used, there are at least two spare pins for con- necting the manual fader. It should be noted that a 7- or 8-way socket accepts a normal S-pin plug. Potentiometers Ps and P6 are connected to the free pins of the DIN sockets as shown in the diagram. These poten- tiometers control the brightness of the lamps in the projectors. The control voltage should be 2.S to S V, and this may be derived from the dimmer PCB as shown in the diagram, lb do that, two further preset potentiometers are re- quired in each projector. The additional presets are connected across Ci in the dimmer circuit, which has a stabilized potential of 12 V. Preset Pi (Ps) is ad- justed so that the projector lamp is at full brightness with Ps (P6) fully clockwise. Presets Ps and Pi are then adjusted so that the projector lamp is just not out with Ps (P6) fully anticlockwise. These adjustments should be repeated a couple of times as the presets affect one another. The photograph shows how the circuit may be built in a small enclosure. The projector selector knobs are located at that side of the potentiometers the slider points to when the associated projector is dimmed. With this arrangement it is seen at a glance which projector is in use. 9.126 LOGARITHMIC READ-OUT When the well-known Type 7106 voltmeter 1C is connected as shown in the diagram, the display shows the logarithmic ratio of the steady input voltages, Ui and U2 (where Ui £ U2). Ex- pressed as a formula: read-out = log(Ua/U 1). The value of Ui may be 20 mV to 2 V, while that of U2 must lie between Ui and Ui/100. For accurate operation of the circuit, the ratio Ri:Ra must be exactly 1:9. The circuit is set up by applying a direct voltage of IV to input Ui and one of 100 mV to input U2, and setting Pi to a position whereby the display reads exactly 1.000 (=logl/0.1). The circuit draws a current of a few millamperes at 9 V (a PP3 battery is perfect). (Maxim Application) Applications, of the well-known Type lighting. As soon as the supply voltage sated by the reservoir capacitors m the 565 timer have still not been exhausted drops below a certain value (set by Pi), equipment being monitored. If it is as shown in this circuit. It is based on the 555 is triggered and pin 3 goes high, desired to detect them as well, C2 must the voltage monitor published some After about 7 s, Da lights to indicate the be omitted. years ago (1) . The difference between malfunction. At the same time, relay Re 1 The alarm delay may be altered to m- the two is, however, that the present one is energized, and this switches on dividual taste by giving a different value measures its own supply voltage and ac- buzzer Bzi. because of the high value of to Ra or C3 (mono time = I.IR2C3). tuates a buzzer which continues to the electrolytic capacitor across the sound after a complete supply failure, buzzer, this will continue to sound for Dr J. Devasundaram and The circuit may be added to an existing about 30 s after the supply has failed. Dr Cariappa Annaiah equipment to detect if and when a Brief supply variations are not indicated power failure occurs. because of the high-value electrolytic The 555 functions as a monostable, capacitor between the wiper of Pi and When the supply is normal, pin 3 of the ground. Such brief variations in the timer is 0 and this is indicated by Di supply voltage are normally compen- .9.127 VOLTAGE MONITOR The MAX690 from Maxim is a monitor IC for computer PSUs that offers the fol- lowing facilities. • Resetting of the processor system by switching the supply voltage on and off. • Switch-over to back-up battery (for RAM, ROM, or other logic circuits) at mains failure. • Generating a reset pulse if the on- board timer does not receive a pulse for more than 1.6 s. 884095 • Giving a warning of low supply or battery voltage. The diagram shows a typical application of the MAX690. The supply voltage is connected to the + terminal (pin 2) and then supplied to the CMOS RAM in the microprocessor via output pin 1. The back-up battery is connected to pin 8. The IC can switch a current of maximum 100 mA. The R(eset) output of the IC is connec- ted direct to the R input of the micropro- cessor. The Power Fail Output (PFO) of the IC is connected to the N MI in put of the microprocessor. The PFO may serve to forecast a mains failure since, if the value of Ri is correct, this output goes low a few milliseconds before the supply voltage begins to decay. A reset is given if the supply voltage drops below 4.65 V. The Watch Dog Input— WDI— may be connected to an I/O line of the microprocessor. This in- put must get a leading or trailing edge at least once every 1.6 s, otherwise the R output of the IC is activated. If this function is not required, pin 6 is left un- connected. The Power Fail Input (PFI) is connected to the junction of potential divider R1-R2 across the unstabilized p ower supply. With values as shown, the PFO is ac- tivated when the unstabilized supply voltage drops below 8.25 V. If a different level is required, the value of Ri may be calculated from Ri = Rz(U— 1.25)/1.25 ohms where U is the required level of unstabilized supply voltage. The IC draws a current of 4 to 10 mA, de- pending on the output current. When it operates from the back-up battery, it draws a mere 1 //A. (Maxim Application) VERSATILE CONTINUITY TESTER This simple continuity tester has 4 resist- ance ranges for quick and reliable faultfinding in electronic equipment. Used with care, the instrument also allows testing diodes, LEDs and electro- lytic capacitors. The four resistance ranges indicated by LEDs are: ■ VLO = very low resistance = green LED. Resistance between test clips is smaller than 5 Q. Buzzer sounds. ■ LO = low resistance = yellow LED. Resistance between test clips is be- tween 5 Q and 100 kQ. ■ HO = high resistance = orange LED. Resistance between test clips is be- tween 100 kQ and 15 MQ. ■ VHO = very high resistance = red LED. Resistance between test clips is higher than 15 MQ. The continuity tester can be used for an initial check on the following compo- Diodes: conductive direction: yellow LED; non-conductive direction: red LED The test current is high enough to enable testing LEDs also. Capacitors: depending on capacitance, the yellow LED will flash briefly, fol- lowed by the red LED lighting continu- ously. Electrolytic capacitors: first, the yellow LED lights briefly, then the orange one. The red LED lights when the capacitor is fully charged. With some skill and ex- perience, the capacitance can be deduced from the charge time. The buzzer produces a continuous or a brief sound when the electrolytic capacitor is faulty. The circuit diagram shows that three op- erational amplifiers compare the drop across the test leads to a fixed voltage, and indicate which of the two is highest by switching their outputs to the positive supply level or ground — see the accompanying Table. The fourth opamp, A4, functions as a rectangular- wave generator for driving the buzzer. The generator is switched on by D7, because it is only allowed to operate when the output of A 1 is low and Di lights (VLO). After completion of the continuity tester on the PCB shown, ranges VLO and LO are adjusted with Pi and P2. Clip the test leads to a 5 Q resistor, and adjust Pi so that Di just goes out, and D2 just lights. Similarly, use a 100 kQ resistor for adjusting P2 until D2 and D3 just go out and light respectively. Current consumption of the tester is less than 20 mA from a 9 V PP3 battery, which should last for 10 to 15 hours of operation. The tester can, of course, also be powered from a mains adaptor. It is recommended to decouple Re with a 22 jjF electrolytic capacitor when the supply voltage is relatively low. To boost the sound output of the buzzer, Ri6 can be replaced with a preset — adjust this until the buzzer resonates. >9.129 0 7 fl COMPUTER- OR SENSOR- M?. CONTROLLED DIMMER The central dimmer chip in the circuit, an LS7331 or LS7332, accepts control commands from a sensor (number 1 or 2) and from a computer (input A). The load is powered when the sensor is briefly touched (between 39 and 339 ms). Depending on the type of con- troller used, applied power is then maximum (LS7331), or equal to the last set value (LS7332). The next touch on the sensor switches off the load. When the sensor is touched longer than 339 ms, ICi slowly varies the lamp intensity be- tween maximum and minimum. Lamp intensity is selected — and retained — simply by releasing the sensor in the ap- propriate instant. All control operations are synchronised to the mains fre- quency by a phase-locked loop in the controller chip. Inputs 1 and 2 are functionally equiva- lent, but the use of input 2 is preferred over input 1 when the sensor is connec- ted to the circuit by means of a cable. Input A makes it possible to control all functions by means of a computer. The dimmer chip sends it status to the com- puter via outputs B, C and D (mains failure; minimum phase angle; dimmer active). Operation of the circuit is relatively simple. Components D1-D2-R1-R2-C2-C3 form a 15 VDC supply for ICi. Capacitor C7 ensures that this supply continues to operate when relatively light loads (< 25 W) are controlled at large phase angles. In the interest of safety, series- connected resistors Ris-Ri6 and R?-Rs may not be replaced with single 4M7 and 10M types respectively. The value of Rs, Rio and R11 may have to be changed when other optocouplers than the 4N25 are used. To prevent the supply voltage for ICi dropping below 15 V, these resistors must not be made smaller than 680R. The total current con- sumption of the LEDs in the optocoup- lers should remain below 25 mA. The 5 V supply for the associated transistors is provided by the computer via ter- minals +5 V and 0. Finally, an important note on safety: the necessary insulation between the user and the mains can only be ensured if the dimmer is fitted in a sealed ABS enclos- 0 8 SYNCHRONISATION SEPARATOR The Type LM1881 from National Semi- conductor is a synchronisation separator that has already found its way in numerous commercial applications. Practical use of the chip is straightfor- ward, because the number of external components is kept to a minimum. The R-C network at pin 6 controls the the in- ternal timing of the chip, and the width of the synchronisation pulses at the out- puts. In the sample circuit, the R-C con- stant is dimensioned for a TV line fre- quency of 15,625 Hz. The input of the cir- cuit can be driven with composite video levels between 0.5 Vpp and 2 Vpp. Gates N 2 and N4 are provided to enable driv- ing monitor inputs that require inverted sync signals. The amplitude of these is determined by the supply voltage — when 5 V is used, TTL-compatible in- puts can be driven direct. 0 8 I I BURN-IN PROTECTION 1 FOR PC SCREENS Many computer users are in the habit of abandoning their machine for several hours without turning it off, or at least re- ducing the display intensity. This negligence readily leads to text or im- ages permanently burned into the phosphor layer of the screen. There ex- ist memory-resident programs that detect the prolonged absence of keyboard actions, but compatibility with the main program is often a problem. The hardware solution presented here is reliable, and should work with most IBM PC-XTs and compatibles. The circuit effectively sits between the keyboard and the computer, and be- tween the computer and the monitor. A CMOS switch breaks the connection between the computer and the monitor when no keyboard action is detected for some time. To prevent having to cut existing cables, the circuit is mounted in a small box fitted with the appropriate sockets and connectors. The supply voltage is obtained from the keyboard. Counter ICi is reset by data from the keyboard. When the data flow stops, the count output of ICi goes high after a predetermined period. This causes IC 2 to switch to high-impedance, so that keyboard data is blocked by Na. When any key is pressed, ICi is reset, and the connection between video output and *r 19889.131 monitor is restored. The first keyboard code is not transferred to the computer, because Na blocks data as long as Ci is not discharged. This arrangement en- sures the suppression, on the monitor, of the first, arbitrary, character typed on the keyboard to restore the video con- nection. The screen saver was tested on an Amstrad PC1640SD and a monochrome monitor (Hercules-compatible video mode). For computers with a GGA, it is probably necessary to break the inten- sity, rather than the video, signal. With PC compatibles other than the Amstrad, the logic levels on the keyboard data line are inverted. To be able to use the circuit with these machines, insert in- verter Ni between the output of N3 and the keyboard input of the computer. The ‘display-off delay can be selected as follows: Oil (pin 1) Q 10 (pin 15) 09 (pin 14) 08 (pin 12) 07 6>in 13) PC-XT: pinning of keyboard connector Assignment secondary green or intensity (I) secondary blue or mono video (V) £ DISCRETE VOLTAGE REGULATOR Low-drop voltage regulators can aid in saving energy by keeping dissipation in power supplies low. Unfortunately, inte- grated low-drop regulators with an out- put current capability of more than about 0.4 A are hardly found on the market. The low-drop series regulator shown here is intended for all appli- cations where more than 0.4 A is re- quired, and supply dissipation is to be kept as low as possible. Constant-current source T2-D1-D2-R6 en- sures high amplification and adequate suppression of hum and noise on the raw input voltage. T3 and T4 form a kind of darlington transistor. This is driven by Ti, whose base and emitter terminals are connected to the output voltage. When this rises, the emitter potential rises above that of the base. The transis- tor blocks, so that the control voltage for T3-T4, and with it the output voltage, is reduced. Diodes D3 to D6 serve to supply the start voltage for this regulator circuit. The output voltage is determined by D7 and R2/R3. Resistor R2 can be replaced with a 5K0 preset to compensate the (usually fairly large) tolerance on the zener diode. It should be remembered that the circuit has no current limiter, so that it is not short-circuit resistant. D1...D6 = 1N4148 884055 This circuit gives a loud warning when even the tiniest amount of water is detected on a special humidity sensor. When this is installed in a- suitable lo- cation, the circuit provides an early war- ning before a defective pump, leaking drainage system, water supply, washing machine or dishwasher can flood the bathroom, cellar or kitchen floor. There exist self-locking valves and automatic switch-off systems to prevent flooding, but these are, in the main, not sensitive enough to afford the degree of protec- tion required, i.e., they are not actuated until the domestic calamity is actually taking place. The circuit is an application of the low- power comparator Type LM1801 from National Semiconductor. The reference voltage for this IC is set with Rz. When the voltage at pin 4 of the chip exceeds the set threshold due to the sensor becoming wet or humid, the chip drives the active piezoelectric buzzer with a current of no less than 24 mA. Stand-by current consumption of the cir- cuit is about 10 A, so that a 9 V PP3 bat- tery should last for about 1 year. Finally, it ■ is, of course, possible to connect multiple sensors in parallel. SOFT-START FOR HALOGEN LAMPS This simple circuit extends the life of halogen lamps in slide or film projectors by eliminating the sudden temperature increase in the lamp filament when this is still cold, and forms a very low resist- Capacitor C2 is discharged yet when the lamp is switched on. This means that Ti conducts, Tz blocks, and the optocoupler-triac is not activated. The initial filament-heating current is limited to a safe 4 A or so by resistor Rs, which shunts the triac. Meanwhile, C2 is charged, so that the base voltage of Ti drops. This transistor is turned off, so that T2 starts to conduct. The crux of the circuit is that the LED in the opto-triac causes the triac to be fired only during the zero-crossings of the alternating supply voltage. Shunt resistor Rs is then effectively short-circuited, and the lamp lights at full intensity. The opto-triac should be fitted on to a large heat-sink. Maximum output cur- rent of the circuit is about 8 A. i9.133 m ELECTRONIC SIGNAL DIVIDER The problem is well-known: the more signal sources, the higher and more tangled the heap of interconnecting wires, and the more hum on recorders. This circuit can aid in avoiding these dif- ficulties and the awkward situations that arise from them. The modular structure of the signal divider allows it to be laid out to individual requirement (the cir- cuit diagram shown is but a suggested configuration). The signal divider shown should be capable of mastering relatively complex equipment set-ups thanks to its two tape recorder inputs, echo input, and auxili- ary input. In the sample circuit, six switches are used for routeing the signals. The function of these switches is summarised in Table 1. The symmetrical 8 V power supply of the circuit is a conventional design that merits no further discussion. The inter- nal structure of ‘black boxes’ ESi to ES« is shown in the top right-hand comer of the diagram. Each channel comprises two pairs of electronic switches which are controlled in complementary fashion by the logic level applied to in- put S. When this is driven positive, the horizontal (series) switches are closed, and the vertical (shunt) switches open- ed. This situation is reversed when input S is made negative (Sx closed). Pro- totypes of the switching units achieved a cross-talk level of -85 dB and a channel separation of more than 75 dB. Signal-to- noise ratio was more than 100 dB, and distortion less than 0.01%. The left and right channel of the auxili- ary input are joined with R7 and Rs. The monaural signal so obtained is then amplified in As and fed to the echo send socket. Amplification of the opamps is defined by the feedback re- sistor (Rs to R13). Th low current consumption of the signal divider (20 mA), makes it possible to replace the mains supply with a pair of 9 V batteries if the unit is not used fre- quently. Switch Function St Input to tape 1 (record) 52 Input to tape 2 (record! 53 Tape 1 (playback) to tape 2 (record) 54 Echo (input) to tape 2 (record) Ss Tape 2 (playback) to tape 1 (record) S6 Echo (input) to tape 1 (record) 9.134, 0 | ALTERNATIVE VOLUME CONTROL Tracking of many types of inexpensive logarithmic stereo (dual-axis) poten- tiometer is usually very poor, and gives rise to audible volume differences in stereo AF amplifiers. This article presents two alternatives for making a stereo logarithmic volume control with adequate tracking characteristics. To begin with, linear stereo poten- tiometers may be used. Plotting resist- ance as a function of spindle position, the single elements in linear stereo potentiomemeters give a straight line, marked 1 in the accompanying graph. Curve 2 is obtained when the indidual potentiometers are connected in series, and curve 3 when they are connected as shown in Fig. la. The latter two curves correspond to a pleasant and natural sounding volume control: with the po- tentiometer set to the centre of its travel, the attenuation is about 5, against about 10 for a standard stereo logarithmic po- tentiometer. The response of the volume control is also improved by the use of a stereo logarithmic potentiometer connected as shown in Fig. lb. This configuration enables an additional attenuation of about 6 dB to be achieved in the first part of the control range. This attenu- ation decreases gradually as the spindle is advanced. 884034-2 LOW-VOLTAGE CONTINUITY TESTER The size of this continuity tester can be kept very small thanks to the use of a 1.5 V penlight battery. The miniature loudspeaker sounds when the resist- ance between the test clips (or probes) is between 0 and 100 Q. Differences of 5 Q are translated in a corresponding change of the output volume. The battery has a relatively long life ex- pectancy because current consumption of the continuity tester is only 30 mfl when the test inputs are short-circuited. I -1 1V5 er 1988 9. 135 SYMMETRICAL VOLTAGE DOUBLER Many circuits based on operational amplifiers and comparators require an auxiliary, low-power, symmetrical power supply. The circuit shown here is an asymmetrical-to-symmetrical step-up converter, which is ideal for use in digital equipment, whether battery- powered or with a large 5 V supply, but lacking symmetrical rails for feeding opamps. Cost and space requirement of the converter should compare favourably with an add-on symmetrical supply requiring a separate trans- former, rectifier, smoothing and regu- lation parts. The circuit diagram demonstrates the simplicity of the voltage doubler, which is an application of the recently in- troduced Type MAX680 from Maxim In- tegrated Products Inc. Output impedance of each symmetrical rail is about 200 Q, and maximum cur- rent about 10 mA. In the quiescent con- 884058-10 dition, the chip consumes hardly any current at all, and ripple on the output lines is then only 40 mV PP . AUTOMATIC VOLUME LIMITER This circuit limits the output power of an which thus allows setting the onset point lights to indicate that the onset level of AF amplifier when predefined levels of of the limiter as a function of output the limiter has been reached. When Ds amplitude and frequency of the output signal amplitude and frequency. The lights at maximum brightness, it is im- signal, or current consumption, are ex- Table shows the configuration of the possible to use the relevant control on ceeded. filter for three ranges of amplifier output the amplifier for turning up the volume Current consumption of the amplifier is power. Capacitor Ci is a bipolar type, further. measured with Rx. Transistor Ti con- The filter is laid out to respond to low The LED and LDR are fitted in a small, ducts and drives the volume limiter, T2, and high frequencies to prevent the light-resistant, enclosure, e.g. a film con- when the drop across Rx exceeds 0.56 woofer and tweeter in the loudspeaker tainer. A single LED (Ds) enables an at- V. The limiter reduces the amplitude of box being overloaded. tenuation of 15 dB to be achieved. This the AF input signal by shunting this with The emitter of T2 is held at a reference can be increased to 20 or 23 dB by fit- a variable resistance formed by a light- potential that follows the supply voltage ting one or two LEDs in series with D2. dependent resistor (LDR), which is il- of the amplifier. Supply voltage fluctua- Distortion of the input signal is low, and luminated by LED D2. tions thus have a direct effect on the feedback problems that often arise in The T-filter connected to the loud- bias setting of T2. This does not con- FET-based circuits should not be en- speaker output of the amplifier can also duct until the threshold of 0.6 V between countered. Series resistor Ri is dimen- actuate the volume limiter via preset Pi, base and emitter is exceeded. LED Ds sioned in accordance with the required 9.136 elektoiindia September 1988 response of the limiter, and the signal level and impedance at the input of the amplifier. It should be noted that the proposed limiter is relatively slow. This disadvan- tage is caused by its response delay of about 100 ms. The circuit can be simplified to limit voltage peaks only by omitting the filter section, and applying the loudspeaker signal direct to Pi. Current consumption of the limiter with Da and D$ on is about 35 mA at a supply of 40 V. NOISE-RESISTANT 5 V POWER SUPPLY The recently introduced integrated voltage regulator Type TEA7034 from CSF-Thomson has been specifically de- signed for use in microprocessor-based circuits whose operation can be ex- pected to suffer from noise, spikes and digital interference. After power-on, the regulator supplies a delayed RESET pulse to the mi- croprocessor. The timing of this pulse depends on the values of Ri and Cz. In the application shown, the delay is about 0.6 s. The regulator is capable of withstanding input voltage peaks of up to 80 V. When the output voltage drops below 4.75 V, shunt regulator Ti is switched on, and Cz acts as a current buffer that temporarily feeds the load. Once the regulator is powered-up, the raw input voltage is allowed to drop to 6 V without affecting the stability of the output voltage. It shoul d be noted, how- ever, that the RESET output does not toggle properly unless th e input voltage is greater than about 8 V. RESET remains low at lower input voltages, with R3 functioning as a pull-up resistor. The circuit introduces a voltage drop of .9.137 £ PF PRESELECTOR FOR SW RECEIVERS The low input capacitance of modem dual-gate MOSFETs makes it possible to realize negative feedback by means of a non-decoupled source resistor. If prop- erly applied, this technique enables making an RF input stage with a rela- tively high dynamic range. No short- wave radio amateur needs to be told that good large signal handling capabilities are a must these days to prevent re- ceiver overloading and cross- interference caused by very strong signals. Unlike input sections in many top-class SW receivers, this circuit has no prob- lems handling RF input signals of up to 2.5 Vpp (not uncommon during night- time reception, and when a good aerial is used). The output then supplies 3 V PP when terminated in 50 Q. Tuning capacitor Ci determines the overall gain, which is mostly due to resonance in the L-C network at the in- put of the circuit. The maximum drain current that may be set by Pi is 12.7 mA, corresponding to 2.29 V on Rs. The minimum drain current is 10 mA (£/rs=1.8 V). The six input inductors are wound on high-quality ceramic formers with a diameter of at least 10 mm. The ferrite bead is slid direct onto the gate 1 ter- minal of Ti to prevent parasitic oscil- lation in the VHF or UHF band. Output inductor L7 is wound as 20 turns (A) and 4 turns (B) on a Type G.2-3/FT16 ferrite ring core. put selector are: 1:30... 100 kHz 2: 100... 300 kHz 3: 300... 900 kHz 4: 900. . .2700 kHz 5: 2700... 9000 kHz 6: 9000... 30 000 kHz SINGLE-CHIP RS232 TRANSCEIVER There are many applications in which only one RS232-C line is perfectly ad- equate for serial communication be- tween computer equipment. In these cases, it is often tempting to resort to the use of the well-known integrated cir- cuits Type 1488 and 1489, which are a quadruple RS232-C line driver and re- ceiver, respectively (the equivalent types from Texas Instruments are SN75188 and SN75189). This solution uneconomical, however, because two chips are required, in which no fewer than six line interfacing devices remain unused. Also, +5V is required in ad- dition to ± 12 V. The new Type SN75155 from Texas In- struments houses an RS232-C line driver, a receiver and a 5 V converter, which makes it possible to feed the chip from a symmetrical 12 V supply. The re- ceiver is provided with a response in- put, which can be connected to a re- sistor, or a resistor and a bias voltage, for noise filtering and optimizing the response for relatively high baudrates and non-standard swings of the input signal. Dimensioning data for the re- sistor, Rc, connected to pin 6 can be deduced from the curves in Fig. 2. It is seen that Rc allows the input threshold of the receiver to be shifted over about 6 V to ensure correct response to the re- ceived signal: • left-hand curve: receiver input threshold with Rc=10 kO fitted between pin 6 and pin 1; • centre curve: response input not con- • right-hand curve: receiver input threshold with Rc=20 kQ fitted between pin 6 and pin 8. Noise rejection of the receiver can be defined by fitting a capacitor, Cc, at the response input. The curves in Fig. 3 9.138 show the input threshold voltage, Vit (y- axis), as a function of pulse duration, tp (x-axis), with Cc as a parameter. It is seen that Cc effectively raises the input threshold for pulses with a relatively short duration. Hence, needle pulses (noise) superimposed on the received RS232-C signal can be prevented from causing digital pulses at the output of the line receiver. The input of the line driver in the SN75155 is TTL compatible. The line driver has a current-limited output (Imax =10 m A). Current consumption of the chip is 10 mA typical, exclusive of the line current. | DECEPTIVE CAR ALARM This circuit is intended to trick car burglars into believing that the vehicle is fitted with an alarm system. A LED, fit- ted at a suitable location on the dashboard, flashes at very high brilliance. This is achieved by feeding it with a pulsating current of about 100 mA, which is far more than permissi- ble for continuous operation. The cir- cuit is based on a relaxation oscillator set up around unijunction transistor (UJT) Ti, which supplies repetitive pulses with a duration of a few milliseconds to darlington transistor T 2 . Via T3, the LED flasher is turned off when the ignition is switched on. Due to the low duty factor of the pulsating current, the circuit has an average current consumption of only 2 mA. Resistors Ri and R3 may have to be redimensioned to compensate the rela- tively high production tolerance on UJTs. High-efficiency LEDs are not suitable for use in this circuit, and care should be taken not to exceed a peak current of 250 mA. Finally, it may be a good idea to fit the LED near the car radio, and to secure adhesives on the side windows of the vehicle warning of the presence of an alarm system. br India September 1988 9. 1 39 I ELECTRONIC MOUSETRAP This is an animal-friendly design, whose operation is apparent from the drawing and the circuit diagram. A piece of cheese is put on a piezoelectric buzzer fitted in the mousetrap. When the mouse approaches the cheese, and trips on the buzzer, an electric signal is generated. This signal is raised in a high-gain amplifier, ICi, whose recti- fied output signal is used for controlling a relay. When a predefined sound level is exceeded, the relay is energized, a spring-operated lever is released, and the door of the mousetrap is closed. The complete box can then be taken out of doors to release the animal. The sensitivity of the sensor can be in- creased by fitting a small screw and metal plate on to the crystal element. The metal plate should have some play to ensure a soft rattle on the buzzer sur- face when this is tripped on by the mouse's feet. Preset Pi is the sensitivity control. 0 | TWO-WIRE REMOTE CONTROL This circuit enables 4 devices to be remote-controlled via a two-wire cable installed, for instance, between the cellar and the attic in the home. Oper- ation is simple: pressing switches Si to St selects the corresponding output, A1 to A4, at the receiver side. The regulated voltage supplied by IC2 is carried by one of the wires (A) in the cable. At the ‘transmitter’ side, this voltage is reduced by the drop across two, four, or six diodes, and fed back to the receiver via wire B, when S3, S2 or Si is pressed respectively. When S4 is pressed, wire B carries the full output voltage of IC2. Circuit ICi translates the returned voltage into a corresponding bit combination at outputs A1 to A4. The circuit is adjusted by pressing S4 and turning Pi until output A4 just toggles. A number of pin-compatible CMOS ICs can be used in position ICi — see the Table for the resulting con- figurations of switches and outputs. Out- puts A1 to A4 can sink 1.1 mA (logic low), and supply 0.4 mA (logic high). The relay driver shown can handle coil cur- 9-CHANNEL TOUCH-SENSITIVE SWITCH This circuit exploits some of the techni- cal characteristics of the 74HC series of CMOS integrated circuits. Contrary to standard CMOS devices, those in the 74HC series are TTL-compatible. Another benefit is that they are less pro- ne to oscillation. The 9-way touch-sensitive switch is simple to build from only 3 ICs and a handful of resistors. Circuit ICi is a 10- to-4 channel priority encoder. By virtue of its high input impedance, the 74HC147 allows 4M7 resistors to be used for creating a logic high level at the sensor inputs. When one of these is touched, the resultant low resistance to the circuit ground causes ICi to read a logic low level. When several sensors are touched sim- ultaneously, the priority encoder sup- plies the 4-bit code that corresponds to the sensor with the highest number. In the de-activated state, all outputs are logic 1. The output code of the priority encoder is latched in quad bistable IC2 — the latch pulse is supplied by NAND gate IC3. When none of the sensors is touch- ed, IC3 supplies a logic low level because the input pattern is 1111. When ICi supplies at least one logic 0, the NAND output toggles, and the 4-bit code is latched in IC2. The state of the bistable is not changed until the en- coder is returned to the de-activated state, and a sensor is touched after- 1 9. 1 41 AUTOMATIC 50/60 HZ SWITCH FOR MONITORS^ j It sometimes happens that a computer program can not be used in a particular country because it supplies the wrong field frequency for the TV set or moni- tor. Unfortunately, it is not always poss- ible to change the field frequency from, say, 60 Hz (American standard) to SO Hz (European standard) by using a conver- sion patch in the video driver. A poss- ible, but not particularly elegant, solution to this problem is to re-adjust the field synchronisation control in or on the TV set to stop the picture from rolling vertically. The circuit shown here performs this task automatically. The field frequency switch can be built into the monitor or TV set, BUT ONLY IF THIS HAS A BUILT-IN POWER TRANS- FORMER THAT GUARANTEES COM- PLETE INSULATION FROM THE MAINS. The circuit has a current con- sumption of only 30 mA, and is con- veniently fed from the supply in the TV set. The input signal can be composite video or just composite sync. The preset drawn near the relay shunts the field frequency control in the TV set when the relay is energized following the detection of 60 Hz field synchronis- ation pulses. It may not be possible in all cases to simply shunt the existing con- trol in the TV set, but this problem can be resolved by the use of a relay with one or more change-over contacts. The two transistors at the input of the circuit form a differential amplifier that functions as a comparator. The base potentials are, in principle, equal when Pi is set to 0 Q. Parts Ci and Di cause the blanking level of the video signal to be shifted such that the synchronisation pulses are at 0.6 V below the base refer- ence potential (switching threshold). To allow the circuit to work with video signals of 1 Vpp, the switching threshold can be adjusted with Pi. The amplitude of the sync pulses is 30% of the full swing of the composite video signal, ie., 0.3 V, so that the switching threshold is optimum when set to 0.5x0.3=0.16 V. The comparator is followed by an in- tegrating network that eliminates the horizontal sync pulses. The next stage in the circuit is a differentiator for the ver- tical sync pulses (50 or 60 Hz), which are given a fixed width. When the pulsetrain so obtained is inte- grated further, the average amplitudes can be arranged to lie just under and above the switching threshold of a Schmitt-trigger. In practice, it is more favourable to use a low time-constant for the integrator, so that the output sup- plies 50 Hz pulses, or a direct voltage when the circuit is driven with a 60 Hz signal. This arrangement alleviates the difficulty in accurately setting the switching threshold, and requires only one more integrator to eliminate the 50 Hz pulses. The digital signals so ob- tained are simple to use for controlling a relay. Adjustment of the circuit is simple: app- ly a 60 Hz video signal and adjust the threshold (500 kQ preset) until the relay is actuated. Then adjust the additional field frequency preset until the picture is steady. Diodes marked DUS are general-purpose, small signal, silicon types, eg. 1N914 or 1N4148. FOX HUNT Well-known among radio amateurs, the fox hunt has nothing to do with chasing an innocent animal, but is the search by a number of radio hunters for a hidden transmitter. The 'fox' proposed here is a small trans- mitter emitting a code in the 80-m band. It is powered by a 9-volt PP3 battery; during operation it draws a current of not more than 30 mA. In Fig. 1, when the output of gate N1 is high and that of N2 is low, N2 generates a pulse stream with a duty factor of about 5%. The pulse repetition fre- quency is about 1 kHz. The bursts of pulses are used to modulate the carrier generated by Ti, which operates on a frequency be- tween 3.3 MHz and 4.3 MHz. Note that Ti can work only when the output of N2 is low. The AM tone burst is amplified in N3-N6 and then fed to an aerial via filter L1-L2-C7-CB. The shape of the filter response is sufficiently sharp to ensure adequate suppression of har- monics. The power transferred to the aerial is about 200 mW. A small auxiliary circuit, consisting of a small VU meter and two diodes, is re- quired for tuning the transmitter - see Fig. 2. Terminate the transmitter with a 50-ohm resistor, connect the auxiliary circuit to the junction of L2 and aerial socket, and adjust C7 for maximum deflection of the meter The transmit antenna is made from 8 metres of suitable wire suspended ver- tically, for instance, from a tree - see Fig. 2. The base of the antenna is formed by three 4-metre long wires laid on the ground to form a suitable earth-plane. The inductor for the tuned aerial circuit consists of 42 turns (tapped at 4 turns) 9.142 Miscellaneous: Xi = quartz crystal 3.3 to 4.3 MHz. Si = miniature SPST switch. PCB Type 884036 Siemens's Type TFA1001W integrated circuit makes it possible to convert light intensity into frequency. The IC contains a photo diode and an amplifier. It delivers a current into its open-collector output that is directly proportional with the light incidence on to the photo di- ode. The pinout of the IC is shown on the circuit diagram. A capacitor connected between the amplifier output and the frequency- compensation connection ensures that the amplifier oscillates. With a capacitance of 1 nF, the output frequency varies between 100 Hz and 100 kHz, depending upon the light intensity (supply voltage = 2.5 V). The output signal has a peak value of 2 to 4 V (depending on the supply voltage). The output load should not be smaller than 50 kQ. The supply voltage may be between 2.5 V and 15 V. A current of not more than 1 mA is drawn with no light falling on to the photo diode; it increases (to an extent that depends on the output load) when the photo diode is illuminated. Siemens Application >9.143 This slide fader is an adaptation of that published earlier in this magazine 111 . It may be connected direct to the user port of a C64. In contrast to the earlier version, it can, however, control only two projectors. The data provided by the user port of the C64 are buffered by latches ICi and IC2. For which projector the data are in- tended is determined by PA2 and SP2. The latches are followed by a switching section and a D-to-A converter, IC2 and IC5 respectively. The converter pro- vides the voltage for controlling the light intensity. Then follows a stage, IC3 (ICs), that transforms data 000000 (A0 to AS) into a voltage of 2.5 V (lamp extinguished) and data 111111 into a potential of 5 V (lamp at full brightness). Lines A6 and A7 are used to control for- ward and reverse transport of the slides via relays Rei and Re 2 (Re3 and Ret). The dimmer is the same circuit as used in the earlier version (see Fig. 2). It is built into the projector; note that the triac needs some cooling. The control signals are transmitted via a 5-core cable terminated into DIN con- nectors. This means that the projectors should also be provided with a 5-way DIN connector wired in accordance with the circuit diagram. Presetting is carried out with all inputs of the D-to-A converters at logic low: Pi and P 2 are then adjusted until the pro- jector lamps just (visibly) light. This im- proves the life expectancy of the lamps. The circuit board contains the control section for the projectors, and this may be cut off. The connector for the user port is soldered to the track side of the control board. Two wires must also be soldered on the component side from terminals 2 and 7 to the corresponding pins on the connector. After the PCB has been populated and fitted in a suitable enclosure, it is in- serted (with components side upper- most) into the connector on the user port. The supply voltage is taken from this connector also. A sample program for the forward and reverse transport of slides via the keyboard, including automatic fade in or out, is given in Fig. 3. The space bar and R-key are pressed for forward and reverse transport respectively. 3 i9.145 The purposes of the prescaler, which is primarily intended as a prestage for the frequency meter in, say, an SSB receiver, are to lower the frequency to be measured and to prevent the frequent switching on the counter preset. The circuit consists of a number of oscillators, a mixer and an output buffer/filter. Its operation ensures that the output frequency is equal to the in- put frequency minus the oscillator fre- quency. Since the oscillator frequency can be altered readily, the output fre- quency is easily adapted to make the meter read the received frequency. The oscillator frequency is altered simply but effectively by switching the supply voltage only to the required os- cillator. The advantage of this arrange- ment is that the inoperative oscillators can not cause any interference. To prevent the inoperative oscillators having any effect on the required oscil- lator frequency, the oscillator signal is fed to the mixer via one of diodes D, to D 3 . This is because with an inoperative oscillator the diode will not conduct, so that there is only a small capacitance to earth. When the relevant oscillator operates, the diode conducts and presents only a relatively small resist- Three oscillators are sufficient for most applications, but for some receivers it may be necessary to add one or two; this may be done without any problems. The crystal frequencies may be calculated as shown in the following examples. In a simple receiver with a range of 1600-4400 kHz and an IF of 5200 kHz, the local oscillator runs between 6800 kHz and 9600 kHz. If the counter input fre- quency is 3 MHz max, the prestage fre- quency must not cause the output fre- quency of the mixer to be higher than 3 MHz. An oscillator frequency of 1600 kHz would, therefore, be suitable. This would result in crystal frequencies of USB: 1600 + 5198.5 = 6798.5 kHz LSB: 1600 + 5201.5 = 6801.5 kHz AM: 1600 + 5200 = 6800 kHz In practice, this would mean three ident- ical crystals that are pulled to the re- quired frequency with the aid of a trim- The SSB receiver in Ref. 1 has ranges of 3500-4000 kHz and 14000-14500 kHz, and an IF of 9 MHz. Prestage oscillator frequencies of 3 MHz for range 1 and 13 MHz for range 2 would give crystal frequencies of: range 1 USB: 8998.5-3000=5998.5 kHz range 1 LSB: 9001.5-3000 = 6001.5 kHz range 2 USB: 13000-8998.5 = 4001.5 kHz range 2 LSB: 13000-9001.5=3998.5 kHz POWER SUPPLY MONITOR The Type MB3773 IC from Fujitsu can Operation of the IC is best seen from the program being run (for instance, via be used to give a reset when (a) the Fig. 1, rather than from the circuit in Fig. an I/O gate). supply is switched on; (b) the level of 2. The upper two graphs show the The points of origin of the graphs corre- the supply voltage has dropped below a voltages monitored by the IC: the spond with the switching on of the certain value; and (c) the program has supply voltage and a pulse-shaped supply voltage, run into difficulties signal that is generated continuously by 9.146, !DD CEMTROINIICS RELAY DRIVE This circuit enables up to eight relays to be controlled via a Centronics printer interface. Since the computer ‘sees’ the circuit as a printer, the relays are driven by 'printing' characters. In essence, the circuit consists of eight bistables (ICO functioning as memory and eight relay drivers (IC2). The open- collector outputs of IC2 can cope with up to 500 mA. When the relay coils are connected be- tween + 5 V and a driver output, a 1 in the printed byte corresponds to an en- ergized relay. Bistables FFi and FF2 keep the data stream from the computer under control. When a byte is written to the circuit, the data are put on to the data lines DO to D7 . and the computer renders the STROBE- line low. This causes FFi to be set and monostable FF2 is started. Because FFi is set, the computer is given the message that the circuit is BUS Y. The Q- output of FF2 transmits an ACK-signal to the co mputer. Subsequently, the STROBE will go high, upon which the data in the bistables of ICi are clocked. Finally, when the pulse duration of FF2 lapses, FFi is reset, after which the next byte may be written. Although the computer 'sees’ the circuit as a printer, there are one or two prob- lems. For instance, GWBASIC transmits a CR/LF to the printer on termination of the program. To many printers, this signifies ‘erase print buffer’, but with the present circuit it means that without m s — e Lpt special precautions (machine language routine) it is not possible to leave the program without a change in the state of the relays. Another problem is posed by com- puters that work with the 7-bit code: only up to seven relays can then be con- trolled. Using only 16 LEDs, this VU meter can indicate74 different signal levels, which •makes it very suitable for use as a peak detector. The input signal is amplified by Atf (10-100 times, depending on the setting of P2), and rectified by D18. The poten- tial across Ci is, therefore, equal to the rectified peak value of the input signal. This voltage is applied to the non- inverting inputs of comparators Ai to A 16. Comparators Ai to Aa are provided with a fixed reference voltage that is de- rived from the supply voltage via the potential divider formed by Pi and resistors Ri to Rs. The outputs of A 1 to As not only drive LEDs Di to D8, but are also connected to the inputs if ICi. This circuit is an 8- bit priority encoder that converts the digital code at inputs DO to D7 to an 3-bit number, Q0 to Q2. This binary number is used to drive a dual 8-channel multi- plexer, IC2 Since the inputs of the multiplexing stages are supplied with the reference voltages for the first eight comparators (always with a voltage-step difference between two identical inputs), the refer- ence level of the second set of eight comparators is automatically matched to the signal level at the input. The step-difference between the refer- ence voltage to comparators A9 to A16 is one eighth that between the reference voltages to AI to A8. In practice, this means that the upper eight LEDs have a resolution eight times better than the lower LEDs. The output voltages of the multiplexers are buffered by Ais and A19, the two sections of a dual opamp Type LF412. This type was chosen because of its low off-set voltage. Dependent on the level of the input signal, some or all of the LEDs Di to Ds light. The bar formed by D9 to Dm div- ide the next voltage range (one eighth of the scale) in eight sections. The advan- tage of this arrangement is that the res- olution is virtually independent of the level of the input signal, and remains good even for low input levels. l SM SMALL LIGHT METER Many electronic components are in- tended nowadays for the camera in- dustry. One of them is Siemens's TFA1001W. This bipolar IC contains, apart from a photo diode, an amplifier and a 1.35 V voltage reference. Possible applications include light meters, elec- tronic flashing equipment, smoke detectors, linear optocouplers, and so The light meter presented here is very sensitive and has good linearity and low power consumption. It is housed in a compact case with six terminals. Other than the TFA1001W, the circuit contains only two other components. The supply voltage may vary from 2.5 to 15 V. The output current, Iq, (in essence the circuit is a light-controlled current- source) is a measure of the incident light flux - see Fig. 2. The circuit may be preset for optimal linearity in the lower region of its range by P,. If the unit is used in a dark- room, its linearity may be set simply with the aid of the diaphragm in the enlarger. Every time the diaphragm is 'stopped', the light flux changes by a factor of 2. In other cases, comparison with an exist- ing light meter is the simplest way of presetting. If the circuit is used as a stand-alone light meter, a nA-meter must be connec- ted between the positive supply line and output Iq. £ T® 2 Photocurrent l 0 = f(£,l i9.149 Ijoj l/O-BUS ADAPTOR FOR IBM PCs AND COMPATIBLES This I/O bus is based on those pub- lished earlier in this magazine (1) for the C64 and MSX computers. It is possible with only five ICs to make the signals of the extension slots on the IBM PC compatible with the timing and levels required by the I/O bus. Circuits ICi, ICz and IC3 provide buffer- ing of the data bus and address bus. The benefit of this is that the extension modules do not load the internal PC bus; it is also safer and more reliable. Circuit ICi, a programmable array logic (PAL) device, is used for decoding the addresses. The I/O bus consists of four slots, each of which has been allocated four adresses. Each slot has its own, active low, slot-select signal (SSI to SS4). Since in the IBM PC addresses 0300hex to 0310hex may be used for I/O extensions, the adaptor card, which uses 16 places) may be placed at two different ad- dresses. When wire link JPi is used, as shown in the diagram, the card is at ad- dress 0300hex. If link JP 2 is used, it is located at 0310hex. The software for communicating with the card may remain very simple as shown in these Pascal examples: writing: Port[$306]:= output reading: Input: =Port[$302] Advice for using the cards is given in Ref. 1 for MSX computers In normal use, the card may remain in the computer without any problems: conflicts with existing PC cards, such as hard disk or video, axe impossible. How- ever, when several extension cards are used, care should be taken with parallel addressing. 9.150 ele CAR-LOCK DEFROSTER The defroster consists basically of a small heating element that is fitted around the barrel of the door lock. Some additional components arrange the switching on (and off) of the el- ement. Switch Si in the diagram is a microswitch that is connected to the door handle in such a way that when the handle is moved the switch is pressed. It would have been possible to use the switch to operate a timer, but this would have actuated the heating element also in the summer months. Therefore, Si is followed by a charge pump (ICia).Every time the door handle is moved, Si is closed and Ci discharges via Si, which cuases the potential across Ca to rise slightly. Because of the ratio Ci:Ca, ca- pacitor C2 is (theoretically) fully charged after Si has been operated ten times. Circuit ICib is connected as a Schmitt trigger that has substantial hysteresis. After Si has been pressed seven times, ;ICib energizes the relay via Ti. Capacitor C2 then discharges slowly via._Rs. After a short time (1 to 2 minutes, preset by PI), the potential at the output of ICu has decayed to a level where the heating element is switched off. The quiescent current is smaller than 1 mA, so that the circuit may be connec- ted permanently to the car battery. The heating element may consist of two 5-watt resistors that are clamped to the barrel with a metal strap. Heat gener- ation is dependent on the value of the resistors: to prevent damage to the paint, it should not be too high. If the control circuit is also fitted in the car door, it should be contained in a waterproof enclosure, because there is always a danger of water entering the door via the window. i9.151 GIANT LED DISPLAY Kingbright's giant LED displays are eminently suitable for use in score boards, counters, large digital clocks, and so on. Each display segment contains four LEDs in series (two for the decimal point). Because of this arrangement and the colour, the operating voltage is fairly high. For safety considerations, it is ad- visable to use a transformer in the power supply. Full-wave rectification is recommended, otherwise the display flickers just visibly. The rectified voltage need not be smoothed, however. A resistor of about 220 ohms (depend- ing to some degree on the supply voltage) may be connected in series with each segment to limit the current to about 20 mA per segment. If the num- ber of operating segments is always the same, only one common resistor is necessary in the anode or cathode cir- cuit; the segments must then be con- nected in parallel. The value of this single resistor, and its power rating, must, of course, be in accordance with the number of operating segments. The displays are available in common- anode and common-cathode versions: the second letter in the type number (A or C respectively) indicates which ver- sion. Both versions come in four colours. A letter on the display (C to M) indicates the efficacy (70 to S600 ^Cd per 10 mA - minimum values). Displays of Type K (minimum 2200 jiCd P er 10 mA) are bright enough to be seen clearly in good daylight. 6-TO-12 V CONVERTER Owners of vintage cars and motorcycles fitted with a 6 V battery will find the con- verter described here useful for operating modern accessories such as a radio, revolution counter, and others. The converter raises the battery voltage to 12 V: its maximum output current is 2 A. It may be used with positive-earth or negative earth chassis. This is made possible by wire links as shown in the circuit diagram. The circuit operates on the flyback prin- ciple, for which use is made of an elec- tronic power switch in ICi that operates at a frequency of 40 kHz. At each oper- ation of the switch, energy is stored in inductor Li and subsequently trans- ferred to capacitor C3 via Di. A filter (L2-C4) has been provided at the out- put to suppress switch-generated pulses on the output line. With negative-earth batteries, wire link A should be used, and an additional link should be fitted between the collector. 9.152 el. BYW 29-100 1 2345 and emitter connections of Ti (Ti is not With positive-earth batteries, Ti is used and wire link B should be fitted. The in- put to the circuit is then as shown in the brackets. The efficiency at full load (6 A in; 2 A out) is around 70%; with smaller output currents, it may be a little higher. R2 = 1K24F R3=12KF Capacitors: Ci = 470(i; 16 V C2 = 470n C3= 1000p; 25 V C4 = 470p; 25 V Semiconductors: Dt=BYW29-100 Tl=BC567B 1C 1 *■ LT1070CT (Linear Technology) O50pH; 3 A ■40nH; 3 A Ki;K2= 2-way terminal block. Heat-sink (or Dt and ICt. .9.153 NEW PRODUCTS • NEW PRODUCTS • NEW Terminal Blocks ‘IEC-Vejay’ Clip-on type Terminal Blocks offer a versatile method of elec- trical cable connection. Any number of terminals can be assembled on a stan- dard mounting rail with a choice of diffe- rent ratings and sizes. Terminal Blocks are available to suit Electrical installa- tions upto 650 - volts AC or DC and can accommodate cables from 1.5 s.q. mm upto 35 sq. mm Asia Electric Company • Katara .Mansion • 132 A Dr. Annie Besant Road • Worli • Bombay 400 018. Cable Stripper The TOR-IC Coaxial Cable Stripper and the TOR-1F Dual Line Stripper are an ideal complement to the electrical and electronic technicians tool kit. Davie Tech Inc. • 2-05 Banta Place • Fair Lawn • New Jersey 07410. Vibration Analyzer Machine Analyzer MK 300 is the porta- ble analyzer (about 6 kg.) with built-in soft ware programmes for automatic machine diagnosis, for on the spot deter- mination of causes and location of machine problems. This instrument, with built-in programmes based on vib- ration frequency analysis technology analyzes the vibration data and displays the resultant diagnostic information, such as misalignment, imbalance abnor- mal bearings or gears, automatically. The analysis unit has the frequency large from 10 Hz to 20 KHz, a built-in unit, a built-in graphics printer, built in memory unit and an RS 232C output standard. Mecord Marketing Pvt. Ltd. • 304 Hill View Industrial Estate, • Ghatkopar (West) • Bombay-400 086. Phone: 588552. Inverter Transformer MU-NETIC Inverter Transformers are essentially inverter cum charger Trans- formers available in 150 VA, 250 VA, 500 VA and 1 KV A models, with 12V- 0-12V or 24V-0-24V at Primary and 10V-O- 190-2 10-230-250V at Secondary. These are air cooled, double wound, base mounted. Class A, 50 Hz. Transfor- mers provided with appropriate termina- tion arrangements. MU-NETIC Inverter Transformers manufactured by Electro Service (India) are tested to meet the routine test re- quirement of IS-6297 (Part-II). Non-standard Inverter Transformers are also manufactured against Customer's specification. Electro Service (India) • 232 Russa Road South • First Lane • Calcutta- 700 033. Rocker Switches “IEC” offers a line of Snap-Fix type Rocker Switches in 6 Amps & 16 Apms, 250 Volts AC or 28 Volts DC. The range includes single pole and double pole with ON-OFF or changeover or momentary contacts. A choice of terminals, viz., sol- der, screw or quick connect plug-in type are available. The switches are available in both illuminated and non-illuminated type. The illuminated Rocker switch with a choice of Red, Green or Amber lense can also be offered as ‘Pilot’ lamp. Indian Engineering Company • Post Box 16551 • Worli Naka • Bombay-400 018. Screws PIC Manufactures a large range of fas- teners for a variety of applications. liiiffli'of 1 ?. Precision Industrial Components • F- Marine House • 11-A Navroji Hill Road • Dongri • PB No. 5153. • Near San- dhurst Road Rly Station (W) • Bombay- 400 009. • Phone: 861213 - 8553750 - 8724815. NEW PRODUCTS • NEW PRODUCTS • N Hi-Tech Materials M/s. A. A. Technology 2000 Pvt. Ltd. represents five leading companies in the field of semiconductor technology and related services. Metal Finishing En- gineers (H.K.) Ltd. - Experts in lead frame plating technology, Alphascm AG - Leading manufacturers of die bon- ders and wire bonders, Sumitomo Metal Mining Co. Ltd. - Manufacturers of alloy preforms, bonding wires, lead frames, crystals, laser rods, thick film paste and Samarium-Cobalt alloys, Pos- sehl Hong Kong Precision Machining Ltd. - Experts in precision machining, moulds and dies, CAD/CAM Interna- tional - Experts in CAD/CAM for the Electronics Industry and Semiconductor Industry. M/s. A. A. Technology 2000 Pvt. Ltd. have already set up an office near the In- ternational Airport, Delhi, and will be soon setting up a sophisticated laborat- ory for providing technical service to the custome customers. For further information, write to: • A. A. Technology 2000 Pvt. Ltd. • A-II-8, Palam Vyapar Kendra, • Palam Vihar, P.O. Carter Puri, • Gurgaon-122 001 • Haryana. Count-Down Timer This new timer provides automatic elec- trical cut-off after a pre-set duration. It has a two-digit, 12.5 mm red LED dis- play and is presettable by two decade switches. Four different ranges are avail- able: 0.1 to 9.9 seconds, 1 to 99 seconds, 0.1 to 9.9 minutes or 1 to 99 minutes. The timer works on 220 ± 15% volts A.C. mains supply and provides a set of Change-over relay contacts, rated 5 amps (resistive) at 220 volts A.C. When the supply is connected, the relay acti- vates instantly and the display reads the set time. It then counts downwards, showing the remaining time. When the reading reaches zero, the relay cuts-off. This is sequence type ‘DDE’, Alterna- tive sequence ‘DE’ can be provided, where the relay energizes when the set interval elapses. The timer can be re- started by interrupting the supply momentarilly, or simply by pressing the ‘PRESET’ button on the front. Barrier terminal blocks are provided at the back side, for external electrical con- ION Electricals • 1/1 Mahalaxmi En- gineering Estate, • 571 , 1. J Cross Road 1 • Mahim (W) • Bombay 400 016 • Tel. 46 77 35, 46 81 57 Modular Computer The Unit comes compplete with two 5 1/ 4” Floppy Drive. Video monitor alongwith keyboard is all that is required to complete the system. This is available in standard 19" Rack with all Eurobus boards. A printer of required capacity and specification can be connected to the Rack. Additional 3 Slots are provided for user defined expansion, such a sys- tem can be used as development system for programme development and debug- ging. Arun Electronics Pvt. Limited • B/125- 126 Ansa Industrial Estate • Saki Vihar Road • Bombay 400 072. Tel: 58 33 54/ 58 71 01. Modular Terminal Blocks G. H. Industries introduces Linear Plug Connector with Screw connection. The Male part can be soldered on PCB and wires are screwed on the Female Termi- nal Block. Pitch 5-0 mm OR 5.08 mm, Pin Dia 1 mm. Current capacity : 5 AMPS. Operating Voltage : 250 Volts Breakdown : 2 KV. Contact resistance : Less than 15 milli ohms. G. H. INDUSTRIES • 84-B, Government Industrial Estate • Kandivli (west) • Bombay -400 067. • Phone: 605 0097, 605 3277 Relimate Connectors M/s. HARISONS introduces Relimate Connectors in Pitch 2.5 mm, Pin Dia 0.8 mm. Range available 2 to 22 way. The connector can be directly replaced to Imported Type of connector without changing PCB Hole dimensions. It can withstand wave soldering temp. The wires are soldered by semiautomatic sol- dering machine. Current Rating 3 AMPS. Operating Voltage 50 Volts. Bombay-400 067 • Tel. 605 3277, 605 0097 9.156, NEW PRODUCTS • NEW PRODUCTS • N jumps to the original setting, ready for the next use. At the same time instant, a buzzer beeps for 10 seconds. The count down can be ‘held’ or resumed by the pressing the start/stop switch. Choronograph function is also available in MQT 86. This mode is useful for measuring time intervals upto 12 hours. Resolution is 1/100 seconds for first 30 minutes. 3'A digit, 7 segment RED LED display, 7 Measurement rages with lowest range of 2 ohm with 1 milli ohm resolution and highest range of 2M ohm with K ohm re- solution . Other ranges 20 ohm, 200 ohm , 2K ohm, 20K ohm and 200K ohm. 4 wire measurement. Accuracy ± .1% of range ± .1% of reading ± 1 digit. Economy Electronics • IS Sweet Home • Plot No. 442 • 2nd Floor • Pitamber Lane • Off Tulsi Pipe Road • Mahim • Bombay 400 016. Alarm Annunciator ICA Alarm Annunciator is a centralised fault indicator for diverse applications including Power Generation, Fertilizers, Chemical & Petrochemical Complexes and variety of process Industries. Type tested from Government approved laboratories like CPRI, Bangalore & IDEMI Bombay the unit uses CMOS In- tegrated Chips. Any sequence can be given with the help of EPROM. Various types of inputs like voltage, current, can be accepted. Inputs are optoisolated. Dual lamp indication for each Alarm is provided with cold current bias. Addi- tional outputs are provided fo mimic in- dication or interlocking. The system is available in Integral and in split architecture with modular design. Tetracon system Tetra-CON is a universal microproces- sor based control system, introduced by TETRATECH ELECTRONICS. It can be used for industrial control and auto- mation applications, cither directly by the user or by manufacturers of control panels and automation systems as an OEM module. Tetra-CON can also be used in various types of machine control applications and for automation of exist- ing machinery. The Tetra-CON system consists of two standard cards, one CPU card and one Keyboard/Display card. Custom built in- terface cards can be used for system ex- pansion. The basic system provides 32 Input/Output lines, one Rs 232 compati- ble serial channel, display upto 8 digits and keyboard upto 20 SPST keys. Analog, Optoisolated Input/Output, Relay output and other interfaces can be provided on interface cards as per cus- tomer requirements. TETRATECH ELECTRONICS • 10 Usha, Opposite Central Bank, • Hanu- man Road, Vile Parle (East), • Bombay- 400 057 Promotion • BIk. # 4, Fir. #1*10, Sub- hash Cross Lane • Bombay- 400 057. Digital Ohmmeter ECONOMY has introduced Bench type instrument for the measurement of vari- ous resistors in manufacturing process of Electrical Heaters, coils, relays. Also useful for the electronic industries for in- spection of proper values of resistances before these are stored for production department, testing laboratories and This battery operated hand-held timer is R&D centres, designed for laboratory, Sports, indust- rial engineering and many other timing applications, and can be also used as an alarm time piece. It features a low power C-MOS LSI Chip, 6 digits LCD display I HE and quartz control for precision timing. ■ Rugged construction, use of standard B-^t penlitc dry battery and case of opeation make MOT 86 an ideal choice (or con- M/s. Industrial Controls & Appliances Pvt. Ltd. • 47-49A, Chakala Road, • An- dheri (East) • Bombay- 400 093. The count-down timer is adjustable upto 1 1 hours and 59 minutes. Once set, it can be started by presetting button. The dis- play then counts down to 00: 00/00 and NEW PRODUCTS • NEW PRODUCTS • NEW Digital Temperature Indicator/ Controllers ‘MECO' Digital Tcmcprature Indicator has been designed for inputs from Thermo couples RTD (Pt 100) or Semiconductor. Depending upon the sensor used, the indicator can be calib- rated from -200°C to 1600°C. The cut off temperature can be set by thumb wheel switches, potentiometer or push button. The control Accuracy is ± 1°C. Indica- tion Accuracy depends on the type of in- dicator used ± 0.1% FS to 1% FS). The output is thoroughly relay contacts (Po- tential free) rated to 5 Amps at 230V AC. The instrument is provided with Automatic cold junction compensation. Bright Red 0.5” LED’s display the temp- erature. The Dimension confirm to DIN standard 96x96 mm. Gujarat Electrical Instruments • Plot No. 624 • G.I.D.C. Estate • Phase IV, Naroda • Ahmedabad-382 330. Digital Panel Meters PUNEET make Digital Panel Meters are made in standard DIN size of 96 x 96 mm. and in the open card Types for mea- surements of 57 parameters like AC/DC Voltages and currents, frequency, wat- tage, resistances, temperature, PH etc. It is operated on Mains or + 5 VDC and measures upto 2000 V and 2000 A. They are suitable for use in Controls Panel and industries like electronic, chemicals, pharmaceuticals, power generation, cable & transformer manufacturing in- strumentation, and by hobyists, servic- ing personnels, R & D and laboratories. M/s. Puneet Industries • H-230, Ansa In- dustrial Estate • Saki Yihar Road • Bombay-400 072. Programmable Sequencer ICA Programmable Sequencer Model 1155PS is a microprocessor based time control unit for any kind of timing and sequencing requirement. The applica- tions range from batch processing in chemical plants. Foods processing Water Treatment Plant. Automation of Process and other Cyclic operations. Operations Cycle of the Sequencer is divided in Number of Stages (Max. 94). Each stage iis on line Programmable upto 99 hrs, 59 Min, 59 second with programmable out- put combination. Maximum capacity of % outputs in Multiple of 16 (Modular design). The Sequencer can be programmed for single run or Cyclic Operation. Upto 8 Programmable inputs for interlocking or for remote operation of sequence are provided. Total Manual Control for each output. Outputs is provided in the form of Potential free contracts, rated at 230V, 5A. Industrial Controls & Appliances Pvt, Ltd. • 47-49A, C'hakala Road • Andhcri (East) • Bombay-400 093. Microsequencer Micronix offers a fully programmable microsequencer for use in automatic control of standard hydraulic presses. The unit has a lot of flexibility in design of press control circuitry which is not of- fered by conventional contactor logic de- sign. The unit also has all Timers and counters build-in. Salient features include: Extremely easy programming method to allow even an unskilled operator to handle the system without special train- ing. Membrane-touch panel Keyboard. 4-digit multipurpose display alongwith status indicators to monitor full system status at all times. Fully modular construction with opto- isolated card designs. Optional battery back-up to retain prog- rammed data during power failures. Mode of operation include Manual, single cycle or fully automatic cycles with all the required interlocks. It accepts a variety of inputs such as mic- roswitch, optical or proximity sensors. It provides change over contacts for con- trol outputs. The unit works on 230 VAC and is enclosed in DIN standard enclo- sures suitable for panel mounting. M/s. Micronix • D-74, Angol Industrial Estate • Udyambag • Karnataka- 590 008. R.N. No 39881/83 MH/BY WEST - 228 UC No- 91 V A S A V I VASAVI ELECTRONICS VtUN OSCILLOSCOPES For direct measurement of Inductance Capacitance & Resistance with the highest possible ranges and Simultaneous display of Tan Delta VLCRl is the only instrument in India covering the Widest ranges of 0. 1 pf/uH/m ohm (i.e. 0.000 1 ohm.) to 20,000 uf/2000H/20 M ohm. Unique 3 Terminal Component Tester lmV/div. Sensitivity, Z-Modulation X-Y Operation, X-Y Magnifiers, X-Y Variable Controls TV Line & TV Frame, Trigger Indicator DIGITAL METER