Annual 1987— Circuits Special September 1987 INDIA Rs. 15.00 Enthusiastic enquiries are flooding our office, Orders are pouring in, and the VEGAKIT Range is expanding to meet new demands from the hobbyists. FOR tHE overwhelming: RESPONSE TOSlGAKil READERS ALL OV«R tWDlA.' - U -- 1 — ,£> -s * 5 VEQAKITS Always the first choice in Electronic Hobby Kits GALA ELECTRONICS 20, Kalpana Building, 1st Floor, 357, Lamington Road, Bombay 400 007. Tel: 363549 VASAVI’s VLCR7 SAVES YOU FROM AGONY OF BRIDGES. Measurement of INDUCTANCE. CAPACITANCE, RESISTANCE are greatly simplified by V/LCR 7. No balancing, no adjustments. VLCR 7 gives you directly the digital reading of value and its loss factor simultaneously. VLCR 7 is the only instrument in India covering the widest ranges of 0.1 pf/uH/m ohm lie. 0.0001 ohm) to 20,000 uf/200H/20 M ohm. OSCILLOSCOPES [fE] 0 0 0. 0 Component tester in our oscilloscopes is altogether different from other’s. In addition to all Oscilloscope funtions component tester provision can test all components passive and active, in circuit and out of circuit. ONLY OCR SCOPESCAN recognise NPN/PNP. Distinguish HIGH/LOWGain, Distinguish AUDIO, R.F., SWITCHING transistors. J DIGITAL FREQUENCY COUNTER VDC 18 , H -ar I l IJ U U U U\ U Smallest size n ever made in INDIA Battery cum mains operated wide frequency range 30MHz (our model VDC 19 is Basic sensitivity of lOmv Bright LED display 7 digit MULTI FUNCTION OSCILLATOR rJJ-LA/ « Pure Sine wave output. * Amplitude settable with ease down to millivolts. X 1. X 1/10 and X 1/100 attenuated outputs available. ♦ No amplitude bounce when freauency varied. VASAVI ELECTRONICS (Marketing division) 630. Alkarim Trade Centre. Ranigunj SECUNDERABAD-500 003. ph: r0995 gms: VELSCOPE elektor mdia September 1987 9-03 SCL 6500 FAMILY COURSEWARF ^ jSfigL Stocks available with WEST: Sales Industrials, Bombay, Ph: 2872169, General Electronics, Bombay. Ph: 4946422/ 897953/4946553 International Electronics, Pune, Ph: 65253. Hari Electronics, Ahmedabad, Ph: 342563/344697 NORTH: Mohan Radios. Delhi.Ph: 234578/2518675. U P. Electronics, Saharanpur.Ph: 7318. Bharat Radio Centre, Lucknow, Ph: 45808. SOUTH: S.P.E.E.D.(P) Ltd., Madras, Ph: 452931/454810. Micro Electronics Incorporated, Madras Ph: 450217, Namtech Consultants Pvt. Ltd., Bangalore. Ph: 561234. Safeguard Micro System, Bangalore, Ph: 568324. Shilba Electronics, Secunderabad, Ph: 73792. Marketing Engineers & Consultants, Hyderabad. Ph:221293. EAST: Sham Electronics, Calcutua, Ph: 274867/269879 Premier Micro Systems Pvt. Ltd., Calcutta, Ph: 359039. India International. Patna.Ph 52616 9-04 elektor India September 1987 READ THIS AD SLOWLY Discover the speed and power of SCL 6502 Microprocessor It's fast Pipeline Architecture enables this Microprocessor to execute an instruction as it fetches the next one, saving few cycles, every time in the process. Easy to handle instruction set A simple set ol 56 instruction when mixed with 13 addressing modes provides a versatile language for your programming needs. Other prominent features. * True indexing capacity ★ Programmable stack pointer * Addressable memory up to 64K* Choice between external and on chip clock ★ 8 bit parallel processing ★ 8 bit bidirectional databus ★ Expandable, using support chips through interfaces. Documentation Extensive documentation, dealing with every bit of the chip thoroughly, helps you to integrate your imagination with the potential of the chip. Support There is a team of live-wire engineers, who can help you to realise the possibilities the chip offers In every facet - generating the concept, conceiving and implementing the design, mapping your needs, etc. - They will be there guiding and working with you. SCI. (5.102 lilt* matchless chip! m Semiconductor ^Complex Ltd. Regd Office Semiconductor Complex Ltd (A Govt of India Enterprise) Phase VIII. S A S Nagar-160 059 (Near Chandigarh) Punjab Phones 87585. 87265. 87397. 87409. 87310. 87443 Telex 0395-270 LSI IN Gram CHALSI Delhi Office 12/48. Maicha Marg. 1st Floor. Chanakyapun New Delhi- 110 021 Phones 30144554 .3010426, Telex 031-65118 TTTTTTTTTTTTTTmm Arif CONTENTS e\®° ijuiv^ ' 0 ANNUAL 1987— CIRCUITS SPECIAL Volume-5, Number- 9 September 1 987 Publisher: C.R. Chand arena Editor: Surendra Iyer Technical Editor : Ashok Dongre Circulation: J. Dhas Advertising: B.M. Mehta Production: C.N. Mithagari Address: ELEKTOR ELECTRONICS PVT. LTD 52. C Proctor Road, Bombay-400 007 INDIA Telex:'(011) 76661 ELEK IN Overseas editions: Elektor Electronics Steadiest House Beth Piece High Street. Barnet Hens EN5 5XE U K Editor Len Seymour Pulitron Publicacoes Tocmces Ltd* Av Ipiranga 1100. 9°andar CEP01 040 Sao Paulo — Brazil Editor: Juliano Barseli Elektor sari Route Nationale. Le Seau B P 53 592270 Bailleul — France Editors: D R S Meyer: G C P Reeder sdorf Elektor Verleg GmbH Susterfeld-StraBe 25 100 Aachen — West Germany Editor: E J A Krompelseuer Elektor EPE Karaiskaki 14 16673 Voula — Athens — Greece Edilor: E Xanthoulis Elektor BV Peter Treckpoelstraat 2 4 6191 VK Beek - the Netherlands EditorP E L Kersdmakers Ferreira & Bento Ida R D Estefama. 32 1° 1000 Lisboa — Portugal Editor: Jorge Goncalves Ingelek S A Plaza Republice Ecuador 2 28016 Madnd Spam Editor A M Ferrer In Pert: Kedhorn Holdings PTY Ltd Cnr Fox Valley Road & Kiogle Street Wahroonga NSW 2076 — Australia Editor: Roger Harrison Electronic Press AB Box 63 Editor: Bill Cedrum The Circuits are lor domestic use only The submission o! designs of articles of Elektor India implies permission to the publisher to alter and translate the text and designed to us the contents in other Elektor publications and activities The publishers cannot guarantee to return any material submitted to them All drawings, photographs, printed circuit boards and articles published in Elektor India are copyright and may not be reproduced or imitated in whole or part without prior written permission ol the publishers Patent protection may exist in repect of circuits, devices, components etc described m this magazine The publishers do not accept resonsibility (or (ailing to identity such patent or other protection MEMBER Printed at : Trupti Offset: Bombay 400 013 Ph 4923261. 4921354 Copyright £ 1987 Elektuur B V. The Netherlands 29 News • News • News 1 34 New Products 161 Datalek 164 Classified Ads 164 Index of Advertisers Audio & music 52 Buzzer driver 86 Compressor 39 Current corrected AF amplifier 89 Digital audio selector 61 Digital volume control 45 Headphone amplifier 42 Integrated stereo amplifier 72 Limiter for guitars 44 Loudspeaker protection 77 Patch catcher 83 Simple preamplifier 88 Single-chip 40 W amplifier 7 1 SMD headphone amplifier 66 Stereo indicator 98 Stereo preamplifier with tone control 102 Low noise riaa preamplifier 113 Audio line drive Computers & microprocessors 92 4 -way DAC extension 47 16-key input for MSX micros 50 32 Kbyte pseudo ROM 60 A-D converter for joysticks 41 Bidirectional serial-parallel converter 79 Bus direction add-on for MSX extensions 58 Communication program for C64 87 Discrete DAC 57 Drive selector 48 Level adaptor for analogue joysticks 38 Light detector 68 Serial data converter 43 Simple D-A converter 67 Synchronization separator 96 Logic families 108 Ram Extension for quantum leap 124 Bidirectional parallel interface for C64 Design ideas 52 Band-gap voltage reference 78 Decoupling in logic circuits 66 HC -based oscillators 89 HCU/HCT-based oscillator 82 Low voltage drop regulators 56 Opamp based current source 65 Pierce oscillator 69 Transmission line for TTL circuits 116 Deglitcher Domestic 64 7 -digit code lock 95 Central heating control 53 Current monitor and alarm 79 Light sensitive trigger 34 Mains failure alarm 65 Thermometer 49 Toilet pointer 105 Starting-pistol simulator 109 Synchronized slide changer 110 Fruit machine 114 Telephone light 123 Fishing aid 1 28 Electronic sand-glass 131 Forced cooling for refrigerator Generators, oscillators and test equipment 80 8-channel voltage display 36 Duty factor analyser 84 Function generator 66 HC— based oscillators 89 HCU/HCT-based oscillator 65 Pierce oscillator 75 Precision crystal oscillator 84 Precision rectifier 93 Variable Wien bridge oscillator 74 Wien bridge oscillator 100 Simple sweep generator 106 Logarithmic sweep generator 1 12 Instrumentation amplifier 1 14 Two-tone RF test oscillator 121 Low current ammeter. 127 Sample & hold for analogue signals 1 30 Divider cascade Miscellaneous 49 6-way channel selector 56 Auto focus for slide projector 52 Buzzer driver 39 Display intensity control 83 Flashing lights 59 Flashing rear light 85 Halogen lamp dimmer 48 Section indication for model railway 62 Speed control for DC motors 90 Speed control for R/C models 38 Timer for fixing bath 81 Tracking window comparator 99 Timer for soldering iron 107 Power switch for cars 1 1 5 Non-interlaced picture for electron 1 1 7 Noise blanker 119 Wiper delay - 1 20 Driver for bipolar stepper motors 1 22 Fast starting wiper delay 125 Ergonomic Thumbwheel switch 1 26 PWM Driver for DC motors 1 32 Computer controlled enlarger Power supplies & ancillaries 52 Band-gap voltage reference 63 Current indicator for 723 53 Current monitor and alarm 35 Digital voltage/current display 96 Loss-free supply protector 82 Low voltage drop regulators 70 Tunnel diode battery charger 1 1 3 Battery charge/discharge indicator RF & video 76 Four-way aerial switch 51 High level passive DBM 34 High level wideband RF preamplifier 73 Morse filters 54 Multi-mode uP-controlled IF module 40 RF module for IDU 94 Switchable bandselector 67 Synchronization separator 91 Synthesizer for SW receiver 97 Front-end for FM receiver 1 1 1 Weather satellite interface 1 1 8 Video distribution amplifier 1 32 Front-end for SW receiver Rlektcw india September 1987 9*05 HIGH LEVEL WIDEBAND RF PREAMPLIFIER A linear RF amplifier can be made in two ways: (1) with the aid of a linear active element, or (2) with a non-linear element operating with negative feed- back. This circuit is of the sec- ond kind, using an RF power transistor as the active element. Feedback is also required to ensure correct termination (SO Q) of the aerial, since bipolar transistors normally exhibit a low input impedance. Also, the noise figure is not increased because virtually no signal is lost. The common-base amplifier is based on a UHF class A power transistor Type 2NS109 from Motorola. The feedback circuit is formed by RF transformer Tri. The input and output impedance of the preamplifier is 50 8 for optimum perform- ance. Network R3-C5 may have to be added to preclude oscil- lation outside the pass-band, which ranges from about 100 kHz to 50 MHz. The gain is approximately 9.5 dB, the noise figure is between 2 and 3 dB, and the third-order output intercept point is at least 50 dBm. The input/output transformer is wound on a Type FT37-75 fer- rite core from Micrometals. The input winding is 1 turn, the out- put winding 5 turns with a tap at 3 turns. B by A Treps This circuit was originally developed to detect and signal interruptions of the mains supply to artificial respiration systems. The signalling is done in two ways: a buzzer is sound- ed, and a small lamp is quenched. The supply current to the monitored equipment induces a variable flux in a small trans- former that serves to keep the relays actuated, so that Lai lights and Bz is off when the mains voltage is available. When a i mains failure occurs, apparatus .X no longer draws current, so that both Rei and Re 2 are de- actuated, resulting in the lamp being turned off, and the battery-operated buzzer being activated. Transformer Tr 2 is a modified 3VA mains type which func- tions as a current transducer: the original primary winding functions in this application as the secondary, while the original secondary winding is replaced by about 7 turns of 20SWG (0 1 mm) enamelled copper wire. Every precaution should be taken to ensure that 3-34 the new winding is capable of safely handling the current demand of X. Thanks to the so created high turns ratio in the transformer, a relatively small current suffices to keep the relays actuated and the smoothing capacitors Ci-Cj charged. Push-button Si makes it possible to test the alarm by simulating the absence of induced current. Tri can be a small bell type, or one salvaged from a mains adapter for a pocket calculator. Switch Sa, finally, is used to turn off the buzzer when apparatus X is disconnected or switched off. if elektor mdia September 1987 DIGITAL VOLTAGE/ CURRENT DISPLAY This V/I display module is eminently suitable for building into an existing DC power supply, where it gives a precise indication of the set voltage or the current consumption of the ‘ load. The circuit diagram appears in Fig. 1. The 3-digit readout is based on A/D converter Type CA3162 and BCD-to-7 segment decoder Type CA3161, both from RCA. The common anode connections of LED displays LD1-LD3 are successively con- nected to the positive supply line via Ti -Tj . Provision has been made to sel- ect the correct position of the decimal point. In the voltage range, the decimal point lights on LDi, and the resolution is therefore 100 mV. Two current ranges are possible: 0-9.99 A Oink a) or 0-0.999 (.999) A (link b). The current sensing resistor is therefore either 0R1 or 1R0 — see Fig. 2. It is important that R« does not affect the output volt- age of the supply in question. It must, therefore, be fitted ahead of the voltage divider that con- trols the output voltage. DPDT switch Si selects between voltage and current readings. When voltage measurement is selected, Pe-Ri attenuates the input voltage by a factor 100. Also, point D is pulled low so that the decimal point on the LS display, and the "V" LED are illuminated. When current measurement is selected, the drop across the sensing resistor is applied direct to the HI-LO inputs of DAC ICi. The sensing resistor has such a low value as to render the voltage divider ineffective. There are four adjustment points in the module: Pi: current range nulling; P 2 : full-scale current calibration; P 3 : voltage range nulling; Pa: full-scale voltage calibration. These points should be ad- justed in the above order. Two presets, Pi and P3, are required to ensure correct nulling of the module. Pi compensates for the quiescent current consumption of the regulator circuit in the supply. The resulting small negative deviation in the volt- age range is compensated by P 3 . The V/I display module is con- veniently fed from the unregu- lated voltage available in the supply (max. 35 V)— see points E and F in Fig. 2; bridge recti- fier Bi may then be omitted. The minimum input voltage for IC3 is 8 V, and this regulator should be fitted with a heat-sink if the input voltage is greater than 12 V. It is, of course, also poss- ible to power the module from a separate 8 V; 200 mA mains transformer. elektor india September 1987 9-35 The unit can be constructed as a double to obtain simultaneous V and I readings. It should be noted, however, that the current sensing resistor is short-cir- cuited via the ground connec- tions when both modules are fed from the same supply. There are two ways to over- come this problem. One is to feed the V unit from a separate supply, and the I unit from the "host” supply. The other is more elegant and entails hard wiring points E to the left side of the current sensing resistor. Note, however, that the highest V indication then becomes 20.0 V (Re drops 1 V max.), since the voltage at pin 11 may not exceed 1.2 V. Higher voltages can be displayed by selecting the lower current resolution, i.e., R$ becomes 0R1. Example: Re drops 0.5 V at a current consumption of 5 A, so that 1.2— 0.5 = 0.7 V remains for the voltage indication, whose maxi- mum reading is then 100x0.7= 70 V. Again, these compli- cations only arise when two of these modules are used in a single supply. fW Resistors (±5%) Ri=82K R2 ;Rj = 82R R« = 15K Rs = 27K R« = 0R1 or 1R0‘ Pi = 50K preset Pz= 10K preset Pr = 10M preset P« = 1 KO preset Capacitors: Ci =270n C 2 ;C3 = lOOn Ca = 470 p ;25 V Semiconductors: Di;D 2 = LED red Bi = BC40C1000 LDi;LDj;LD3=7750 Ti. Ts incl. = BC640 T. = BC547B Ts = BC557B ICi = CA3162 ICj = CA3161 ICs = 7805 Miscellaneous: Si = miniature DPDT switch. PCB Type 87468 (see Readers Services). DUTY ANALYSER by R. Behrens straightforward. A PLL, ICs, is used to multiply the input signal by a factor 100 and to clock counter IC 6 -IC 7 , whose BCD outputs are applied to dis- play drivers IC 2 -IC 3 . The carry output of IC7 is fed back to the phase comparator in the PLL. The counter state is only latched and displayed upon the falling edge of the input signal. Since the counter always counts up to 100 (leading edge of the input signal); the output state that exists upon detecting the trailing edge corresponds to the percentage of the pulse duration in relation to the Applications of this duty factor meter include adjusting and setting up ignition systems, switch mode power supplies, PD modulators, and sensor signal converters. The circuit itself requires no adjustment, and has a duty factor resolution of 1%, or 1° in terms of the dwell angle. The duty factor range is 1% to 99% in the frequency range from 1.5 Hz to 10 kHz. The analyser is fed from 12 V and consumes only 50 mA, so that it can be readily used in a car. The measuring principle is -36 elektor india September 1987 period. Example: assuming that the duty factor of the input signal is 60%, the counter is started at state 00 on the leading edge of the input signal, and is at state 60 when the trailing edge commences, so that ’60’ is latched and displayed. The latch pulse is generated with the aid of monostable ICi and timing parts Rt-Ci, while R22-C4 ensure that the display does not flicker when the input fre- quency is equal or close to the sample frequency. Each display value is so retained for about 0.5 s. Switch Si selects between duty factor (position 2, 0-99%) and dwell angle readings (pos- ition 1, 0-90°). The latter scale is obtained by programming a divide factor, and hence a PLL multiplication factor, of 90 with the aid of NAND gates N3-N4. The input impedance of the duty factor analyser is 100 kQ. Input signals should be at least 8Vp P : a suitable preamplifier set up with a switching transis- tor may be added to increase the sensitivity. Finally, it is rec- ommended to fit the dashed diodes at the input of Ni to afford protection against too high or reverse voltages. D Parts list Resistors t±5%): Ri - 10K R 2 . . Ris incl.;R 2 > =470R Rn= 100K Rir = 2K2 R.« = 470K Ru = 47K Rro = 33K R 22 = 1M5 Capacitors'. Ci = 120p C2 = 100p Cj = 10p; 16 V C 4 — 1 m; 16 V C»;C? = lOOn C« = 220n Semiconductors: Di = 1N4148 D 2 = LED green Ti = BC547B ICi = 4528 IC2:ICi = 4543 104 = 4093 ICs = 4046 IC«;ICr = 4029 LDi;LD2= common anode type, e g. 7651 or 7766. Miscellaneous: Si = miniature SPDT switch. PCB Type 87448 Isee Readers Services). N1...N4= IC4 = 4093 2 elektor india September 1987 9 "37 by A Behrens 9V When, after developing, photo- graphs are immersed in the fix- ing bath at irregular intervals, it becomes difficult to observe the correct fixing time for each of these. This problem is solved by the present timer, which is capable of "remembering” up to 32 immersion times, and auto- matically provides a signal when a photograph is to be taken out of the bath. Any time a photograph is immersed in the fixer, the user presses the start key on the timer, which responds by lighting a LEU When the fixing interval has lapsed, the timer provides a short beep. The circuit is composed of a 64-stage shift register which is loaded with zeroes on power up, because it lacks a reset input. Electronic switch ESi connects the frequency deter- mining capacitor to the input of clock oscillator Ni. The logic level that exists at the Dm ter- minal of IC3 is shifted towards output 0 at a speed that is defined by Pi, which enables defining fixing times between roughly 1 and 10 minutes, 9 min- utes being a commonly used value. When the start button is pressed, S-R (set-reset) bistable N2-N3 toggles, and LED Di- lights. A logic 1 is written into the shift register with the aid of a positive pulse transition applied to terminal CP. After 64 clock pulses from Ni , the logic high level is available at the out- put of the shift register, and enables oscillator N< to drive piezoelectric buzzer Bzi. The LED is turned off shortly after the start button is pressed, because the bistable is reset by the CL. OUT pulse from IC3 The timer is conveniently fed from a 9 V battery, and should not consume more than about 10 mA. Th ■W ,'MO .li LIGHT This circuit provides a com- puter with information about the presence of daylight. Poss- ible applications include auto- matically measuring the dur- ation of the daylight period in an autonomous weather station, or in control systems for outside lighting around the home. The computer can be programmed to monitor the output of the light detector, and automatically arranges for the relevant lights to come when it gets dark. The circuit is simple enough to enable ready construction on a piece of veroboard. Its output is TTL compatible, and logic low when the phototransistor de- tects light. The sensitivity can be made adjustable for par- ticular requirements by replac- ing Ri with a series connected 10 KS preset and a 270 Q re- sistor. St -38 elektor india September 1987 The majority of modem AF power amplifiers drive the loudspeakers) with a voltage that is simply a fixed factor greater than the input voltage. It is fairly evident, therefore, that the power delivered by such amplifiers is inversely pro- portional to the loudspeaker impedance, since the cone displacement of a loudspeaker is mainly a function of the cur- rent sent through the voice coil, whose impedance may vary considerably over the relevant frequency range. In multiway loudspeaker systems, this diffi- culty is overcome by appro- priate dimensioning of the crossover filter, but a different approach is called for when there is but one loudspeaker. This amplifier is based on cur- rent feedback to ensure that the current sent through the voice coil remains in accordance with the input signal. The current through the voice coil and R? develops a voltage across the resistor. A negative feedback This is a light dependent voltage source that regulates the supply to 7-segment dis- plays in accordance with the intensity of ambient light. The regulating action is positive, ie., a higher ambient light intensity results in the circuit raising the supply voltage to the displays. Phototransistor Ti does not con- duct when it detects darkness, and the base of Tj is therefore grounded via R 2 and Pi. This causes the voltage at the emit- ter of this pnp darlington tran- sistor to be about 1.2 V. The voltage across Rs is the refer- ence potential, 1.25 V, of the Type LM317 regulator, so that Irs is about 5.7 mA, and the out- put voltage, Uo, of the circuit is Uo = 1.2 + (5.7 x 10 ’(Rs + R 3 )] = 1.2 + 1.82 = 3 volt loop is created by feeding this reference voltage to the invert- ing input of ICi. The overall amplification of the circuit depends on the ratio of the loudspeaker’s impedance, Zl, to the value of R?. In the present case the amplification is 16 times (Zl/R7= 8/0.5 =16). The connection of the opamp’s output to ground is slightly unusual, but enables the base current for output transistors Ti-Ta to be drawn from the supply rails, rather than from the opamp. Capacitor Ce func- tions to set the roll-off fre- quency at about 90 kHz. The quiescent current of the ampli- fier is of the order of 50 to 100 mA for class A operation, and is determined by R3-R4 and Rs-Rs. The complementary power transistors should be closely matched types to avoid fairly large offset currents (and voltages) arising. Some redi- mensioning of either R 3 or R 4 may be required to achieve the correct balance for the power ; output stage. The emitter cur- rent of Ti and T 2 is about 500 mA when the amplifier is ' fully driven. The harmonic distortion of this amplifier is less than 0.01% at Po = 6.25 W and Ub= +18 V. Sv Source: Texas Instruments Linear Applications. 87410 # A DISPLAY INTENSITY CONTROL when Ti detects darkness. ■ When it detects a relatively high light intensity, the base j and emitter voltage of T 2 ! increase. When the base voltage of T 2 exceeds 2.7 V, R 4 limits the emitter voltage to 3.9 V due to the constant cur- rent of 5.7 mA. T 2 no longer j conducts and the output I voltage of the circuit is 5.7 V, because the total resistance between the regulator output and ground is Rs + R3+R4 = | 1,000 S, and the current through it is still 5.7 mA. The sensitivity of the regulator is adjustable with Pi. The maxi- mum output current is of the order of 700 mA when ICi is adequately cooled. The input voltage range of the circuit is 8 to 15 V. Th 87496 elektor indie September 1987 9-39 RF MODULE FOR INDOOR UNIT by R v Tetborgh Electronics, has met a great deal of interest from our readers. Many have success- fully ventured out into the world of centimetre waves and SHF construction methods, and are proud to watch the pictures pro- duced by a home-made Indoor Unit The construction and alignment of the RF input stage and the local oscillators is undoubtedly the most difficult phase of the project. However, an attractive alternative is now available for those constructors hesitant about their skills in dealing with very high frequency compo- nents and techniques. The Type HL ECS51 is a ready-made, tunable, 950-1750 MHz to RF in (from LNB) LNB supply AGC input ♦ 12V supply tuning voltage (0...20V) IF output 480 MH* osc output 86082 - 1 87503 ■ 1 RF board is already complete, it is recommended to L8 480MHz remove MXi, R 3 and band selector St. The screen between the RF input stage and the local oscil- lators may also be omitted, but not the remaining screens on the board. Before aligning the "modular" Indoor Unit, the intermediate frequency requires lowering from 610 to 480 MHz. The VCO in the SL1451 PLL can be tuned to the new centre frequency by increasing the inductance of L«. This is easiest to do by making a new inductor as shown in Fig. 2. Use about 5 cm of silver plated, 0 1 mm (SWG20) wire to make the single turn inductor, ensuring that the underside of it is just above the PCB surface. The alignment procedure of the module-based Indoor Unit is essentially identical to that set forth on page 54 in Part 2 of 1 . The reference there to TV chan- nel 36 (600 MHz) should then be read as TV channel 21/22 (approx. 480 MHz). The IF band- filters can be tuned to the new frequency when the tmm 0 CuAg is the AGC input, which is grounded here to achieve maxi- ted in series with the existing +33 V tuning voltage rail to ensure the correct maximum input to pin 5. The IF output on the module accepts a common phono plug, to which a short length of thin (RG174) coax cable is soldered for connect- ing to the short track between pm 3 and 4 of MX. on the RF board Coupling capacitor Cz should be left in place, but MXi, the RF input stage, and both local oscillators may be omit- ted, since the module takes over their function. In case the 9-40 respective trimmers are set for nearly maximum capacitance. Finally, a note on the results obtained with the module in combination with the RF board as described here: the original, completed, and properly aligned RF board with the BFG65, local oscillators, and MXi fined gives a slightly better performance when relatively elector india September 1987 weak signals (C/N < 12 dB) are being received. This is mainly due to its noise figure being lower than that of the HL ECS51 module, which is specified for no less than 15 dB in this respect. None the less, the module gives good results with relatively strong input signals. RCK The HL EC51 module is avail- able through Bonex Limited • 102 Churchfield Road • Acton • London W3 6DH. Telephone: (01 992) 7748 or (01 993) 7631. Literature reference: 1,1 Indoor Unit for Satellite TV Reception Parts 1-3; Elektor Electronics, October 1986 and following issues. BIDIRECTIONAL SERIAL- PARALLEL CONVERTER by R. Baltissen This interface circuit enables doing rather more than nor- mally possible with the com- puter’s serial (RS232) port. Serial output data from the computer is converted into parallel for- mat, and parallel data applied to the interface is converted into a serial bit stream for reception by the computer. The interface is based on the industry standard UART (univer- sal asynchronous receiver/ transmitter) Type AY-5-1013, or the CMOS version of it, the CDP1854 from RCA. Serial data from the computer is received at input RXD, and inverted in Ti for driving the RSI input on the UART, which converts the received word into 8-bit parallel format (RD 0 -RD 7 ). The shifting in of serial bits is clocked by the 19,200 Hz signal applied to the RCP and TCP input. This fixes the baud rate of the interface at 1200 (19,200/16). The baud rate generator is a conventional design based on a binary counter/divider with built-in clock oscillator, which is crystal controlled here and operates at 2.4576 MHz. The parallel output of the UART is buffered with the aid of IC 2 to enable controlling 8 relay drivers Bi-Bs. The paral- lel word applied to the UART at its TD0-TD7 inputs is converted into serial format and output via the TSO terminal, where the signal is inverted and fed to the TXD output. The serial data format can be defined with the aid of wire links B-F: Table 1 lists the func- tion of each of these. Inverter T« automatically resets the receiver in the UART by driving RDAR (received data available reset) low whei. ?.DA (received data available) goes high to signal th t a complete word has been ch 'ted into the receiver hold register. W hen wire link A is install ed, R DA can also con- trol the TDS (transmitter data strobe) input, so that a new parallel word (TD0-TD7) is loaded into the transmitter holding register. Thus, jumper A makes it possible to use the CTS (clear to send) hand- shaking signal. The TEOC (transmitter end of character) j pulse is used here to generate the RTS handshaking signal, and also to control the TDS input, together with CTS. This handshaking input, when active, prompts the UART to out- put a new serial word. Set-reset bistable Nt-Nz precludes con- ' flicts arising between the elektor indin September 1987 9-41 signals in question. Power-on network C 1 -R 1 ensures that the UART is properly reset and initiated. TSO and TEOC then go high, while RDA is forced low. When link A is not fitted, the presence of the inverted TEOC pulse at input TDS causes the transmission pro- cess to commence. The author has developed this circuit mainly to enable two IBM PCs to communicate with the aid of the Turbo Pascal program listed in Table 2. Before this can be run, the status of serial port COM1: (AUX:) should be defined by typing DOS command MODE COMl:1200,n,8,2 (1200 baud, no parity, 8 data bits and 2 stop bits). Pins 6 (DSR) and 20 (DTR) on the 25-way D socket should be interconnected, and the same goes for pins 4 (RTS) and 5 (CTS) when no handshaking is being used. When it is intended to use the handshaking facility on the bidirectional interface, link A should be removed, and socket pins 4 and 5 connected to interface terminals CTS and RTS respectively. Sv repeat writetaux.chrfn) ) read ( aux , v) ; write(ordtv) , ' * ) rs := n+1; delay (500) ; until (n = 256) or until keydepressed end INTEGRATED AMPLIFIER The Type TDA1521 from Valvo/ Mullard is an integrated HiFi stereo power amplifier de- signed for mains fed appli- cations such as stereo TV. The device works optimally when fed from a + 16 V supply, and delivers a maximum output power of 2 x 12 W into 8 2. The gain of the amplifiers is fixed internally at 30 dB with a spread of 0.2 dB to ensure optimum gain balance between the chan- nels. A special feature of the chip is its built-in mute circuit, which disconnects the non-inverting inputs when the supply voltage is less than ± 6 V, a level at which the amplifiers are still correctly biased. This arrange- ment ensures the absence of unwanted clicks and other noise when the amplifier is switched on or off. The TDA1521 is protected against output short circuits and thermal overloading. The SIL9 package should be bolted onto a heat- sink with a thermal resistance of no more than 3.3 K/W (Rl=8 2; Vs= ± 16 V; Pa = 14.6 W; Ta = 65 °C). Note that the metal tab on the chip package is intern- ally connected to pin 5. The accompanying photograph shows that this high quality stereo amplifier has a very low component count, and is readily constructed on a piece of Veroboard. Distortion at Po=12 W: 0.5% Quiescent current: 40 mA Gain balance: 0.2 dB Supply ripple rejection: 60 dB Channel separation: 70 dB Output offset voltage: 20 mV 3 dB power bandwidth: 20-20,000 Hz 220n The following technical data are stated as typical in the data- sheets for the TDA1521 (Rl=8 2; Vs= + 16 V): 9-42 elektor mdia September 1987 from an idea by M Wiegers Two simple to build 4-bit digital-to-analogue converters are described here. One trans- lates a 4-bit BCD code into 10 analogue voltage levels, the other accepts a 4-bit binary code and outputs 16 voltage levels. Both circuits comprise a digital decoder with open collector outputs for control- ing a resistance ladder. The analogue voltage is obtained by controlled connection to ground of a particular section of the ladder, and buffering the drop so obtained with a tran- sistor. Notwithstanding their relatively low resolution (10 or 16 steps), the circuit should have many possible applications, includ- ing driving digitally controlled power supplies, triangular wave and sawtooth generators, and A- D converters. Table 1 lists the relative values of the resistoi| in the ladder network, starting from Ri = lK0. Three values are given for each resistor: the lefi-hand column shows the theoretical value, while the nearest equivalent from the E24 and E96 series appears in the centre and right hand column, respectively. Note that the starting value can be changed to individual requirements, provided all other resistors are dimensioned accordingly, ie., their values should be multiplied with the same factor with respect to 1K0. It is a relatively simple matter to add an Uth or 17th output level by driving the decoder such that none of its output tran- sistors is enabled. This results in an output voltage which is 0.6 V lower than the supply for the ladder network. In the case of the 74LS145, this condition is obtained by applying a non- valid code to the inputs, i.e., one greater than 9io (100h). Similarly, on the 74159, enable input G1 or G2 can be made logic high. Sv Table 1 Resistor values relative to 1 kQ — Rn 10-step 8CD versior 16 step binary vers ion R. 1000 1K0 1K0 1000 1K0 1K0 R* 111 110 110 66.7 68 665 R» 139 130 140 763 75 76.8 R« 179 180 178 87 91 86.6 Rs 238 240 237 103 100 102 R. 333 330 332 122 120 121 R; 500 510 499 145 150 147 R. 833 820 825 178 180 178 R» 1667 1K6 1K69 222 220 221 Rio 5000 5K1 4K99 286 270 287 Rn 381 390 383 R >2 533 510 536 R«i 800 820 806 Ri« 1333 1K3 1K33 Rio 2667 2K7 2K67 R.o 8000 * 00 8K06 s 34 mA 5V 87418 elektor india September 1987 9-43 Many modern AF power output stages are capable of deliver- ing considerable power levels in the supersonic frequency range. When the loudspeaker can not handle that power, the voice coil is rapidly overheated, and causes a short-circuit. If the power output stage is not prop- erly protected, it breaks down and supplies a direct current that effectively destroys the loudspeaker. The present loudspeaker pro- tector is composed of three sections: a measuring amplifier, a detector, and a relay driver. Four channels are shown here as an example. Potential divider Ri-Rj determines the sensitivity of the protection circuit, while Di-Dz protect the input of Ai. Opamp As is set up as a low pass filter with a cut-off fre- quency of 0.5 Hz, so that it can PROTECTION function as a DC detector. The second section of the circuit is composed of four detectors A9-A12. As compares any negative direct voltages to a ref- erence set with Re-Rs, while C3-R7 determine the delay time. Opamp A10 has a similar func- tion for positive direct voltages. The circuit is actuated when VmRz Q gg I5R2 Comparators An and Ai? func- tion as the power limiter. Positive and negative peak voltages are rectified in D3-D4 and averaged with the aid of R- C combinations R36-C33 and R26 - C23 . The relatively long periods of these networks precludes erroneous triggering of the circuit on peaks in the input signal. The power limiter is actuated when Vmfo /2 q gg 15 R 2 « R1+R2 R28+R2S This equation is also valid for the positive detector set up around A12. The stated component values result in P max -- 30 W in 82 . When the input signals are all right, the open collector out- puts of A9-A12 are in their high impedance state, so that the out- put voltage is + 15 V via R r^f 1 > — fen -0 1 BC857B Cl JTTT-Lf'P) 01 — vjy BC847B three wire links and jumper B-C or A-C as required. Do not forget to solder the terminals of D- (not an SMA part), and the battery connections, at both sides of the board. Also, through-contacting with short lengths of component wire should be effected at four locations. Push all the pins of two 14-way IC terminal strips through the straight rows of holes on the component side of the board, ie. the side that holds the transistors, then solder the pins to the islands on the copper side, i.e., the side that holds the 74HC00s. The pins should protrude at least 4 mm. The use of a centrally cut wire-wrapping socket is not recommended here in view of the thickness of the pins. Locate the pin that protrudes from the hole marked 1, and cut it off. Mount a turned IC pin holder next to pin 28, 27, 22 and 20 of the right-hand side terminal strip, and solder these at both sides of the PCB. These pins should not protrude at the cop- per side, and their tops should be 1.5 to 2 mm above those in the terminal strip. When it is intended to use the RAM in its 2 x 16 Kbyte configuration, wires are connected to points OE2 and CE2 at the copper side, and guided between pins 5-6 and 9-10 respectively. Remove pin 1 of a standard 28-way IC socket, before carefully push- The mixer is one of the most important sections in any good- quality SW receiver, since it determines to a large extent the sensitivity and the dynamic range. The so-called switching mixer is often used, because it has none of the technical im- perfections of active mixers. The most commonly found switching mixer is the diode- based double balanced type (DBM), which is, unfortunately, a notoriously expensive compo- nent, especially when a high intercept point is required to ensure low levels of inter- modulation. The application of active devices, such as bipolar tran- sistors and J-FETs, in a passive fitting this onto the 27 pro- truding pins at the copper side. Connect the battery supply wires and the wire to Si (NWDS) to the respective points at the component side. Use a pair of precision pliers to carefully bend pins 28, 27, 22 and 20 of the 43256 or 6264 slightly to the right of the other pins in the row. This enables pushing these four IC pins in the previously mentioned, separ- ate, socket pins, while the 24 others are inserted in the usual manner. The battery is con- veniently mounted at some distance from the module. When a miniature battery is available, this can be fitted underneath the RAM chip. For BBC users: wires OE2 and CE2 are conveniently connected to pins 22 and 20 respectively of a 28-way IC socket for plugging into the adjacent ROM/RAM socket on the BBC's sideway extension board; the NWDS signal is available at pin 8 of IC77. Switch Si is mounted at a convenient location on the com- puter's rear panel, and when opened inhibits writing into the RAM. It is recommended to open Si after turning the com- puter off to prevent the battery having to supply some 50 \i A for prolonged periods: this current flows into the NWDS driver via Rio. Non-BBC or Electron Plus-1 users should note that the NWDS signal is the same as Parts list Note: all parts Surface Mount Assembly types unless marked *. Resistors: Ri. . ,R* incl.;Rr:R»;Rio = 47K . Rs = 180R Ro;R»-1KO Capacitor: Ci = lOOn or 47rt Semiconductors: Di = AA119 • Ti;Tz = BC857B or similar pnp SMA type. Tj = BC847B or similar npn SMA type. ICi;IC 2 = 74HCOO IDo not use HCT types). IC3 = 43256C-10/12/15L (NEC) or 62256 LP10/12 32Kbyte CMOS static RAM \ Miscellaneous •: PCB Type 87500 (see Readers Services page) 2 off 14 way terminal strips with 7 mm pins. 4 off turned pins for IC leads. Suitable battery (see text, Vb S2.4 V) WRITE, not READ/ WRITE. The MOVE command in the ADT ROM available for the BBC computer enables exchanging data between resident and sideway memory. Programmers should have little difficulty, j however, in writing a short i routine that selects the relevant sideway socketfs) via the socket latch at FE3 Fh, and copying one or two 16 Kbyte blocks. Th It is regretted that information on software for this project is not available. (Ed). iW Jr HIGH LEVEL PASSIVE DBM mixer is less well established, j present design is based on a And yet, these components pair of well-known UHF tran- enable the mixer to remain rela- sistors, which require no supply tively simple, since the RF input voltage or bias circuits, signal can be thought of as j The input and output trans- electrically insulated from the j formers are wound on two-hole 1 local oscillator output. The 1 ferrite cores (Baiuns). The primary of Tra is 8 turns with a centre tap for the RF input, the secondary is 4 turns. Tri is wound such that the indicated LO amplitude is available at the secondary. Only the RF input or the IF output requires correct termination on 50 Q, the other connections are then fairly uncritical. The input intercept point of this mixer is excellent at between 31 and 36 dBm, while the noise figure and con- version loss are acceptable at about 6 dB. The LO rejection is roughly 25 dB, and depends mainly on the construction. The mixer is suitable for RF and IF signals up to 30 and 50 MHz respectively. B elekiof mdia September 1987 9 - 51 Piezoelectric resonators, also referred to as buzzers, are fre- quently used for providing aud- ible signals in all sorts of electronic equipment. Buzzers are small, light, simple to use, and yet provide a loud output signal. They are either of the passive or of the active type. The former are driven by an AF signal source, while the latter feature a built-in oscillator, and require a direct voltage only. This circuit is a double AF oscillator for driving passive buzzers. It ensures a richer out- put sound than normally obtain- able from a piezo buzzer due to the use of two oscillators, Ni and Nj, whose output signal lies between 1 and 10 kHz. Gates N3-N4 form an S-R bistable which is controlled by the out- puts of N 1 -N 2 , and drives the buzzer direct. The spectral composition of the output signal is fairly complex, due to ^ .n II I . Illl llllllllllliilllillllllllllililllllill i It is generally known that the accuracy of measurements in electronic circuits is mainly a function of the stability and reliability of the reference against which the unknown quantity is compared. There- fore, everything feasible should be done to maintain the stability of the reference, i.e., counteract the presence of both the fun- damental notes and the differ- ence and sum frequency. The timbre so obtained varies as a function of the ratio between the oscillator frequencies, which are adjustable with the aid of presets Pi -P 2 . Note that diodes D 1 -D 2 reduce the duty factor of the oscillator signals to about 25%. Optimum effects are achieved when a simple ratio is set between the oscillator fre- quencies, e.g. 3:4. The resulting waveform is always composed of rectangular signals, but these differ in respect of their period to ensure that the buzzer pro- duces a rather agreeable sound. The buzzer driver is controlled by a logic level applied to point X. The quiescent current con- sumption is virtually negligible, while about 10 mA is drawn in the actuated state. D m My ,| !j| «y J Cl MlN4148 6nfl c i 10n D2^|1N4148 P2__X R2 / 470k iTV— inr □ J yN4 u~~ir N1...N4 = ICt = 74HCT132 BAND-GAP VOLTAGE REFERENCE the adverse effects of variations in the ambient temperature, supply voltage, and load cur- rent. The zenerdiode in Fig. 1 is a usable reference device for applications where the above three parameters are not sub- ject to appreciable variation. The ’’super zener” in Fig. 2 features excellent stability and is hardly affected by variations in the supply voltage and the load current. Although the tem- perature coefficient of the super zener circuit can be optimized by careful dimen- sioning of the components, there exists a still better way for making a precision voltage ref- erence. The term band gap refers to the difference between two dis- crete energies of the outer four electrons in a semiconductor atom. Electrons in the highest energy band contribute to the conduction of the material. As the temperature is increased, some electrons gain enough thermal energy to escape from the valence (non-conductive) band, cross the band gap, and enter the conduction band, leaving the valence band unfill- ed. Thus, conductivity is a func- tion of temperature. With reference to Fig. 3, the temperature coefficient of cur- rent mirror T 1 -T 2 is compen- sated by that of T 3 . The following conditions should be met if the circuit is to function optimally: (1): Ib-lORi; (2): R 3 is dimensioned such that Vr = 1.204 V; and (3): the tran- sistors are exactly matched. The latter condition is probably best satisfied by using tran- sistors on one and the same chip carrier, e.g. those in a tran- sistor array such as the Type CA3083. The value of R de- pends on the supply voltage and the maximum output cur- rent. It should be noted that T 3 carries the output current if the circuit is not loaded, so that the resulting dissipation may give rise to temperature differences on the chip. It is, therefore, rec- ommended to permanently load the band-gap reference. The accompanying calculations prove that the output voltage of the circuit is not affected by temperature variations. Sv Band-gap reference. The reference voltage, Ur. is obtained from Ur Uwiu + l»T». R» and Ri are dimensioned such that li - 10b, so that R> drops |Uutni-Uum>| volts. When the current amplification of T* is sufficiently high, R» carries virtually all current la: I7 — UimiiWR) whence Uh Uwiift + fUatiT!. Ubcit7>)R?/R}. For identical transistors Ust rs given for different values of las as U«ni Um4t» - *77g(og«Oi/l>). Uw of T» is also expressed as Uwir».- Uacll — 17Ti*) + Umo<77To) so that Ur can be written as Ur Ubc.(1 - TVTrI 4 UwolTVT*) + R»/Ra*77gk>g*
  • SPACE 00 FOR ALL CHARACTERS: TRANSMIT CHARACTER ERASE CHARACTER TRANSMIT MARKER "RET” (END OF MESSAGE) RECEIVE ) SUBROUTINE BLINK RECEIVE CURSOR READ RX CHARACTER RX CHAR.oEMPTY RX CHAR. = RET THEN RECEIVED CHARACTER TO RECEIVE SCREEN LF FOR RX CURSOR CLEAR CURRENT LINE 87461-2 elektor mdia September 1987 other types of computer, the iunction of the major lines can be summarized as follows: 100-125: initialize the screen and the sound generator. 130: open the serial port with parameters 300 baud, 8 data bits, 1 stop bit, no parity, no handshaking, full duplex. 140: T is the base address of the transmit screen, and T0 is the associated index. R and R0 are similar variables for the receive screen, while R1 in addition gives the maximum number of character per line. 160: blink the cursor and read the keyboard buffer. 180-200: test for DELETE, and erase the previous character. 210-230: test for RETURN and transmit message. 240-260: toggle the BORDER colour when the screen is full. 270: go to the receive sub- routine. 280: repeat the above loop. 710: transmit the "begin of message" marker. 720-750: transmit and erase all characters. Monitor the receive channel for messages, after transmission of every character, reception has the highest pri- ority. 760: transmit the "end of message" marker. 810: blink the cursor and read the receive buffer. 820: buffer empty? 830: end of message. 840: have the sound generator produce a beep. 850-870: advance the cursor to the next line. 880: clear the new line. 900: display received character on REMOTE screen. 910-920: advance cursor to next position. w 3 100 FOKE5 3281 . 1 2 .PRINT " :POKE53280 . 9 : P0KE53281 .0:PRINT CHRf ( 152) : POKES 3272 . 23 110 SI=54272: POKE 24 + SI . 15 : POKE SI.207:POKE 1 +S I . 34 : POKE 54-SI. 10 120 FOR H= 1033 TO 1044: READ A: POKE H . A : NEXT H 125 FOR H= 1273 TO 1283: READ A: POKE H . A : NEXT H 130 OPEN 2,2,0.CHRf <6)+CHRf (0> 140 T= 1104: T0 = 0 : R=1344: R0 = 0: R1=0 150 REM MAIN 160 POKE T+T0.60: POKE T+T0 . 32 : GET Tf 170 IF TS = M " THEN GOTO 270 180 IF Tf OCHRS ( 20 ) THEN GOTO 210 190 IF T0>0 THEN T0-T0-1 200 POKE T+T0.32: GOTO 270 210 IF Tf OCHRf (13) THEN GOTO 240 220 GOSUB 700 230 GOTO 270 240 IF T+T0>-R-80 THEN GOTO 260 250 POKE T+T0 . ASC ( Tf ) : T0=T0+1 : GOTO 270 260 POKE 53280.1: FOR H=0 TO 15: NEXT H: POKE 53280,9 270 GOSUB 800 280 GOTO 150 700 REM TRANSMIT 710 PR I NTH 2 , CHRf ( 62 ) ; : PRINT# 2 . CHRf ( 32) ; 720 FOR K=T TO T+T0-1 730 PRINT# 2 , CHRf ( PEEK < K > > ; : POKE K . 32 740 GOSUB 800 750 NEXT K 760 PR I NT# 2 . CHRf (13);: T0 = 0 770 RETURN 800 REM RECEIVE 810 POKE R+R0.60: POKE R+R0.32. GET#2 . Rf 820 IF Rf= M - THEN GOTO 930 830 IF Rf OCHRf (13) THEN GOTO 900 840 POKE 54276.0: POKE 54276.33 850 IF R1 =40 OR R1=0 THEN GOTO 870 860 POKE R+R0.32: R1«R1+1: R0=R0+1: GOTO 850 870 R 1 = 0 : IF R+R0 = 2024 THEN R0 = 0 880 FOR H=R+R0 TO R+R0+39: POKE H.32: NEXT H 890 GOTO 930 900 POKE R + R0 . ASCCRf > : R0 = R0 + 1: R1=RH-1 910 IF R1 =40 THEN R1=0 920 IF R+R0=2024 THEN R0=0 930 RETURN 950 DATA 42.32,84.82.65.76.83.77.73.84.32.42 960 DATA 42.32,82,69.67,69.73.86.69.32.42 970 END READY. on FLASHING ! ii iiii i 11] ill REAR LIGHT ill 1 IlllnlmiiLl III 1 j ; i i i : 1 ■ i -• i ! . ; j i . « f i 1 1 WJJililii by J Donhauser The circuit is essentially com- [ minute ride, i.e., enough for the bulb is illuminated. Since C 3 is posed of a battery charger and light to flash for about 4 minutes only slowly discharged via Rt , This rear light for bicycles is fed a logic switching section. The after the bicycle is halted. A Na remains enabled for about 4 from a battery charged with NiCd battery is charged from a relay is used to switch between minutes after halting. Push- current from the dynamo, and voltage doubler C 1 -C 2 -D 1 -D 2 -C 3 operation while riding and button Si enables immediately starts to flash when the cyclist to ensure a charge current of while standing still. When the switching off the rear light, halts. To preserve battery about 20 mA when riding at a bicycle is in motion, the voltage because R 2 then discharges C 3 I power, the unit automatically reasonable speed. This makes from the dynamo, G, ensures in a few seconds, switches off 4 minutes after it possible for a charge of that N< is enabled, so that Ti Gate Ni monitors the dynamo waiting. 3 mAh to be available after a 10 actuates Re, and the small 6 V ' voltage, which is rectified by elektor mdia September 1987 9 -59 D4-C4-R3. When the direct voltage drops below approxi- mately 2 V, Ni switches on multivibrator N2-N3-N4 which causes the relay to toggle at a rate determined by R4-C5. The 5 V DIL relay requires only 11 mA, while the current con- sumption of the 4093 is virtually negligible at about 1 jiA. It should be possible to fit the circuit and the battery in a somewhat larger than normal bicycle headlight, equipped with terminals for connecting the dynamo and the rear light. Of course, due care must be taken to avoid the battery con- tacts touching the metal inside of the light. R 87446 A-D CONVEI JOYSTICKS from an idea by f Berben Although joysticks come in an astounding variety of versions, their internal organization is vir- tually always a standard con- cept, based on either a set of relatively fragile, springy, mem- brane contacts, or two poten- tiometers. Many computer en- thusiasts will agree that the latter, analogue, type offers bet- ter reliability and quality. Unfor- tunately, however, these can not be used in conjunction with a popular home micro such as the Commodore C64, and that is where the present circuit comes in. The four comparators in ICi function as switches to translate the handle movement into digital signals. The outputs of the comparators are buffered in IC2 to enable interfacing to the computer’s joystick port. The two remaining inverters in IC2, Ns and Ns, along with two inverters in IC3, function as drivers for the LEDs that indi- cate the handle position. Gates N9-N12 are set up as a wired NOR function to enable LED Ds to light when the joystick handle is in the centre position. Finally, the current consump- tion of the converter is about 25 mA. D 9-60 elektor mdia sepiember 1987 32 DIGITAL V CONTROL Many of today’s HiFi amplifiers feature a "clicking” volume control, but this is only rarely a real stepped attenuator based on a wafer switch. In nearly all cases, this expensive system is based on a normal poten- tiometer, whose spindle is fitted with a mechanical construction to simulate the stepping move- ment. A normal rotary switch is not suitable for adjusting the volume of an amplifier because it briefly disconnects the input from the signal source when operated, and so readily gives nse to clicks and contact noise. Different problems crop up when designing an electronic volume control. Of these, distor- tion is probably the hardest to master, but reasonable results are still obtainable, as will be shown here. Basically, there are two methods for making an elec- tronic potentiometer. One is to create a tapped resistor ladder which is not much different from a normal potentiometer), the other is to change the resist- ance of the two "track sections" such that the total resistance remains constant. The circuit proposed here is based on the second method, and features .5 steps in its basic form. The number of steps can be in- creased to, say, 64 by adding four switches and resistors. The electronic potentiometer is composed to two equal sec- tions, which have a total resist- ance of 15 kQ each. The electronic switches in each section are controlled by binary- counter ICs. Since the switches m section ES 1 -ES 4 and those in ESs-ESs are controlled in com- plementary fashion, the total resistance of the potentiometer remains constant. Resistors R,- cu and R 7 -R 8 serve to keep the potential at the input and output sr 0 V so as to preclude clicks when the step switch, S 2 , is operated. Switch Si is the up down selector. Gates Ns-Ns farm a bistable to ensure that the counter is clocked with debounced step pulses. The number of steps can be m creased by adding a counter and the required number of e.ectromc switches, divided I ov er the two "track sections", shown in the circuit diagram. ] of the theoretical resistance These switches are then con- Fortunately, precise binary values, and as long as the actual j nected in parallel with resistors j ratios are not required here, resistors are kept equal in both | whose values correspond to J since adequate results are sections. D binary order 1-2-4-8, etc., as \ obtainable with approximations elektrv mdia September 1987 9 * 61 Simple DC operated motors with a permanent magnetic stator behave as an indepen- dently energized motor. The speed of an ideal motor with an infinitely low internal resistance is in direct proportion to the voltage applied, irrespective of the torque. The motor thus runs at a speed at which its reverse electromotive force (e.m.f.) equals the supply voltage. The reverse e.m.f. is directly pro- portional to the force of the (constant) magnetic field, and the motor speed. In theory, therefore, the motor speed can be held constant with a constant supply voltage. The speed reduction observed in practice arises from the voltage drop across the internal resistance, Ri, of the armature winding. Thus, when the motor is loaded, its current consumption, and hence Vm, increases, reducing the effective supply voltage. This effect can be eliminated by means of Ri compensation, which essentially entails measuring the motor’s current consumption, relating this to the motor’s instantaneous drop across Ri, and increasing the supply voltage accordingly. In fact, this calls for a voltage source with a negative output impedance, since it caters for a higher output voltage when the load is increased. The basic set-up of the supply required here is shown in Fig. 1. The load current is measured as the drop across sensing resistor R 3 . The DC transfer function of this amplifier is written as U2 = Ui +ILR2R3/R1 which accounts for the negative output impedance because then Rout = — R2R3/R1 For optimum results, this impedance must be kept about equal to that of the motor. Figure 2 shows the practical cir- cuit of the motor driver based on a power operational ampli- fier. The Type L165 from SGS can supply up to 3 A at a maxi- mum supply voltage of 36 V, and is therefore eminently suitable for the present appli- 9-62 elektor india September 1987 cation. Capacitors Ci and C 2 ~ suppress noise on the reverse • e.m.f. from the motor. Due care should be taken, however, in so extending the circuit, because this readily leads to instability. The motor itself already forms a fairly complex load, since the revolving rotor winding is mainly inductive, and the rotor itself represents a fairly large capacitance. Noise suppression components such as R4 and C3 add to the complexity of the load and may result in control instability, which becomes manifest in the motor’s tend- ency to alternately reverse its direction at a relatively low rate. Also, the response to a fast change in the torque may be impaired, and high-frequency oscillation may occur (notice- able as exessive heating of ICi and/or R4). When the circuit was tested with a small PCB drill, best results were obtained by omitting R4-C3 and including C 2 . If the motor has a noise sup- pression network, C 2 must be omitted, and Rs added to pro- tect the opamp inputs against too high differential voltages as a result of commutation voltage peaks. Clearly, Di and D 2 have been included with this in mind. Preset Pi is adjusted until the motor remains stable. Over- compensation of the motor will give rise to apparently uncon- trolled movement. The adjust- ment of Pi should be carried out when the motor has not yet reached its normal operating temperature, because its self- heating gives rise to an increase in the internal resistance. The use of a symmetrical supply (+ 18 V max.) enables twoquadrant operation of the motor (cw/ccw rotation), which can then be used to power model trains and the like. The motor is halted when P 2 is set to the centre position. The ground rail may be connected to the negative supply rail if only one direction of revolution is required (PCB drills). The maxi- mum supply is then 36 V, mak- ing a greater voltage available for the motor, so that 24 V types can be controlled also, although it is not possible to completely halt these. The motor can be protected against overloading by selec- ting a supply voltage that causes the opamp to clip when it outputs the maximum motor current. Finally, ICi is capable of supplying considerable cur- rent, and must, therefore, be fit- ted with a fairly large heat-sink. The quiescent current of the circuit is about SO mA. TW tab connected to pin 3 by P Needham i this is either R 6 , Rs//Rs, or has a diode characteristic, the due to the use of resistors from Re//R 9 . A difficulty arises if it is associated voltage drop is the E12 series. Close tolerance Although the Type 723 voltage | intended to provide an overcur- always lower than that of the is especially important in the regulator has been with us for j rent indication for the shutdown transistor internal to the 723. 5 V range, since the value quite a few years, it is still a i circuit with the aid of an exter- The three output voltages from j shown for R 3 gives a theoretical favourite component for mak- nal transistor fitted in parallel this supply are probably the output of 4.9 V. This can be mg simple and good quality ; onto pins 2 and 3, since the most commonly used for increased readily by fitting a. power supplies. The 723 j external and internal transistor testing asymmetrically fed resistor in parallel with R3, until possesses excellent character- are highly unlikely to have pre- designs: 5 volt for many TTL the output voltage is 5.0 V pre- tties, including a highly stable ! cisely the same characteristics, and CMOS circuits, 9 volt for | cisely. Switches St and S 2 are output voltage, adjustable cur- [ When the internal transistor has battery operated equipment or ' preferably SPDT types with a rent control, and short-circuit the low B-E voltage of the two, logic circuits equipped with centre position, but three-way protection, but it lacks an out- 1 the indication will not work, a 7805 regulator (this requires , rotary switches should also do if put for signalling the activity of j while in the other case the an input of at least 8.5 V), and in both cases the centre contact the built-in current limiter. external transistor takes away 12 volt for RS232 drivers, and , is not used. W The current limiter in the 723 ; the base current for the internal miscellaneous opamp or tran- consists of only one transistor, transistor, so that the current sistor based circuits. The cur- whose base and emitter are limiter is rendered ineffective. rent limiter can be set to 10 mA, brought out to chip pins 2 and 3 In this design of a power ] 100 mA, or 1 A for safely power- respectively. When the voltage supply, a current overload indi- ! ing- experimental circuits, across these pins exceeds 0.5 to cation was realized by fitting Power regulator Ti should be 0 6 V, the transistor is turned on the external transistor with a , fitted with a heat sink sized at and cuts the drive to the output high value base resistor, R?, to | least 10 x 10 cm. LEDs D? transistors. In most applications, ensure that the current limiter ' (green) and Ds (red) are the ~e voltage drop for the B-E in the 723 is not disabled. A power on and current overload -nction of the current sense further transistor, T 3 , has been indicator, respectively, transistor is developed across added to keep the base current The output voltages of the an externally fitted resistor. In for T 2 as low as possible. Since supply may not be as accurate I ~e the supply proposed here, the base-emitter junction then as required, and this is mainly 1 elektor mdia septembef 1987 9-63 I* lllllll *• !“'ll 1 This code lock provides a high degree of security whilst being a very simple design. At the heart of the circuit is the Type 4022 octal counter. In the non- active state, C 2 is charged via Rs, so that the reset (R) input of the counter is kept logic high. This causes output Qo to be actuated, while all other outputs are logic low. When Si is pressed, Ti is switched on via debouncing network R2-C1, and ICi receives a clock pulse. Also, C 2 is discharged via R«-Dv ending the reset state of the counter and enabling it to advance. The time required for Rs to recharge C 2 , i.e., to reset the counter, is the maximum time that can lapse before the next key is pressed. The above cycle is therefore repeated only when S? at the 0’ output is pressed in time. When all keys have been pressed in time and in the correct order, Qr goes high for about 4 seconds to enable driving the unlock cir- cuitry, e.g. a relay driver for an automatic door opener. The code for the lock shown in the circuit diagram is 1704570: this is but an example, however, and the combination code is readily altered by swapping connec- °| “ a code = 1704570 *unlcck © I2 r ~ — 1 A w qoM— rJkTj— 0 - 101 4022 5 S_G 06P— I 4k7 1—0- »n| - 4 — u- tions between the counter out- which makes it possible to add sumption of the code lock is puts and the switches. When two keys. This means that the negligible at 0.5 pA, so that bat- the 7-digit code is considered number of combinations is 10 9 tery operation is feasible. The too simple to crack, the 4022 instead of 10 7 . circuit works well from any can be replaced by a 4017, The quiescent current con- supply between 6 and 15 V. Resistors ( ±5%): Ri;R 3 - 10K R2 = 220K R« = 100R Rs = 1M0 Re. . . R 12 incl. = 4K7 Capacitors: Ci;Cj = 100n Cd = 4p7; 16 V; axial Semiconductors: Di = 1N4148 ICi = 4022 Ti = BS170 Miscellaneous: So . . So incl. = Digitast momentary action button (SE or S version, ITT Schadowl. PCB Type 87463 (not available through the Readers Servicesl 9-64 elektor mdia sepiembet 1987 The accompanying photograph shows that the code lock can be built as a very compact unit thanks to the use of a printed circuit board that holds the 10 keys also. Sv i from an idea by P Needham At the heart of this simple cir- cuit is the well-known Type KTY10 temperature sensor from Siemens. This silicon sensor is essentially a temperature-de- pendent resistor, which is con- nected as one arm in a bridge circuit here. Preset Pi functions to balance the bridge at 0°C. At that temperature, moving coil meter Mi should not deflect, i£., the needle is in the centre position. Temperature vari- ations cause the bridge to be unbalanced, and hence pro- duce a proportional indication on the meter. Calibration at, say, 20°C is carried out with the aid of P 2 . The bridge is fed from a stabil- ized S.l V supply, based on a temperature-compensated zenerdiode. It is also possible to feed the thermometer from a 9 V battery, provided D 1 -D 3 , Ri and Ci are replaced with a Type 78L05 voltage regulator, because this is more economic as regards current consump- tion. TW 86503-1 In addition to the description elsewhere in this magazine of HC and HCT based R-C/L-C escalators for use up to 20 MHz, this design brief concentrates on quartz-controlled oscillators which find applications in digital equipment and micro- processor systems. Such oscillators can only be made with HCU gates, because HC and HCT ones have buffered outputs that make them unsuit- able for use as analogue ampli- fiers. The circuit diagram shows a Pierce oscillator set up around a single gate in a Type RI LiilLI ami ^ rii 5)5V Raj- ► — — s © IC1 © Y — ® — “O— Cl ■■ T - ' B8p -L N1 = 1/6 IC1 = 74HCU04 87407 74HCU04 package The inverter functions as an inverting ampli- fier with a phase shift of 180°. The circuit can be modified into a Collpits oscillator by replacing the quartz crystal with an inductor. It should be noted, however, that the use of a quartz crystal is more appro- priate because it ensures minimum current consumption and adequate suppression of the third harmonic frequency. Finally, Ra must be replaced with a 33p capacitor if the oscil- lator is operated above 4 MHz. St elektor mdia September 1987 9 - 65 w< HHiilliliW HC-BASED OSCILLATORS Two inverters, one resistor and one capacitor are all that is required to make a HC(T> based oscillator that gives reliable operation up to about 10 MHz. This sort of circuit is well-known, and appears in Fig. la. The use of two HC inverters gives fairly good symmetry of the rectangular output signal. In the same circuit, HCT inverters give a duty factor of about 25%, rather than about 50%, since the toggle point of an HC and an HCT inverter is ViVcc, and slightly less than 2 V, respect- ively. When the supply voltage for the oscillator is switched on, C initially has no charge, and the output of Ni and Nz are at the same logic level. Capacitor C is then charged via R, until it has acquired a charge voltage that corresponds to the toggle volt- age, Us, of Ni. Assuming the output of N 2 initially to be logic low, the waveform of the signal at the input of Ni is essentially as shown in Fig. 2. When C is charged up to level 1 , the out- put of Ni toggles, and so does that of N 2 . This causes the voltage at the input of Ni to rise, via C, to about 1.5Vcc, so that C is reverse charged to level 3 . From there on, the amplitude changes in a mirror-inverted way to reach the initial state again (level 5 is identical to 1 ), and the circuit oscillates. In practice, the curve in Fig. 2 is slightly flatter, because the peaks at levels 2 and 4 are clamped to + 5 V and 0 V by the protective circuits internal to the inverters. - If the oscillator is to operate above 10 MHz, the resistor is replaced with a small inductor, as shown in Fig. lb. The output frequency of the cir- cuit in Fig. la is given as about 1/1.8RC, and can be made vari- able by connecting a 100K preset in series with R. The solution adopted for the oscil- lator in Fig. lb is even simpler: C is a 50 pF trimmer capacitor. D On most FM tuners, the stereo indicator lights upon detection of the 19 kHz pilot tone. How- ever, this need not mean that the programme is actually stereophonic, since the pilot tone is often transmitted with mono programmes also. A similar situation exists on stereo amplifiers, where the stereo LED is simply controlled from the mono-stereo switch. The LED-based stereo indicator described here lights only when a true stereo signal is fed to the inputs. Differential ampli- fier Ai raises the difference between the L and R input signals. When these are equal, the output of Ai remains at the same potential as the output of Az, which forms a virtual ground rail at half the supply voltage. When Ai detects a dif- »• 9-66 elektor india September 1987 ilillllllilllillillllllilliliiiiili llllil ill 11 1 :reo indicator , ference between the L and R When building the circuit into input signals, it supplies a an amplifier, care should be positive or negative voltage taken to select the right point with respect to the virtual from which the input signals are ground rail, and so causes Cs to ' obtained. In general, this be charged via Di , or C 4 via Dj. should be before the volume The resistors connected in and balance controls, but parallel to these capacitors behind the mono/stereo selec- ensure slow discharging to tor. The signal level should not bridge brief silent periods in j be less than 100 mV to compen- the programme. Comparator j sate for the drop across Di or A3-A4 switches on the LED j Dz . Also observe that the driver via OR circuit D3-D4. impedance at the selected I ’’tap” location is relatively low. I tion is less than 7 mA when the I Should the stereo light come on J LED is off, and about 20 mA when a mono programme is \ when it is on. being received, the input ! fyy signals are different, and the sensitivity of one of the ampli- fier channels should be altered, j If this is impossible or undesir- 1 able, R 3 may be replaced by a series connected preset and a resistor. The sensitivity of the stereo indicator is adjustable with Pi. The current consump- from an idea by J W v Dijk Many monitor chassis currently offered by computer surplus stores have separate inputs for horizontal and vertical synchronization signals. Most home micros, however, have a composite video output, so that some form of interfacing is required to drive these bargain monitors. The Type TBA950-2 is a sync separator chip which is fre- quently encountered on TV chassis. In its standard appli- cation circuit, it requires to be driven by a flyback signal derived from the output of the me frequency oscillator. With- out this signal, which is applied to pin 10, the sync pulse would end up somewhere among the picture lines. To be able to use the TBA950-2 in the present application, the horizontal pulse is slightly shifted with the ard of a double monostable multivibrator, ICj. The operation of the circuit should be clear from the accompanying timing diagram. The output pulse from the TBA950 is fairly wide (26 (is), and its positive edge triggers the first MMV (01), whose negative output pulse transition m turn triggers the second MMV in the 4538 package. The line sync pulse for the monitor a available positive and nega- tive at IC 2 outputs 02 and Q2, respectively. Adjust the circuit as follows: set Pj to the centre of its travel, and adjust the frequency control, ?- , such that the image is stable. Next, position the image by adjusting P 3 . If the correct pos- 1 ition can not be obtained, the phase control, P 2 , must be care- fully readjusted, followed by Pi . The vertical sync pulse is avail- able at pin 7 of the TBA950-2. Finally, the dashed resistors and diodes are required if the moni- tor inputs are designed to ac- cept signals with a peak-to- peak amplitude of 5 V. D elektor india September 1987 9 - 67 SERIAL DATA CONVERTER Some computers and communi- cation programs are unable to output serial data composed of 7 data bits and a parity bit. The present circuit has been de- signed to output this data format when it is driven with serial data organized as 1 start bit, 8 data bits, no parity bit, and 1 stop bit. This format is widely used for accessing bulletin boards, data banks, and the like with the aid of a modem, and should be available on most computers equipped with an RS232 port. The converter has a built-in clock generator which can be set to the baud rates shown the circuit diagram, Fig. 1. Both odd and even parity can be gener- ated, and no handshaking is required with the computer or console. The basic operation of the con- verter is as follows (also refer to the timing diagram in Fig. 2). The rising edge of the start bit in the incoming 10-bit word clocks bistable FFi, whose out- put 0 goes low and so enables counters ICi, IC 2 a and IC 2 b, which were previously blocked by the high level of RST. Binary counter IC 2 a starts counting the clock pulses provided by baud rate generator ICe. The fre- quency of this clock signal is 16 times the bit rate on the serial input and output line. Bistable FF 2 and counter ICi are clocked with signal CK, whose period corresponds to that of the bits in the data stream. The received start bit and the next seven data bits are passed through FF 2 , while ICi keeps count of the number of transmitted bits, and actuates output 9 during the reception of the ninth bit (i.e., databit 7). The rising edge of the counter output pulse is dif- ferentiated in Cs-Rs and then applied to NAND gates N 1 -N 2 . These make it possible for FF 2 to be set or reset, depending on the state of parity counter ICzb, which keeps count of the logic high bits in the serial word applied to the converter. Its out- put Qa indicates whether the number of detected high bits is odd (Qa=1) or even (Qa=0), and causes FF 2 to toggle when the differentiated pulse from ICi makes the output of Ns or Ns 9-68 elektor india September 1937 1 87S16 I 0 / i START L _n_ go high for a very short period. When Qa is low, the parity bit a: 0 of FF 2 is high because in that case the S (set) input is driven high. Similarly, the parity bit is low when Qa is high because the R (reset) input on FF 2 is then driven high. These two situations can occur when even parity is selected by fitting wire links A-D and B-C as shown in the circuit diagram. Odd parity is obtained by fitting links A-C and B-D, and perma- nently low parity by fitting C-E and D-F (note that a "low" parity evel means that the relevant bit is logic high in the RS232 con- vention). After transmission of the parity bit, the circuit is prepared for the next word by the carry (CY) output of ICi providing a high level to differentiator C 2 -R;. This j 7 resets FFi, which in response ^ drives the RST line high to reset the counters. The convention adopted for the logic high and low levels of the data bits in the proposed con- verter requires that this is inserted in an RS232 or RS432 data line. Line driver N 1 1 may be omitted, and the serial output signal taken from 0 on FF 2 , if the driven input can operate with pulse levels of 0 and +5 V. Finally, Fig. 3 shows a suitable alternative for the crystal-oper- ated clock generator, which may be considered too exten- sive if the circuit is to work at a fixed baudrate of 1200. Multi- turn preset Pi is set for an out- put frequency of 19,200 Hz. Th _ 5V 87516 - 3 / r 1 1 TRANSMISSION LINES iil!il!li! Iiiiiii iiiiiii lli m .ill FOR TTL CIRCUITS Although cable connections between TTL circuits are nor- mally not as critical as those for, say, RF applications, it is still worth while to reflect on this subject because strange things often happen when a TTL trans- mission line is not correctly ter- minated. In particular, this discussion is about terminating coaxial cable and flat ribbon cable. The latter is frequently used for driving Centronics compatible inputs. A commonly used coaxial cable is RG59B/U, which has a characteristic impedance of 75 2 and a propagation delay of 5 ns/m. With signal rise and fall ernes of 4 ns, the cable may be considered electrically long if it exceeds 40 cm. One of the most common terminations used when driving a long coaxial cable with an LSTTL gate is shown in Fig. 1. This set-up is unsuitable for a HCT bus driver, since the termination provides a poor impedance match, and requires a current sinking capability of 20 mA. An improved termination cir- cuit is shown in Fig. 2: this ensures reliable signal trans- mission for cables up to 15 m. Note that the 1 kQ pull-up resistor is only required when the driver is an open collector gate or buffer. Flat ribbon cable often introduces considerable cross- talk between wires, especially when terminated in HC(T) gates, which form a high input impedance. In general, a flat ribbon cable should not be longer than about 60 cm, but longer runs are possible when individual wires are separated by grounded wires (1.8 m max.), or when each wire is termin- ated with a 1 kQ pull-up resistor (1.2 m). A combination of these methods makes it possible to use flat ribbon cables with a length up to 2 m, but this is also attainable without ground wires— see Fig. 3. The com- bined use of this termination network and grounded wires in the flat ribbon cable should enable a cable length of about 5 m. St 1 a. fcl 1 °l I ~ -A 2 A- A nllNr, -A . j. l °w 3 fcl lol ■ ml Q^>°- fcl lol 87455 i>lektor india September 1987 9-69 TUNNEL DIODE BATTERY CHARGER After a longish absence from 1 the semiconductor scene, the tunnel diode, also called Esaki diode, after its Japanese inven- tor, has been re-introduced thanks to the fact that it can be used to save energy. In the nineteen fifties and sixties, tun- nel diodes found many appli- cations mainly in RF circuits, where their rather special properties were exploited for making fast level detectors, oscillators, mixers, and the like As compared with a normal diode, a tunnel type utilizes a semiconductor material with an extremely high doping level, causing the depletion layer between the p-n junction to be roughly a thousand times nar- rower than that in even the fastest of normal silicon diodes. When the tunnel diode is for- ward biased, tunnelling of the electron stream occurs across the p-n junction. Tunnelling in doped semiconductors is a phenomenon not readily ex- plainable on the basis of tra- ditional atomic theory, and can not possibly be expounded in this brief article. When measuring the corre- lation between a tunnel diode's forward voltage, Ur, and cur- rent, If, it will be found that the device possesses a negative resistance characteristic be- tween the peak voltage, Up, and the valley voltage, Uv, as shown in Fig. 1. Thus, if the diode is operated in the shaded section of its 1f-Uf curve, the forward current falls as the voltage rises. The resistance of the diode is undisputedly negative, and usually given als The present design exploits the above property of tunnel diodes by having a series con- nected set of these devices charge a battery with the aid of solar energy. As seen in Fig. 2, seven or more Gallium-Indium Antimonide (GISp) tunnel diodes are series connected and fitted on a large heatsink, which does not serve to dissi- pate their power (tunnel diodes get colder as Uf rises), but to effectively accumulate solar, or otherwise applied, heat, whose energy is converted into a charge current for the NiCd battery. 87465 1 The operating principle of this unique design is remarkably simple. If a normal, pure, resist- ance, R, discharges a battery with current I=V/R, a negative resistance charges the same battery, since the sign of I reverses: — I=V/— R. Similarly, when a normal resistance dissi- pates P=I 2 R watts, a negative resistance delivers this wattage into the load: P=— I 2 — R. When the load is itself a voltage source with fairly low internal resistance, the negative resist- ance must, of course, output a higher voltage for the charge current, Ic, to flow: Ic = <5[ l (Urj-Uba. j / 1 (Ra) + Rb4, From the notation I(Rd) it is immediately seen that all diodes in the series chain must be operated in the — Rd area, because any single diode with a +Rd characteristic would cancel the object effect. To ensure that all diodes exhibit a negative resistance, a simple test circuit can be made as shown in Fig. 3. Note that the meter must be capable of indicating the direction of the current, since it may well hap- pen that a particular diode has such a high Ip:Iv ratio ( tunnel slope) that the battery is unintentionally charged on applying a small forward bias. The test must be carried out at an ambient temperature below 7°C (use a cleared out refriger- ator), and the Uf-If curve is noted for every individual diode by carefully raising the forward bias with the aid of the potentiometer, and recording the resultant values of If, read from the meter. Put an FM radio nearby to make sure that the diode under test does not oscillate at 94.67284 MHz (F res for GISp at doping level 10 7 ). If it does, it is unsuitable for the present application. Establish the range of Uf that ensures — Ra for all diodes. Depending on the production tolerance of the diodes in the available lot, this span may be as restricted elektor india September 1987 4 (TySiDKXle (2^ T unnetdiode &l / 1 130 120 110 0.068 ufd n ^ Mfd 0.068 ufd 0.1 ufd <0.1 ufd M,d ^•>*0.22 ufd '•0.33 ufd. .05 0.20 0.35 1 I ' 0.50 1 f ' 0.65 1 — I — r- 0.80 1 — T - 0.95 CAPACITANCE M F elektor India September 1987 LIGHT-SENSITIVE TRIGGER From an idea by R de Haan This circuit activates a relay upon detecting the absence of light on an LDR (light depen- dent resistor). It is particularly well suited to control outside lighting as used for driveways and garage entrances. Contrary to its normal use as an astable or monostable multivi- brator, the Type 555 IC in this circuit functions as a compara- tor. To explain this rather unusual application, it is nec- cessary to note that the oper- ation of a 555 is normally as follows: the output goes high upon receipt of a trigger (start) pulse on input pin 2. This pulse is a voltage whose level is lower than '/a of the supply voltage. The output goes low again when the voltage at the second input, pin 6, has briefly ex- ceeded % of the supply level. In the present design, the sec- ond input is not used, but the output of the chip can none the less revert to the low state, since pin 6 is connected direct to the positive supply rail. This set-up is accounted for by the ac- companying Table, taken from the 555’s data sheets. In principle, the supply voltage for the circuit must equal the coil voltage of the relay. Do not apply more than 16 V, however, as this may damage the 555. The current consumption of the cir- j cuit is 4 mA, exclusive of the ■ relay, at a supply level of 12 V. Components R 2 and Ci ensure a delay of about 10 s before the relay is energized, so that the circuit is rendered insensitive to rapid changes in the light intensity. Basically, the circuit has no hysteresis effect. However, when the supply is not regu- lated, the actuation of the relay will lower the supply level somewhat. This lowers the internal threshold of the IC, since the trigger point is de- fined as % of the supply level (pin 2). Therefore, the hysteresis of the circuit can be dimen- sioned as required by fitting a resistor in series with the supply. It is also possible to fit a resistor between pins 5 and 7 of the 555, as shown in the circuit diagram. The amount of hyster- esis is inversely proportional to the value of the resistor, and 100K is a reasonable starting point for experiments. The sensitivity of the trigger cir- cuit can be controlled if Ri is replaced with a 1M0 poten- tiometer or preset. W NE555 FUNCTION TABLE RESEf 141 TRIGGER VOLTAGE (2) THRESHOLD VOLTAGE (61 OUTPUT (31- DISCHARGE SWITCH Low Irrelevant Irrelevant Low On High < Vs VOO Irrelevant High Off High > 'h Voo > % Vdd Low On High > 'h Voo < % Voo As previously established i * >2 ■1 BUS FOR DIRECTION ADD-ON MSX EXTENSIONS The majority of MSX computers do not require a BUSDIR (bus direction) signal from add-on circuits plugged into slots. A problem arises, however, if the extension circuits published in Elektor Electronics are used in conjunction with, for example, a Sanyo MSX machine, which has a few peculiarities in its exter- nal I/O concept. In general, the more slots on an MSX com- puter, the higher the prob- ability that either one of, or both, these circuits are re- quired to be able to use the home-made extensions. Two solutions are offered to provide for the BUSDIR signal. One is usable for the Universal I/O Bus and the I/O & Timer Cartridge, the other for the Car- tridge Busboard. Each of these circuits consists of one IC only. Circuit A is used with the two I/O extensions, and is readily incorporated in the computer, at a suitable location near the slot that receives the extension. If necessary, all slots on the computer are fitted with this cir- cuit, but this makes it imposs- ible to uti lize cart ridges that do supply a BUSDIR pulse, unless Si is included to disconnect the output of N» from slot pin 10. Note, however, that this switch 1 must not be operated when the elektor mdta September 1987 9 - 79 A B /T1 BC547 87430 computer is on. As I/O range 40h-FFh is re- served for the computer-resi- dent hardware, address lines As and A 7 must be low for the selection of external I/O cir- cuitry. Moreover, IOREQ and RD must be low to ensure that BUSDIR is only active when the CPU reads data from an I/O device. Interrupts from an exter- nal device can only be pro- cessed correctly when BUSDIR is low in response to MT and IOREQ being low also. This requires an OR function for logic low levels: BUSDIR = Ml IOREQ + IOREQ RD A7 A 6 If you are hesitant about open- ing the computer to install cir- cuit A, you may consider the use of a part of the EPROM car- tridge board to hold the 74HCT32 as shown in the ac- companying photograph. Note that the 50-way track connector plugs straight into a computer slot, and that a slot connector is fitted at the other side of the ”adaptor-PCB" to receive car- tridges. Circuit B is intended for use on the Cartridge Busboard. Its function is to pass BUSDIR pulses from cartridges to the computer. To this end, it is necessary to first break the interconnecting tracks between slot pin s 10 so as to make all car- tridge BUSDIR outputs separ- ately available for wiring to 8 -input NAND gate Ns. Inverter Ti turns this simple add-on unit into an 8 -input OR gate for logic low levels. The collector of this transistor is wired to pin 10 of Ks on the busboard. It may well happen that both cir- cuit A and B are required for a specific I/O arrangement. In that case, it is suggested to fit circuit A on one slot of the Car- tridge Busboard, and conse- quently use only that slot for external I/O. Pin 8 of N a is then connected direct to the relevant input of Ns. Note: articles in the series MSX Extensions were published in the following issues of Elektor Electronics: January 1986, February 1986, March 1986, January 1987, March 1987, April 1987. 8-CHANNEL VOLTAGE DISPL Simultaneously monitoring the trends of 8 slowly varying voltages is normally very diffi- cult, if not impossible, even with the aid of 8 analogue or digital voltmeters. This circuit turns a common oscilloscope into a versatile 8-channel display for direct voltages. The trend of each of the 8 input levels is readily observed, albeit that the attainable resolution is not very high. The circuit diagram shows the use of an 8-channel analogue multiplexer, ICi, which is the electronic version of an 8-way rotary switch with contacts X 0 - X7 and pole Y. The relevant 9-80 channel is selected by applying an binary code to the A-B-C inputs. Example: binary code 011 (A-B-C) enables channel 7 (Xe -> Y). The A-B-C inputs of ICi are driven from three suc- cessive outputs of binary counter ICa, which is set to oscillate at about 50 kHz with the aid of Pl As the counter is not reset, the binary state of out- puts Qs, Os and Q? steps from 0 to 7 in a cyclic manner. Each of the direct voltages at input ter- minals 1 to 8 is therefore briefly connected to the Y input of the oscilloscope. All eight input levels can be seen simul- taneously by setting the time- elektor mdta September 1987 i” .... |L. ^ h- rB r ! I' 1 -J r’% 1 . .' M oil © N1...N4 = ICI = 74HCT32 (J { qS2 Vi (?) L. base of the scope in accord- ance with the time it takes the counter to output states 0 through 7 on the Qs-Qs-Q? out- puts. The correct starting time for the oscilloscope trace is ensured by using the Qe output of the counter to supply the trig- ger pulse. Diodes Di and D 2 provide for some space between adjacent bars on the display, and create a horizontal reference line. The timebase on the scope should be set to 0.5 ms/div, and triggering should occur on the positive edge of the external signal. Set the vertical sensi- tivity to 1 V/div. The input range of this circuit is from —4 V to +4 V, and connected channels are terminated in about 100K. Adjusting the 8-channel voltage display is straightforward. Simply select the previously mentioned scope settings, and adjust Pi to make all 8 channels [ visible over the full width of the scope screen— see the accom- panying photograph. [ The circuit has a modest cur- rent demand of less than 5 mA from a simple +5V supply, or from two 4.5 V flatpack bat- teries. Th “3 IIIIIIIIF ,Jf TRACI ONG WINDOW 1 ill i* hi COMP •ARATOR by H Gulitz The use of comparator circuits in many different appearances and practical realizations is common in a wide variety of electronic control and measure- ment systems. Usually, the volt- age from a sensor device is fed to a comparator which, as its name implies, compares the measured level, Um, with a fixed reference, Uret, and pro- duces a negative output (0) or positive output (1) when UinUret, respect- ively. A window comparator can be made by connecting two comparators with different reference levels, which define the upper and lower limit of the switching range. In practice, these references are usually adjusted with presets to dimension the win- dow as required. This arrange- ment makes it impossible, however, to automatically shift the window up or down in ac- cordance with, say, ambient light conditions to be measured with a light dependent resistor. This circuit has no fixed I threshold levels, but derives its reference from the measured signal, so that slow changes in this cause the window to track along. Capacitors Ci at the inverting input of Ai, and C 2 at the non- inverting input of A 2 store the input voltage. When the voltage at the non-inverting input of Ai rises, this opamp toggles. The eleklor india September 1987 9 8 1 associated inverting input lags this change because of the r delay introduced by the capaci- tor. LED Di lights. The process is similar in the A 2 section of the circuit when the input voltage drops. This is indicated by LED D 2 lighting. Diodes D 3 and D 4 form an OR function to actuate a simple relay driver set up with Ti. The relay is energized when the cir- cuit detects a fast change in the input voltage. The ability of the circuit to accept a variable input voltage makes it suitable for use in burglar alarms— see Fig. lb. Several break contact arrangements Rt3-Si-Ri» may be connected in series and to the input of the window compara- tor. Alarm relay Rei is activated when either Si is opened or Si-Rk is bypassed. To prevent burglars from fooling the alarm, Rt4 must be fitted into Si, because no alarm signal is given when only Si is shorted. The sensitivity of the tracking window comparator is defined by the ratios R2/R3 and Rs/Rs. The relevant component values indicated in the circuit diagram give 1:100 ratio, so that, for example, a fast change of 30 mV is detected when the input voltage is 3 V. The sensitivity also depends on the input voltage. Although the circuit can in principle handle any input between 0 V and the supply level, the ICs used give reliable operation only when driven between 1 and Ub— 1 volt. The tracking window compara- tor is preferably fed with a supply between S and IS V. Its current consumption, inclusive of the LEDs but exclusive of the relay, is 10 mA maximum (note that the relay can be fed sep- arately). W ! W ii LOW VOLTAGE ■ii r / J m Jill DROP REGULATORS The fast spreading incorpor- ation of CMOS, HC and HCT chips has created a need for voltage regulators with a very low internal drop to enable powering CMOS-based equip- ment from a set of batteries delivering 6 V. The recently introduced Types LP2951 and LP2950 from National Semicon- ductor are micropower voltage regulators with a variable out- put voltage of 1.24-29 V and a fixed output voltage of 5 V, respectively. The former features an internal voltage div- ider with a 5 V tap bonded out to a pin, a logic compatible shutdown input, and an open- collector ERROR output which warns of a low output voltage, often due to an insufficient bat- tery voltage at the input. The ERROR output is extremely useful for an early warning system that arranges for a microprocessor to be reset properly before the supply voltage falls to a level that would upset the operation of the system it controls. The voltage drop across the LP2951 is only 0.4 V at a load current of 100 mA, so a 6 V bat- tery pack can be used to power a S V circuit. The quiescent cur- rent drain of the regulator is about 12 mA at an output cur- rent of 100 mA. This is fairly high as compared with a con- ventional regulator from the 78XX family, and mainly due to the internal series regulator transistor being driven into saturation, which causes it to have a relatively low current amplification factor (the base 9-82 current flows into the ground return line, instead of into the output load, as with the typical 78XX regulator). The application circuit shown in Fig. la should be fed from an input voltage of more than S.4 V, while its maximum output cur- rent is 100 mA. Note that both the LP29S0 and LP2951 feature internal current and thermal limiting circuits. The decoup- ling capacitor at the output of the regulator should be a good quality tantalum type, fitted as close as possible to pins 1 and 4. At relatively low output cur- rents, less capacitance is required in this location. For currents below 10 mA, 0.33 piF is satisfactory, while the minimum value is 0.1 jiF for currents below 1 mA. These values ap- ply to an output voltage of S V; for lower voltages, more output capacitance is needed. The circuit in Fig. lb is a 2 A low dropout regulator based on the LP2951, The output voltage is calculated from Vo=(1+Ra/Rb)1.23V where 1.23 stands for the volt- age at the feedback input, pin 7. For an output of 5 V, Ra and Re may be omitted, and the feedback input pin 7 can be connected direct to the S V tap (pin 6) output. The sense input, pin 2, is then connected to the Vo rail. In this application, Vm must be at least 0.S V higher than Vo. National Semiconductor appli- cations. R elector india September 1987 N» This design answers the need for an inexpensive, yet good quality, preamplifier equipped with a tone control section. Fig. 1 shows the circuit dia- gram. The amplification of the input stage set up around opamp Ai is adjustable between 10 and 20 with preset iPt The 0 dB level at the input is 50 mV, while the input im- pedance and capacitance are 47 KQ and 47 pF, respectively to enable ready connection of most record players and cassette decks. The tone con- trol section is a standard Baxan- dall type with P3 and P< as the respective bass and treble con- trols. The gain vs frequency curves for various settings of the tone controls appear in Fig. 2. Here the 0 dB level cor- responds to 1 V. The current consumption of this preamplifier is modest at about 5 mA. When the circuit is cor- rectly balanced, the indicated measuring points should all be very nearly at ground potential. The circuit shown here must, of course, be duplicated to obtain a stereo preamplifier. St FULL CUT '/I CUT 1 This application of the well- known Type 555 timer is in- tended for model railway en- thusiasts wishing to construct a two-lamp flashing beacon with a minimum of components. With reference to the circuit diagram, the number of LEDs need not be restricted to two: several may be connected in parallel to achieve a higher light intensity, but a total current con- sumption of 200 mA should not be exceeded to prevent the destruction of the output stages in the 555. Each LED added sv IS SO mA should have its own current limiting resistor, similar to D1-R3 or D2-R4. The flashing rate is defined with Cl The stated value of this com- ponent is likely to be optimum for applications in model rail- ways. The supply voltage for the circuit is not critical, but should remain within the range from 5 to 10 V. With two LEDs fitted and a 5V supply, the flashing circuit should con- sume less than 50 mA. The in- tensity of the LEDs can be adapted to individual pre- ference by changing Rs and R4, but too low resistance values should be avoided to prevent the destruction of the LEDs. St elekior iivdia September 1987 9-83 PRECISION This precision rectifier oper- ates from an asymmetrical supply, handles input signals up to 3 Vpp and has a frequency range that extends from DC to about 2 kHz. Its amplification is unity, and depends mainly on the ratio R4/R3. Opamp Ai is connected as a voltage ampli- fier (Ao = 1), A2 as an inverting amplifier (Ao= — 1). Opamp A2, transistor Ti and diode D2 ensure that the output voltage, U2, is identical to the positive excursions of the input voltage, Ui. When Ui is positive, the out- put of Ai is held low at about 0.25 V, so that T 2 is disabled and can not affect the rectified out- put signal. Components R 2 and Di pro- tect the pnp input stage in A 2 against negative voltages, which are effectively limited to —0.6 V. For negative excursions of the input signal, the function of Ai, T 2 and D 2 is similar to the previously mentioned compo- This is a downright simple de- sign for an AF function gener- ator that supplies a rectangular and triangular signal, and can be fed from a single 9 V supply. The signal generator proper is a Type TLC272 dual CMOS opamp from Texas Instruments. This chip is remarkable for its low current consumption and wide operating range. The circuit is essentially com- posed of two functional parts. Opamp Ai is connected to func- tion as a Schmitt-trigger whose toggle point is set to 4.5 V, while A 2 is an integrator that converts the rectangular signal from Ai into a triangular waveform. The oscillation frequency of the circuit is fixed solely by the ratio R/C and can be calculated from f 0 = R2/4RRiC. Resistor R may be replaced by the combi- nation of the 10K resistor and 100K potentiometer as shown to 9-84 elektor india September 1987 nents. The peak output voltage of the rectifier circuit is deter- mined mainly by the maximum output swing of the opamps and the voltage drop across the transistors plus D 2 : this amounts to about 3 V in all. When the circuit is not driven, it consumes about 1 mA, and is therefore eminently suitable for building into portable, battery- operated equipment. Sv iiiiiiiiiiiiiiiiiiiiifiiiiiiiiuaiiiiiii m it i! Hi llllllllll! 2 mA 9V * effect continuous adjustment of the output frequency within the AF signal band. The generator should not be terminated in less than 10K. St o— i a i I \ i — t — o 100k \ 1 | |- IC1 = TCA280 A The circuit proposed here is suitable for fitting into slide pro- jectors without a dimmer fa- cility (24 V AC fed halogen lamps). With a few small alter- ations, it can also be used for dimming 12 V halogen lamps, but not those in a car, because these are fed from a DC source. The circuit shown in Fig. 1 is intended for operation from a 24 V AC supply, and can handle a lamp load of up to 150 W. For loads up to 250 W, the TIC236 should be replaced by a TIC246. The illumination of the halogen lamp is controlled by applying a direct voltage to pin 5 of dim- mer chip ICi. A voltage of +2.5 V gives maximum illumi- nation, while + 5 V results in the lamp being turned off com- pletely. The lamp intensity con- trol range— 2.5 V to 5 V— can be extended upwards by de- creasing the value of C 2 . The TIC246 should be used when the circuit is to control a 12 V lamp that consumes more than 50 W. Figure 2 shows details of the connection of a potentiometer to the intensity control input of the TCA280A. Voltage divider R 10 -P 1 -R 11 is fit- ted externally and can be fed from the stabilized voltage available at pin 11 of ICi The minimum and maximum inten- sity of the lamp are determined by Rio and Rn, respectively, so that the control range can be dimensioned to individual pre- ference. When potentiometer control is used, C 2 must always ! be lOOn. Figure 3 shows the signal wave- forms at various points in the circuit. The halogen lamp dimmer is constructed on a printed circuit board as shown in Fig. 4. When the lamp power is greater than 15 W, the triac should be fitted onto a heatsink, and the tracks to the al and a2 terminals should either be covered with solder, or strengthened with short lengths of wire. R Parts list Resistors I ± 5%): Rt - 470R; 0.5 W Rj;fl7= 100K Ra = 22K R< = 330K Rs= 150K Re - 270K R« = 82K Re - 150R Capacitors: Ci = 470p; 16 V; axial Ci- Ip. 16 V; axial 1 Ca - 1 n5 Semiconductors: Di - 1N40O1 Tri. TIC236 or TIC246* 1C i = TCA290A Miscellaneous: PCB Type 87452 (see Readers Services). Heatsink for Trie. ' See text elrkior india September 1987 9-85 COMPRESSOR by S G Dimitriou This versatile circuit serves to raise the average output power of an AF amplifier. Its simplicity makes it suitable for appli- cations in intercom systems, public address and disco- theque equipment, and also in various types of transmitter. Compression of music and speech essentially entails reducing to some extent the dynamic range of the AF input spectrum in order to drive an AF power amplifier with a fairly steady signal level just below the overload margin, thus increasing the average output power of the system. However, some distortion is inevitably incurred in the process of amplifying the relatively quiet input sounds and attenuating the louder sounds. It is evident, therefore, that the control of the amplifier/attenuator function in the compressor determines to a large extent just how much dis- tortion is introduced by the circuit. Before inserting any type of compressor in an AF signal path, due consideration should be given to the attack time i.e., the time it takes the circuit to detect and counteract a sudden increase in the amplitude of the incoming signal. Allowing for persona] preference and the character of the input signal (speech, popular music, etc.), the attack time of a compressor generally lies in the range from 0.5 to 5 ms. The release time of the compressor is the time it takes the circuit to return to the settings that existed before the rise in amplitude occurred. Contrary to the attack time, the release time is usually of the order of seconds. If it is made too short, the compressor’s attenuating action may cause interference with the lowest components in the frequency spectrum. On the other hand, too long a release time (10-15 s) is also undesirable as this will give rise to an unrealistic and unpleasant effect caused by the output sound remaining com- pletely muted long after the increase in input signal ampli- tude. In practice, the release time of a compressor will need to be adapted to meet the de- mand of the particular input signal; speech generally re- quires a longer release time than music. Some compressors have a provision for the setting of the release time, but the one proposed here is an auto- ranging type, that is, it arranges for the release time to change automatically with the instan- taneous amplitude of the input signal. Figure 1 shows the circuit diagram of this compressor. Despite its simplicity, the design responds adequately to a good number of contradicting requirements. As to its dynamic characteristics, an input signal change from 25 mV PP to 20 V PP (=58 dB) is compressed into an output signal change from 1.5 V PP to 3.4 V PP (=7.1 dB). For a less extreme signal change, e.g., from 25 mV PP to 2.5 V PP Measurement values: A = 0 V B = +4.5 V C = 6 mA D = 3.9 V All values are typical and within 10%. All voltages measured with respect to ground with a DMM IZm = IMP). (-40 dB), the compressed out- put signal changes from 1.5 V PP to 2.25 V PP (=3.5 dB). The cir- cuit has an extended frequency response from about 20 Hz to 40 kHz nominally, thanks to the use of a fast opamp, the Type LF357 (ICi), which is set up here to provide an amplification of about 471 }(Rs + Rs)/Rs]. Capaci- tor Cj blocks the .direct voltage at the inverting input of ICi, and with R5 sets the low-frequency roll-off of the opamp alone at about 16 Hz. Resistors R 3 and R* bias the non- inverting input of the opamp— and hence its output— at half the supply voltage, ensuring opti- mum linearity. Capacitor C 2 feeds the input signal to the opamp while blocking the bias j voltage at pin 3. Its value is not critical, but it has some effect ; on the low-frequency response J of the compressor. The attenu- j ator section in this circuit is essentially composed of R 2 and Tt. The collector of this transis- tor is held at 0 V with the aid of Ri and R 2 . In this way, Ti is always operated in its saturation region, and its collector-emitter junction acts as a variable resist- ance controlled with the cur- rent fed to the base. The higher this current, the lower the c-e resistance, and the higher the instantaneous attenuation of the signal fed to ICi. The control- ling rectifier is composed of D1-C5-R7. Transistor T 2 functions to provide the charge current ! for C 7 so as to avoid distortion : otherwise incurred by too heavily loading the ICi output. The rectified voltage across Cs is a direct measure of the output ! signal amplitude, and forward- biases the base of Ti, which j regulates the attenuation as dis- 9-86 elektor mdia September 1987 cussed. The use of a diode with a low internal resistance, Di, and a buffer, T 2 , ensures fast charging and slow discharging of C 5 , and thus a short attack time and a long release time, respectively. As Cs is dis- charged via R ? and the base resistance of Ti, the release time of the compressor is the product of the value of these three components. When the base bias is reduced, the base resistance of Ti increases, lenghtening the release time. This is a most welcome feature, especially with speech signals. The output of the opamp is fed A digital-to-analogue converter (DAC) that is easy to build from a handful of readily available parts. The 8-bit digital input for the circuit is applied to resistors Ri?-R 24 inch, each of which drives an associated current source composed of two series- connected diodes, a transistor and a current defining resistor fed from the positive supply rail. A logic high level at the input causes the relevant cur- rent source to be switched on, a logic low level switches it off. The sum of currents from Ti-Ta inch is arranged to pass through preset Pi, which thus drops a voltage Uo in accordance with to Ci-Pi-Rio, which provide DC insulation and level adjustment. Two compressors are readily combined to make a stereo ver- sion by- feeding them from a common battery and connect- ing points X and points Y (never X to Y!). In this case, Ti and Di in both compressors must be matched types to ensure proper operation.. Figure 2 shows two simple test circuits for selecting transistors and diodes with matching DC characteristics. The basic method is to start with noting the voltmeter reading for a par- ticular device, and then find a the magnitude of the 8-bit word written to the circuit. The current supplied by each current source is about 700/Rx [mAj, where Rx is the value of the associated resistor between the emitter and the +V rail. In order to ensure satisfactory linearity of the analogue output voltage, resistors Ri-R 8 inch must be dimensioned to obtain a current ratio of 1:2 between any two adjacent sources. In practice, it is wise to first apply a logic high voltage to the MSB (most significant bit) input of the circuit, leaving the remaining inputs low, and measure Uo with the aid of a good-quality matching type from an available lot by inserting devices until one is found that gives the previously noted test voltage. In the diode test circuit, the LED lights to indicate the absence or reverse connection of a diode under test. Provision has been made to use the circuit as a noise sup- pressor. Referring to Fig. 1, closing Si connects Cs across the regulator transistor to form a low-pass filter in conjunction with Ri and R 2 . The cut-off fre- quency of this LPF is a function of the current sent into the base of Ti. The overall effect thus voltmeter. Next, drive De high and all other inputs low, and make sure that Uo drops to half the previously obtained level by dimensioning R? as required. The other current determining resistors are similarly estab- lished; the value of Ri-R 8 incl. that gives the correct level of Uo is obtained by making suitable combinations of series and/or parallel connected high stability resistors. Alternatively, it is possible to use multi-turn presets. As all resistors R 1 -R 7 incl. must be dimensioned starting from a particular value of Rs, this resistor must first be calculated considering that the obtained is an effective elimin- ation of noise from quiet passages in the programme. For louder passages, the suppres- sion of noise is not so important, as it is then virtually inaudible. Finally, when using this com- pressor, make sure that your amplifier has ample cooling provision, because it may well be continuously operated at the top of its power rating. For the same reason, check whether the loudspeakers can handle the available power. Sv output linearity of the circuit is affected unless 1.4Pi/Rs 2R; 1KQ < R < IMS; C > 10 nF. With Rs and R calculated for a given frequency and value of C, both resistors can be realized as presets to enable precise set- ting of the output frequency and the duty factor. Do not forget, however, to fit small series resistors in series with the presets, in observance of the minimum values for R and Rs as given in the design equa- tions. The values quoted for Ic i are only valid if the inputs of the remaining gates are grounded. Source: Philips CMOS Designers Guide, January 1986, p. 105 ff. St __ III 1 , DIGITAL v ill 1 / 1 Jr .,1111 AUDIO SELECTOR by R Shankar Switching audio signals digi- tally could be done with the aid of CMOS analogue switches or multiplexers. Simple as this may seem, there is, however, an inevitable loss in the quality of the sound due to the noisy nature of CMOS switches. Furthermore, the high on- resistance of these devices together with the large parasitic capacitances generally present in CMOS circuits causes a high susceptibility to crosstalk. The circuit given here is a novel way of selecting one out of ten audio signals digitally without any of the foregoing drawbacks. As shown in the circuit diagram, the ten input signals numbered 1-10 are applied to the bases of transistors T,-T,o via capacitors C,-C,o respect- ively. The bias voltages for the transistors are obtained with the aid of Ri-Rio. Depending on the binary state applied to ICi, one of its outputs Q 0 -Q 9 goes low. For example, if the input code is 0010 O 2 goes low, pulling the base of T 3 to 0 V, while the bases of all other transistors are raised to nearly + 15 V. There- fore, T 3 works as an emitter follower while the other tran- sistors are effectively reverse isv 33 nA biased. The output rail of the transistor array is connected to voltage follower IC2, which provides the output signal of the digital audio selector. Voltage regulator IC3 is required only if a +5 V rail is not available. If the number of channels required for a par- ticular application is less than 10, the relevant components can be omitted. If a mute facility is required, simply short one input to ground to silence the output on selection of the cor- responding channel. This circuit can handle input signals up to 4 Vm,s. The total distortion does not exceed 0.01% for frequencies up to 20 kHz. The crosstalk incurred in this circuit is less than —80 dB. This value can be attained by paying due attention to the layout of the practical cir- cuit, the decoupling of the supply lines (fit C11 and Cis direct to the relevant pins of the opamp), and the use of good quality components. The measuring values indi- cated in the circuit diagram were obtained in a prototype. All voltages are measured with respect to ground with the aid of a DMM (Zm = 1M0). The chan- ti^i ■sihexnfeti -was iiuinuti '1. W elektor india September 1987 9 - 89 97443 67 by P Techer The speed and direction of rotation of a motor in a radio controlled model aeroplane or boat is generally controlled by pulse width modulation of the supply voltage to the motor driver stage. In the present circuit, shown in Fig. 1, bistable FFi is set up rather unconventionally to func- tion as a monostable multivi- brator, whose period is set with R 1 -C 1 -P 1 . This period deter- mines the toggle point at which the motor s direction of rotation is reversed. Output 0 of FF* goes high when the pulse at the D input (PWM signal) is shorter than that at the CLK input (signal from FFi). This causes Tt to actuate Rei, so that the motor direction is reversed. The PWM control signal applied to the cir- cuit is also fed to N 2 , whose out- put pulse width is the differ- ence between that of the input signal and that from FFi The pulse width at the output of N 2 therefore increases as the rel- evant control handle on the transmitter is moved further towards either extreme, and is maximum when the handle is in the central position. The output of N 2 is integrated by At to obtain an output voltage pro- portional to the pulse width. As compares this output voltage with the triangular signal at the wiper of P 3 , so that a variable duty factor signal is obtained for driving the power output stage comprised of T4-T5. Meanwhile, A 2 compares the proportional voltage from At to the level set with P 2 . When the output of At is lower than the threshold, i.e„ when the motor speed exceeds the preset level, T 2 activates Re 2 . This causes the collector-emitter junction of series regulator Ts to be by- passed by the relay contact, and so enables the motor to run at full speed, because the forward drop across Ts is eliminated. The frequency of the triangular signal from A 3 is of the order of 2 kHz. which is suitable for most motors. Capacitor C6 may be increased to lower the frequency for non-standard 9-90 SPEED CONTROL FOR R/C MODELS A 1...A4 - TC1 = LT4W4 FF1.FF2= (C2 CD 40 13 NT N3 - 4/4 1C 3 - CD 4070 T3 BC 547 a] motors. Conversely, if the fre- quency is increased, care should be taken to observe the maximum switching speed of Ts, which is a commonly available, but relatively slow power transistor. Presets P 4 and P 3 determine the limits of the inoperative range of the handle, and the point that corresponds to maximum motor ficiently large control range for the handle, and also to avoid the risk of Re 2 clattering or being blocked. Be sure to fit the 470n capacitor across the motor terminals, and the 47n capacitor between one of these and the motor body- see Fig. 2. The coil voltage of the relays should be equal to the voltage for the battery that powers the motor, while the contacts must be capable of handling the current demand of the motor. Transistors T 4 and Ts should be fitted with a heatsink. Note that although the Type 2N305S can handle currents up to 10 A, it may be a good idea to fit two in parallel with 0R1 emit- ter resistors for equal current distribution when heavy loads are to be controlled. The cur- 87426 2 respectively. More specifically, P 3 sets the amplitude of the triangular signal, while P 4 sets the offset level, to enable Ai to output the triangular wave undistorted and with the maximum possible voltage swing. Preset P 2 is used to define the point at which the motor is switched to full speed. Some care should be taken in this setting to allow a suf- rent rating of Db and D? must also be observed: for the stated lN5401s, If(nwx)= 3 A, and two may have to be connected in parallel when this current is approached. Finally, U+ is the model's battery voltage (4.8 V), and +Ucc is the supply voltage for the motor. Pi 1 H — . 1 w r — elekior india September 1987 SYNTHESIZER FOR SW RECEIVER The synthesizer shown in Fig. 1 I is computer controlled, and outputs a local oscillator signal (LO) between 48 and 78 MHz for driving the mixer in the SW receiver proposed on page 00. The circuit is based on the Type MC145156 synthesizer from Motorola. This IC is rela- tively inexpensive, and ensures good LO suppression in the receiver when used in combi- nation with a good mixer. Also of interest is its serial control input, which enables the output frequency to be programmed from a computer. The internal reference fre- quency, 1200 Hz, is obtained by dividing the signal from oscil- lator Ts-Te by 2048. The DAC connected to the output of the first LO gives a resolution of 1200/255=5 Hz. The divider I composed of ICi, IC2, IC3 and Ni has a prescale factor of debtor India September 1987 9 - 91 Last data bit in (Bit no. 16) First bit in (Bit no 1) ‘ 128/129. Opamp ICe is connec- ted as a simple loop filter with a reference signal rejection of about 60 dR An alternative filter that ensures a rejection of 80 dB, but has a slightly longer settling time, is shown in Fig. 2. This circuit is driven from the phase detector output of the synthesizer chip. Opamp ICi is used in a speed-up circuit that may be included to equal the settling time of the filter with ICe. Diodes D 1 -D 2 also serve to shorten the lock-in period of the synthesizer. The use of the Type E420 (Ti) is not obligatory: other types of AF double FET should also work in this appli- cation. The power supply for the synthesizer is shown in Fig. 3. The L-C filter in the +S V rail suppresses noise on the synthesizer supply, and D 2 has been included to compensate for the drop across choke L 2 . The data format for program- ming the MVC14S156 is shown in Fig. 4. Bits SWi and SW 2 con- trol the switching outputs, and are not used here. The syn- thesizer divides by 128N+A: when counter A reaches state 127, N is increased by 1, and A becomes 0. Data is latched into the synthesizer on the trailing edge of the clock signal. When the control word is complete, the enable signal is briefly made high to transfer the data from the shift register to the programmable dividers. The squelch is then enabled to sup- press locking and tuning noises. The construction of this syn- thesizer requires some experi- ence in building RF circuits. The ECL dividers and the syn- thesizer chip should lie upside down on an unetched board to enable effective grounding and cooling. The chips are intercon- nected with the shortest poss- ible wires. Great care should be taken in the construction of the VCO and the TXO. These sec- tions should be screened and built such that mechanical stab- ility is ensured at all times. VCO inductor Li is especially critical in this respect: make sure that the wire turns are secure on the core. Finally, the winding data for the home-made inductors in this circuit: (use enamelled copper wire): Li (VCO): 14 turns 22SWG (0 0.8 mm) on a T50-12 core, tap at 4 turns from ground; Lj (+5 V rail): 8 turns 30 SWG (0 0.3 mm) through a ferrite bead. B 4-WAY DAC EXTENSION This extension circuit makes it possible to use a single DAC (digital-analogue converter) for generating four analogue volt- ages. Evidently, the cost of the extension described here is only a fraction of that of four DAC chips. The operation of the 4-way DAC is fairly simple. Assuming that inputs A, B and E of multi- plexer/demultiplexer ICi are driven low, the output of Ai is fed to the + input of A 2 , while the output of this opamp is con- nected to the — input of At via the demultiplexer and Ri. Capacitor C 2 functions as a storage device. The output volt- age available at terminal 1 equals Udac because Ai is dimensioned for unity gain. When the E input is driven high, or when a new code is applied to inputs A-R the input voltage for Aj is derived from Cz, so that the programmed voltage remains available at the output. The function of the other output buffers and capacitors is, of course, similar to that of A 2 -C 2 . For optimum performance, C2-C5 should be low leakage capacitors, e.g. multilayer MKT, and the input current to A2-A5 9 92 elektor mdia September 19b/ AI = IC2 = TLC271 A2, A3 = 1C 3 = TLC 272 * 87453 A4, AS = 1C 4 = TLC 272 I should remain low. The latter | condition is satisfied by using opamps with FET inputs (typi- cal bias current: 1 pA). Only At requires an offset compen- sation since feedback is pro- vided via the lower multiplexer in ICi. The E (enable) input serves to disable ICi during switchover to another channel. R; then gives Ai unity gain to prevent the — input being left open. When a Type HCT4052 is used | in the ICi position, standard TTL levels can be used to drive inputs A, B and E. A "normal” 1 CMOS 4052 requires 5K6 pull- up resistors to be fitted on these inputs, but only if TTL signals , are used to drive the extension. j The current consumption of the | circuit is less than 10 mA. Udac should be between —3.5 V and 1 +3.5 V. Th 70 VARIABLE WIEN BRIDGE OSCILLATOR A Wien bridge oscillator can be made variable by using two frequency determining parts that are varied simultaneously at high tracking accuracy. High- quality tracking potentiometers or variable capacitors are, how- ever, expensive and difficult to obtain. To avoid having to use such a component, this oscil- lator was designed to operate with a single potentiometer. The output frequency, fo, is calculated from fo=l/(2nRCV r a) where R=R 2 = R3 = R4 = R6, C = Ci = C 2 , and o=(Pi+Ri)R. Preset P 2 allows adjusting the overall amplification such that the output signal has a reason- ably stable amplitude (3.5 V PP max.) over the entire frequency range. The stated components allow the frequency to be adjusted between 350 Hz and 3.5 kHz. Other frequency ranges are readily defined with the aid of the above formula, although it should be noted that the upper Parts list Resistors I + 5%): Ri = 10K Ri;Ri;R4;R« - 100K Rs - 2M2 Pi - 1M0 linear potentiometer P 2 = 5k0 preset Capacitors: Ci;C 2 = 1n5 C3.C4 = lOOn Semiconductors: Di;D2 = 1N4148 ICi = TLC272 or TL072 or OP 221 Miscellaneous: PCS Type 87441 Isee Readers Servicesl . frequency limit is determined mainly by the gain-bandwidth product of the opamps Type OP-221 and TLC272. The current consumption of the oscillator depends on the type of opamp used. The following values were measured: OP-221: 0.5 mA; TLC272: 2 mA; TL072: 2 mA. [ The construction of the oscil- | lator should present very few I problems since a ready-made 2 j circuit board is available through our Readers Services. ! Sv (PMI Application) elcktor mriia seplembef 1987 9 - 93 In many older types of SW receiver, intermodulation in the mixer was generally avoided by including a tuneable, often automatically tracking, pre- selector. In a computercon- trolled preselector, the use of varactor diodes for tuning the inductors often leads to con- siderable intermodulation dis- tortion. A different approach is therefore used in this design. The circuit diagram shows the use of PIN diodes Type BA244 for selecting one of 5 band- filters followed by a low-pass section. Selection of a filter is effected by having the com- puter drive the associated input high. An impedance trans- former is provided at the input to enable connection of 50 8 as well as 500 S aerials. For most purposes, the 500 Q input is preferable, since it allows short aerials to be correctly ter- minated. Input transformer Tri is wound on a ferrite core Type FT37-75 from Micrometals. The total number of turns is 19, with a tap at 2Vz turns from the ground connection. The input should be provided with an overvoltage protection if the aerial is a large, outside mounted, array. B 9-94 elektor india September 1987 CENTRAL HEATING CONTROL 72 This circuit is used for optimum regulation of the flow of hot water in a central heating system. It measures the water temperature, and arranges for a particular valve or pump in the system to be switched on to achieve a user-defined tem- perature distribution in the home. Residual heat in the cen- tral heating system can thus be used to lower the cost of fuel. Fig. 1 shows that water in tem- perature range I can be used for the central heating and the storage vessel, while that in range II is also suitable for directing to the boiler. In most cases, it is not recommended to re-use water with a temperature below 30 °C. The circuit ar- ranges for an alarm to be activated when the water tem- perature falls below 5 °C, or exceeds 95 °C. The circuit diagram of the cen- tral heating control appears in Fig. 2. Relays Rei and Res are activated upon measuring the maximum and minimum per- missible temperature, respect- ively. The temperature sensor is a Type LM35, which has a scale factor of + 10 mV/°C. Its output voltage is amplified in Ai and fed to the non-inverting inputs of comparators Pa -At. The presets at the inverting input of each of these is used to set the toggle voltage. i.e., the tempera- ture at which the relevant relay 1 95 90 --- » Rei > ■■ • < ♦ TCC) is switched on or off. The relay drivers are open-collector power buffers with built-in freewheeling diodes to afford protection against inductive surges. The use of the Type ULN2003 makes it possible to use relays with a coil voltage of upto 50 V without the need for additional interfacing. Each temperature setting has a hysteresis of about 2 °C. Tran- sistors Ti -T 3 serve to disable the previously energized pump or valve upon detecting a water temperature that falls within another, predefined, range. In this manner, only one relay is activated at a time. It stand to reason that the tem- perature sensor, ICi, must be mounted such that it is in ther- mal contact with the water in the heating system. Make sure that the device is well-insulated, and that it does not cause leakage. The temperature range settings for the presets are shown op- posite. Relay Preset Temperature range 1 P. 93-103 °C (upper limit alarm) 2 P 2 77-93 °C 3 Pj 33-77 °C 4 P« 11-33 °C 5 P S 5-11 °C Gower limit alarm) (hysteresis on all toggle points: 2 °C). St ULN2003 I ± A A A x i ^ 1 r 87412-1 87412 2 elektor india September 1987 9 - 95 LOSS-FREE SUPPLY PROTECTOR by R Kambach the wrong polarity. The coil voltage of the relay may be lower than the input voltage, because Re is activated within a few milliseconds, and then receives the correct coil volt- age via Ti-Dt Since the hold voltage of a relay is generally lower than the actuation volt- age, D 2 can be dimensioned such that the relay operates reliably with a minimum of zener current taken from the supply. D Any diode-based circuit that protects against reversal of the supply polarity introduces a certain voltage drop. Also, when relatively high currents are involved, the choice of a suitable diode, and its dissi- pation, may become problem- atic. This circuit utilizes a relay con- tact to break the positive supply line when the input voltage has 1N40011 <7102 LOGIC FAMILIES The introduction of new, faster, CMOS techniques has given rise to a considerable increase in the number of available logic families. Understandably, this may cause confusion on part designers and users of logic circuits. Up until a few years, 3 families were commonly known: the CMOS 4xxx series; the TTL 74xx series; and the 74LSxx low-power Schottky series. TTL and LS chips are mutually interchangable, but TTL consumes considerably more current at the same switching speed. The 4xxx series is about 10 times slower than the TTL family, but is more economic as regards current consumption. In many cases, TTL chips are no longer con- sidered suitable for new design. The new HC and HCT CMOS families are just as fast as TTL and LSTTL, and have a greatly reduced current consumption. HCT chips can work in LS based circuits, provided they are not driven from TTL or LS. This is because of the differ- ently defined switching levels. It is, however, possible to use HCT for driving HC. With this in mind, it is possible to replace the LS family by the HC family. This is preferable since the HC family offers the highest noise immunity. Figure 1 shows the current con- sumption of a HCMOS gate as a function of the input voltage. The shaded area represents the (logic high) output voltage of an LS chip. From this, two con- clusions can be drawn. Firstly, the noise margin is very narrow: the HC gate sees 2.7 V as a logic "high level already. Secondly, the current consumption of the gate is a few mA higher than necessary. Although usable in practice, driving HC with LS is, therefore, not recommended. Another new logic family was recently introduced: FACT (Fairchild Advanced CMOS Technology), also referred to as ACL (Advanced CMOS Logic) by other chip manufacturers. There are 2 versions: AC and ACT. ACT, like HCT, is fully LS compatible, while AC gives the same drive problems as HC. Both series are typically 2 to 3 times as fast as LS or HC. Figure 2 shows the correlation between the propagation delay, tp. and the power consumption. P, of various logic families. It will be noted that the modern CMOS families are almost as fast as the ECL series, hitherto renowned for its unbeatable speed. It is expected, therefore, that a CMOS equivalent will soon be available for ECL, and that ECL will gradually become obsolete. Replacing bipolar chips in existing circuits with CMOS types is not very useful if rela- tively high frequencies are involved. Finally, a rule ot For further reading: thumb for working with chips of RCA CMOS Databook different families in a single cir- Fairchild FACT Logic Data Book cuit: HCT can replace LS, unless driven by LS. W 4000 9-96 elektor mdia September 1987 l TTL LS HC/ ALS ACT ' a F « T AS S 23 _| | ■ _ _ _ FRONT-END FOR FM RECEIVER Among the most important i Tv and again filtered. The technical characteristics of a overall gain between the aerial VHF preamplifier are the noise input and the mixer input is figure, and the large signal about 12 dB at 87 MH2, and handling capability. Although 17 dB at 108 MHz. The differ- these are in principle conflict- ence is caused by the adopted ing requirements, a compro- method of filter coupling. A mise can be found in the use of wideband Schottky DBM high-quality RF components, (double balanced mixer) is The receiver's ability to with- used for the mixer in this stand high input levels can be design. The Type SBL-1 (LO = enhanced by providing suf- + 7 dBm) is probably the best ficient selectivity ahead of available of the 3 DBMs stated, the active element(s). This is Tuneable local oscillator T2 pro- especially important for the duces very little phase noise, mixer, since it generates most and DG MOSFET T3 provides a intermodulation products. LO power of 50 to 100 mW at a In this FM tunerhead, the aerial drain current of about 25 mA. signal is first passed through a FET T« enables driving a slightly overcritically coupled prescaler or a synthesizer with band filter, amplified with the the LO signal. Series network j aid of low noise UHF transistor R9-C20 is fitted at the input of the < £ 100 mA) it ampuner Decause any pass- ive DBM should be correctly terminated on at least two of its ports. To compensate for the 6 dB conversion loss in the DBM, and to ensure some spare IF gain, medium power RF J- FET Ts is dimensioned to pro- vide a gain of about 12 dB at a drain current of 25 mA. The proposed front-end gives fairly good results: its third- order intercept point is better than 0 dB when a mixer is used with IP = +20 dBm, while the noise figure is about 4 dB. This sort of performance should enable the reception of quite weak transmissions even with a powerful transmitter within a tew mites trom tne receiver. Finally, due account should be taken of the fact that the IF out- put easily delivers 10 mW, which may well give problems if the IF amplifier is not prop- erly dimensioned. Inductor data for this project: L1...L5 incl. = ES26HNA10014 (Toko). L 6 = E526HNA10013 (Toko). L?. . .Ls;Lu= 6 turns 36SWG (G 0.2 mm) enamelled copper wire through a ferrite bead. L11 = 9 turns 24SWG (0 0.6 mm) enamelled copper wire on a T25-12 ferrite core; tap at 3 turns from C35-R15-R16. elektor India seplember 1987 9-97 This simple, one-chip, stereo preamplifier is ideal for build- ing into an existing AF power amplifier. It is based on a re- cently introduced integrated circuit, the Type TCASSOO or TCA5SS0 from Motorola. This double AF amplifier chip with inputs for balance, volume, and bass and treble controls forms a sound basis for a good quality preamplifier with a minimum of components. The onset points for the bass and treble controls are defined with C 3 and C 4 | respectively. All (mono) poten- i tiometers are best fitted direct | onto the circuit board to make for simple mounting into a cabinet, and also to prevent hum and noise being picked up in the wiring that would other- wise be required. [ The preamplifier has a current j consumption of 35 mA, of ! which 5 mA is drawn by voltage , regulator IC 2 . Zenerdiode Di ■ and power resistor Rs should be i added if the positive supply voltage available in the power I amplifier is more than about 30 V. Fe STEREO PREAMPLIFIER WITH TONE CONTROL Specifications of the preampli- fier: Distortion: Si 0.1% at nominal output level. Channel separation: 245 dB. Supply voltage: 8.8-18 V. Tone control range: 14 dB. Volume control range: 275 dB. Maximum input voltage: 100 mV. Amplification: 10. Low output impedance. Parts list Resistors t ±5% I: Ri..,R« incl. = 100K R5= see text Pi ... Pr incl. = 100K linear potentiometer Capacitors: Ci;C«;Cit = lOOn Cz;C« tOu: 63 V; radial C3;C4;C®;C7;Ci8 : 220n C5,Ci7 = lOOp: 40 V: radial Cio;Cis = 4p7; 63 V; radial Cn;Cn = 4p7; 40 V; radial Cu;Cn = 47n Semiconductors: Di= zenerdiode 27 V; 1 W (see text! ICi = TCA5500 or TCA5550 (Motorola) ICz = 7815 Miscellaneous: PCB Type 87405 (available through the Readers Services). TIMER FOR SOLDERING IRON ! By K Feigl when it is out of its holder for I activated. Pressing key Si Switch S 2 should be closed more than about 20 seconds. causes the circuit to become when the soldering iron is in its | It often happens that the The output frequency of clock operative again. Capacitor C 2 is holder or stand. This causes the soldering iron is left switched oscillator Ni is adjustable with not yet charged, so that ICi is counter to remain reset, and on, but unattended and out of Pi. Decimal counter ICi divides reset by N 2 . Gate N 3 is hence the relay to remain ener- its stand. Evidently, this is a the clock by 10. When both Qa therefore driven with two logic gized, until S 2 is opened. The waste of energy, and an un- and On are high, i.e.. when low levels, so that Ti energizes above timing sequence is then necessary fire hazard at the counter state 9 is reached, N 3 Rei. The mains voltage is now started, and can be interrupted same time. This circuit arranges turns off Ti. Hence both the applied to the soldering iron only by placing the iron in the for the soldering iron to be soldering iron and the timer are (= load Rl) and Tri via contact stand within 20 seconds. W switched off automatically switched off because Re 1 is de- re. q qq ' elektor india September 1987 ZJ-sjZj 78 SIMPLE SWEEP GENERATOR The sweep generator is an I mum duration is adjusted with indispensable piece of measur- | P 4 . The sawtooth voltage at pin ing equipment for testing the 6 of ICi has an amplitude of frequency response of AF 5 V PP , and can be used to drive amplifiers, filters, and loud- the horizontal deflection (X) speaker systems. At the heart of input of an oscilloscope via ter- this design is the well-known minal K. The amplitude of the Type XR2206 function gener- sawtooth voltage is determined ator chip from EXAR. It is seen by the zener voltage of Di and to the right on the circuit the base-emitter voltage of T 2 , diagram, in a standard appli- which is briefly turned off when cation with 3 capacitors and a the output of ICi exceeds 5 V. rotary switch for selecting the The collector of this transistor is frequency range, and a poten- then pulled to ground via R3. so tiometer, Ps, for adjusting the that Ti is switched into conduc- amplitude of the output signal, tion. The integrator is reset by The signal frequency is a func- making the — input of ICi posi- tion of the current drawn from tive with respect to the + input pin 7 on the XR2206: with the aid of Ts, Rs and Re. Capacitor Ci serves to lengthen fo= 3201/C [Hz] the on-time for Ti and Tj to ensure that the flyback of the where I is in milli-amperes, and sawtooth is finished. C is in micro-farads. It should Potentiometer Pi is a voltage be noted that pin 7 is internally divider to define the sawtooth kept at 3 V. which is available at amplitude, and hence the pin 10 also. sweep range, while Si makes it The left-hand part of the circuit possible to turn off the sweep I comprises the sawtooth gener- function (position F). ator, ICi, and a buffer, IC 2 . The Opamp IC 2 is configured as a former is set up as an integrator, buffer stage for inverting and i ! whose sweep period depends attenuating the sawtooth volt- ! on the voltage at terminal C. age, to which a direct voltage is j ; Potentiometer P 2 enables set- added also. The output of IC 2 ; ting the sweep period between carries a sawtooth voltage with 0.01 and 10 seconds; the maxi- an amplitude between 0 and | 2.8SV, or a direct voltage \ 3 frequency ranges. The fre- between the same limits when i quency scale can be calibrated Si is set to position F. Bearing in j with the aid of P3. mind that the reference voltage | of IC3 is 3 V, the current through R13, and hence the output fre- quency, can be varied by a fac- tor 20, which is the maximum attainable deviation factor in all 9-100 elektor india September 1987 Parts list Resistors { ± 5% ): Ri =22K R 2 ;R4;Ri7 = 10K R s - 4K7 Rs = 1K2 Re - 10R Rr= 1M0 Re = 68K R9 ,Rio-820K Rii;Ri2 = 470K R 13 = 2K2 Ru.Ris = 33K Rie-220R Pi - 50K linear potentiometer P 2 - 100K linear potentiometer P 3 - 100K preset P 4 = 100R preset Pe 1K0 logarithmic poten tiometer Capacitors: Ci = 3n3 C 2 = 12n C 3 = 68p C 4 - 1 M ; 16 V; radial Cs = 22n Ce = 220n Ct = 2m 2; 16 V; radial C«= 10 m; 16 V; radial C 9 = 2n2 Cio = 220m; 16 V; radial Cn;Ci 2 — lOOn Semiconductors: Di zenerdiode 5V6; 400 mW T i;T 2 = BC557 T3-BS250* ICt; IC 2 = CA3140 IC 3 - XR2206* + Miscellaneous: Si= miniature SPOT switch. Sz= miniature SPST switch. S 3 - 1 pole, 3-way rotary switch. PCB Type EPS87419 (available through the Readers Services). * Available from Universal Semiconductor Devices Limited, Cricklewood Elec tronics Limited, or Elec- troValue (28 St Judes Road, Englefield Green, Egham, Surrey TW20 OHB. Tele phone: (0784) 33603; telex 264475). • Available from Cricklewood Electronics Limited. 1( n tQ^SV o| f I - ® -© U 2 : U,: Us I *s I j I 1 Usync is applied to the Y input, and the vertical sensitivity is adjusted until the maximum excursion of Us reaches the top of the display. Set Si to position A (sweep 0.1 s), and adjust P 3 until the peak of the exponen- tial voltage is displayed in the top right-hand comer. This is repeated with Si set to position B (sweep 1 s), and the scope set to 100 mV/div. (adjust P 4 ). This completes the adjustment pro- cedure, and Us can be connec- ted to the VCO input. The current consumption of the cir- cuit is less than 25 mA or 15 mA with a 555 or a 7555 fitted, respectively. Th References: 1,1 Function generator. Elektor Electronics, December 1984. 121 Audio sweep generator. Elektor Electronics, November 1985. 9-106 elektor mdia September 1 987 POWER SWITCH FOR CARS Motorist are generally well aware that car fuses do not blow just like that. None the less, when something appears to be amiss in the electrical circuit, a new fuse is nearly always fitted prior to investigating the poss- ible cause for the malfunction, which then, of course, costs two fuses. The circuits shown here are short circuit proof power switches, or electronic fuses with switch control dimen- sioned for relatively heavy (lamp) loads in a car. Both cir- cuits are composed of a power switch, Ti, and a current limiter, T 2 . The circuit is fully short- circuit and overload resistant, provided Ti is adequately cooled, and the whole unit is constructed in a sturdy enclos- ure. The circuit in Fig. la has the lower voltage drop of the two, while that in Fig. lb is used when a TO-218 style Type MJE2955T or TIP29S5 is not obtainable. It is interesting to note that the plastic TO-218 package is mechanically inter- changeable with the well- known TO-3 outline, and en- ables ready mounting of the Resistors ( ± 5%) Ri;R 2 - see text Semiconductors: Ti = MJE2955T or TIP2955 (Fig. la) Ti - MJE3055T or TIP3055 (Fig. 1b) T 2 =BD136 or BD140 (Fig. la). T 2 - BD135 or BD139 (Fig. 1b). BD136 I BD140 1 BD135 BD139 Miscellaneous: Si - see text PCB Type 87467 (not available through the Readers Services). 87467- la 87467 ■ 1b transistor onto a flat surface using an insulating washer— see Fig. 2. The use of a die-cast enclosure and TO-3 style tran- sistors is illustrated in Fig. 3. This unit houses two power switches, one of which has its contacts at the rear side. Pay great attention to the correct electrical insulation between the transistors and the enclos- ure, and, if required, that between the enclosure and the car body. Switch Si is the exist- ing control for the relevant lamp in or on the vehicle Note the difference in respect of the connection of Si in Fig. la and lb. Table 1 shows how Ri and R 2 are dimension,, d in accordance with the current requirement of the load, and also gives a suggested area of the cooling surface. Finally, when the printed circuit board is used, Ti should be a TIP29SS or a MJE295ST, not a MJE295S, since this has its outer terminals (B-E) reversed. Table 1 Application Ri (W1 1 IAI R. IS] R 2 IQI Cooling T 1 -T 2 Dashboard lighting 1 0.08 5.6 3300 not required Courtesy light 2 0.17 2.7 1500 not required Rear light or parking light 5 0.42 1.2 680 25 cm 2 Brake light 18 1.5 0.33 (5 W) 180 (1 W) 225 cm 2 Fog light or trafficator 21 1.75 0.27 (5 W) 150 (1 Wl 225 cm 2 elefclor India September 1987 9-1 07 83 III 1 ' M9 mi # RAM EXTENSION FOR QUANTUM LEA The Sinclair Oantum Leap (QL) computer is eminently suitable for a low-cost introduction into working with Motorola’s 68000 true 16-bit microprocessor. Many computer enthusiasts did not fail to note the spectacular price cuts for the QL when its production was discontinued. An excellent support program, TOOLKIT II, became available and is still considered indis- pensable by many for getting to grips with the QL. The present 512 Kbyte RAM extension should be very welcome for running a RAM disk, and/or programs such as ICE and QIMP. The circuit is based around the Type THCT4502 RAM control- ler from Texas Instruments. This dedicated controller takes care of all the DRAM controlling, including the refresh timing, and the addressline multi- plexing. The address decoder is made with a single XOR gate, Nt. The DSMCL line is made high within 30 ns with the aid of three-state buffer Ns. Bistable FFi delays the ASL signal some- what, so that DTACL is only activated when the RDY output of ICt is stable. The databus is buffered by bidirectional octal transceiver IC23. The extension memory is div- ided in two banks of 256 Kbyte. Note that CAS, unlike RAS, is common to both banks. It is possible on the QL to omit the second bank without altering the address decoding. This is thanks to QDOS, which searches for correctly oper- ating, continuous, and unique, i.e., non-mirrored, memory. It is N1...N3 = 3/4IC2 = HCT 32 FFI = 1/2IC3 = HCT 74 N4...N6 = 3/4IC4 - HCT 125 N7...N8 = I/2IC5 = HCT 86 5 V © r-j 77 " las '/f, ACW 0 RFFRFO y/ O 1 ' s % MAI 1L ///„ £§ ^-O D3 ! s° m RA6 IC1 MA6 — RA7 THCT MA7 21 . CAO 4502 J^ 8 — CA1 ’ CAS* 3^-1 \ CLK « K J, | _141 42| 30 1 I2|37 ■ 3 — r/i^/csT (?) o”=, a; 1 I t Qi) ^ w ^s) C 2 I c 2ol , C 2 IC3 IC4 IC5 IC23 18. | 16v KK>< 7 ) ( 7 ) ( 7 ) ( 7 ) ( 10 ) 9-108 elektor mdia September 1987 interesting to note that machine code in the extension memory runs at almost double the nor- mal speed. The RAM chips used should have an access time of 150 ns or less. Current consumption of the extension is low at 50 mA or 150 mA in the non-active and active mode respectively. Non- used inputs on gates should be tied to ground. Finally, note that the Type THCT4502 controller may not be available everywhere yet. W Distributor for TI Semiconduc- tors in the UK is DC Distribution • Freepost • Hitchin Road • Arlesiy • Bedfordshire SG15 6BR. Telephone: (0462) 834444 or (0454) 273333. SYNCHRONIZED SLIDE CHANGER Sound and vision can be syn- chronized fairly easily for mak- ing slide presentations: the left and right channel of a cassette or tape recorder are used for separate recording and piaying back of the music and accom- panying speech, and the slide change control signal. The cir- cuit around Ai is a square wave generator that produces the 1 kHz slide change signal for recording on one track of the tape. Si is pressed when the next slide is to appear. Voltage divider Rs-Rs ensures that the recording amplifier is not over- driven. If necessary, Re can be replaced by a 8K2 resistor and a 2K5 potentiometer as shown to enable adjusting the output amplitude. The circuit around A 2 is the playback amplifier for driving the slide change relay. The quiescent current of the slide changer is about 15 mA. Some projectors have a slide feeder that can be moved in reverse with the aid of two short control pulses, or one long pulse. This is of course also possible with the present cir- cuit. R elektor India September 198? 9-1 09 FRUIT MACHINE cally. When a counter is dis- abled by the high level at its CE input, one of the LEDs in the 3 groups remains illuminated. The output state of the counters is not predictable because of the inconstant delay between the disable instants. NAND gates N13-N15 detect the win- ning combinations, i.e., LED D2 is illuminated, and Bzi is sounded, when 3 identical counter outputs are activated. Note that diodes D3-D5 form a 3 -input OR gate, and that the buzzer also produces sound when the LEDs are flashing, since the pulses at output O2 of IC3 enable the oscillator inter- mittently. The play is ended when the voltage across C3 is high enough for gate N? to change By F Pipitone This is one of the very few "one- armed bandits” to which the maxim the sole way to win is not to gamble is not applicable. In other words, this circuit does not have a slot for inserting coins: every play is free. Actuation and release of the "PLAY" button, Si, causes the circuit to become operative. Series regulator Ti is driven into saturation by T2, which is con- trolled by N2-N7. The outputs of N2, N11, and N10 go high in suc- cession, and disable counters ICs, IC4 and IC3, which are all clocked by oscillator N12-N9, and reset by the pulse at their Q3 output. The 3 LEDs driven by each of the counters are, therefore, illuminated cycli- the fruit machine, thanks to its very low current consumption in the de-activated state. R state. T2 is turned off, and Ti no longer powers the circuit. An on/off switch is not required for 9-110 elektor india September 1 987 Parts list Resistors (±5%): Ri;R3;R4;Rs;R8;Rio= 10K R 2 = 4K7 Rs = 220K R7 = 1M5 Rs = 470K Rn =3M3 Rl2;Ri4-33K Ri3=100K Ris = 1K0 Ri6=330R Capacitors: Ci;C 3 = 4p7; 16 V; axial C2;Cs;C6 = lOn C 4 = Ip; 16 V; axial C?;C8;C9= lOOn Semiconductors: Di;D3;D4;Ds=1N4148 D 2 ;D 6 ;D 7 ;D 8 = LED (red) D 9 ;Dto;Dn = LED (yellow) Di 2 ;Di 3 ;Di 4 = LED (green) IC1.IC2 = 40106 IC3;IC4,IC5 = 4017 IC 6 = 4073 Ti = BC557 T2 = BC547 Miscellaneous: Si = momentary action push button. 82 = PB2720 buzzer (Cirkit stock no. 43 27201). PCB Type 87476 (available through the Readers services). I WEATHER SATELLITE INTERFACE An increasing number of elec- tronics enthusiasts is becoming interested in weather satellite reception. Most non-geostation- ary weather satellites, like those in the NOAA series, operate in the 138 MHz carrier. For opti- mum reception, the detector should feature a relatively high carrier suppression. It is assumed here that a picture signal is available on a cassette tape. Opamp Ai has an amplifi- cation of 48, while Az-Ai form a precision two-phase rectifier. The 2,400 Hz ripple arising from the slightly different specifi- cation of the opamps amounts to no more than 0.2%. For com- monly used A-D converters, this corresponds to an error smaller than >/ 2 (LSB). The main ripple signal is 4,800 Hz. This is readily removed by a a double n filter set up around Li and L 2 . At 2500 Hz. the attenuation is about 3 dB. at 4,500 Hz about 45 dB. The parallel R-C and L-C networks at the + input of As compensate for the ohmic resistance of the inductors in the n filter. Li, L 2 and L 3 are preferably ferrite- encapsulated chokes from the Toko 10RB senes, available from Cirkit PLC (Li & Lz: 181LY-473. Li: 181LY-104). The interface is suitable for processing carrier frequencies up to 4,800 Hz, so that it is possible to play the tape at double speed for read- ing into the computer (pro- vided, of course, the program can handle this). Components R 11 , D< and Ds protect the A-D converter against voltages higher than 5 and lower than 0 volt. The use of the Type CA3130 BiMos opamp ensures an output voltage swing of 5 V when a +6 V supply is used. The maximum supply level and current consumption are +9 V and 15 mA respectively. The input signal amplitude should be greater than 68 mV rms for a 5 Vpp output. B elektor india September 1987 9*1 1 1 I £ 1 5mA 6V I INSTRUMENTATION AMPLIFIER This instrumentation amplifier was originally designed for the senal digitizer described in " , but should be suitable for many other applications also. The amplifier makes it possible to use a relatively long, inter- ference-free, connection be- tween the transducer or sensor and the digitizer input. The theoretical basis for the cir- cuit is summarized in the accompanying Table. It is seen that the common mode rejec- tion of the amplifier serves to suppress interference. In prac- tice, however, the low drive margins of the inputs and out- puts of the opamps impose some limitations. Both suggested types have PNP input transistors capable of handling input voltages be- tween 0 and Ub-1.5 V. The out- put of the OP-220 can deliver voltages between 0 and Ub-1 V, that of the LM358 between 0 and Ub-1.5 V. The current consumption of the opamps is low at about 150 /iA for the OP-220, and 1 mA for the LM358, while the slew rate is about 0.04 V/j£ and 0.4 V/|£ respectively. For optimum accuracy it is recommended to use high stability (1%) resistors in positions Ri-Rs inclusive. Sv Reference " Universal peripheral equip- ment (2): Serial Digitizer Elektor Electronics. September 1986 p. 23 ff. Micropower instrumentation amplifier Consider an input UCM iUd at the input of the circuit, and UCM * V>Ud at the r input. This corresponds to a common mode input UCM. and a differential input Ud The currents at the inverting input of each opamp can be summed to form two equations (Ub-UCM-f ViUd) (1 R.) + (Ud/Ro) HU. ♦ ViUd) 11 R3) (UCM- !6Ud) Cl R2> (U* UCM Vi Ud I (1 R4|tUoViUdJ (1 R5> Ud Ro ( 2 ) When Ri R? 2Ra 2R« 2R* 2Rx, (1) and 12) can be combined to Uo 2(1 *R*/Ro)Ud + ViUb which shows that the common mode input (UCM) has been rejected. The dif- ferential gain. A*, of the circuit is therefore A* 2 t <2Rx Ro) and is adjustable between 0 and 1,000 by varying Ro R3= R4 = R5 = R x PMI Application 87464 9-112 elektor india September 1987 By R Baltissen Many of today’s cars and motor cycles are equipped with a meter for monitoring the bat- tery voltage. However, this meter does not provide infor- mation on the battery condition, or whether it is being charged at all. When the voltmeter reading is too low, the battery is generally in such a poor state as to necessitate switching off heavy loads to save power for use of the starter engine later. Especially on motorcycles, the battery capacity is relatively low, which justifies the need for a reliable monitoring system. A standard 30 A ammeter offers too low resolution, and is rather awkward to fit permanently. In this charge/discharge indi- cator, the measured current is converted into a potential differ- ence by R*. which is either two 1R0 5 W resistors, a fuse, or a few turns of copper wire. The direction of the current through R* is detected by comparator ICi, which then indicates whether the battery is being charged or discharged by light- ing the relevant LED. The 100R preset enables shifting the indi- cation threshold somewhat. Input terminal + on the indi- cator unit is best connected to a point behind (that is, electri- cally behind) the contact switch, although it is also poss- ible to fit the circuit with a separate on/off switch. Finally, the circuit is only suitable for use in or on vehicles having a 12 V battery. R * see test B = remaining loads C = starter motor 87474 01 = cnarge 02 = discharge 59 AUDIO LINE C >RIVER Integrated operational ampli- fiers are not always suitable for applications where a high signal level (Uo<10 Vims) is required for driving a relatively low impedance (Z = 50— 600 Q). The amplifier described here is eminently suitable as a high dynamic range line driver or power buffer in public address systems and AF distribution amplifiers. The input amplifier of the line driver is formed by a low noise opamp Type OP-37 from PMI. This ensures the following technical specification of the line driver: Uo=70V PP max.; Io = 400 mA PP max.; Dtot=0.01% at Uo=10 Vrms, Zl=50 Q and S/N > 90 dB. Regulators T 1 -T 2 bring the supply voltage for the OP-37 down to ± 15 V. The comp- lementary power output stage is formed by T 3 -T 4 . The ampli- fier has a standard nega- tive feedback circuit R 1 -R 2 , which results in a voltage gain Av=— (R 2 /R 1 ). A local feedback R 3 -R 4 has been included to keep the output voltage of the opamp within safe limits, while capacitors C 1 -C 2 serve to improve the stability. It should be noted that the value of Ci and C 2 depends on the con- struction of the line driver: typical values are 680 pF for Ci and 22 pF for C 2 . In a prototype of the circuit, neither capacitor was required for the frequency response to remain flat (±1 dB) up to 100 kHz. Resistors Rb should drop just enough voltage for Ts and T< to start conducting (class A-B operation). The quiescent cur- rent of ICi is about 3 mA, so that 150 Q can be taken as a suitable starting value for Rb. The quiescent current in the power output stage should be be- tween 20 and 50 mA. Higher values of Rb cause the quiesc- ent current, and hence the power dissipation, to increase, resulting in less distortion. The power output stage is not pro- tected against thermal over- loading, so that due dare should be taken in adjusting the quiescent current. Sv 36V elektor India September 1987 9-1 1 3 jr TWO-TONE RF TEST OSCILLATOR This test oscillator is useful to ensure optimum operation of RF amplifier stages designed to work on the short-wave bands. Based on two crystal oscilators, it provides considerable output power (10 to 100 mW) to enable measuring intermodulation characteristics of high level and RF power stages. The quartz crystals used here not only serve as the frequency deter- mining elements (2...20 MHz), but also as output filters to pre- vent one generated signal being lost in the other oscil- lator. With this in mind, tapped inductors Li and Ii 2 ensure freedom of mutual interference when the oscillator is used for frequencies higher than 10 MHz. Both inductors are wound as 12 turns of enamelled copper wire with a centre tap, on either a small balun or a suitably rated core with an air gap. Outputs of equal ampli- tude can be obtained by ad- justing Pi. The test oscillator consumes about 250 mA from a 60 V supply. This means that both transistors should be fitted with a heat-sink, and that chokes L 3 and 1,4 should be capable of carrying about 150 mA. B 10... 2048m 50 n * see text 87478- A TELEPHONE LIGHT In this circuit, a relay is ener- gized when a call signal is received, or when the receiver is lifted. The relay remains ener- gized for a short period after ringing off and after the call signal has ended. An opto- coupler is used here in view of the direct connection to the telephone line, and the rela- tively high bell voltage on it. Current is passed through the I LED in IC, via rectifier D,-D<, so j that the phototransistor con- ducts. C, is discharged via Ds, N 3 changes state, and Ti ener- gizes the relay. Also, C 2 is charged via De, and so ensures that the lamp remains illumi- nated for a few seconds after the receiver has been put on the rest, or after the bell signal has ended. The current con- sumption of the circuit in the stand-by mode is less than 10 mA. The coil voltage of the relay should be equal to the supply voltage used. St Note: connection of this circuit to the British Telecom tele- phone network is not permit- ted. € 10 mA 9V 9-114 elektor india September 1987 NON-INTERLACED PICTURE FOR ELECTRON Owners of the Acorn Electron home computer may well object to its interlaced, and therefore slightly instable, pic- ture. There is a trace of display flicker in non-moving areas on the screen, and this is mainly due to the internal video pro- cesing circuitry operating on the basis of interlacing, a tech- nique used in conventional TV transmission for smoothing the appearance of moving picture areas. Arguably, interlacing is not very useful in computers, since these work with text in most applications. Special dis- plays with a relatively long afterglow time are no remedy for this awkward problem, and that is why the present circuit was designed. It effectively switches off the interlace func- tion, and so ensures a restful display, albeit that the indi- vidual lines that make up the characters become slightly more prominent. Figure 1 shows that a TV pic- ture is composed of 625 lines divided between 2 rasters of 312.5 lines each. In an inter- laced picture, these rasters are vertically shifted by one line. This is done by starting the sec- ond raster x and a half time later than the first raster. Interlacing can thus be rendered ineffec- tive by starting the second raster half a line period earlier (ie., after 312 lines rather than 312.5). To retain the normal number of lines (625), the sec- ond raster is arranged to com- prise 313 lines. The ULA chip (Uncommitted Logic Array) in the Electron computer provides a horizontal and a composite synchroniz- ation signal, which are shown in Figs. 2a (HS) and 2b (CSYNC) respectively. With reference to Fig. 2c, and the circuit diagram in Fig. 3, MMVi forms a new vertical synchronization pulse, VS, with the aid of the CSYNC signal. The period of pulse VS is different for the first and second raster, so that MMV 2 is needed to make VSYNC equally long in both. MMV 2 is triggered on the first line pulse (HS) that occurs when VS is active, and is retriggered when VS goes low— see Fig. 2d. The length of the VSYNC pulse so made is about 160 \i s, or about 2.5 times the line time (64 pis). The HS and the new VS signal are combined in XOR gate N 2 for driving the video modulator. Gate Ni serves to buffer the HS output of the ULA. The final results obtained with the circuit depend mainly on the type of TV set or display used, and may not be optimum when the TV is driven via its RF input. On an older type mono- chrome set, the central area of — -- first raster - - - flyback “ (blanked) second raster ir~ u u u; u u it 11 = period of MMV 1 12 = period of MMV 2 87485-3 elektor india sentember 1987 9"1 1 CSYNC N2/6 pcb Circut 87485-4 the picture was stable, but the upper and lower areas gave a less favourable look. Good results were obtained, however, from the use of Type TX chassis, which are currently the basis of TV sets sold under many different names and licenses. Even better perform- ance can be expected from a video monitor, whose (TTL compatible) H and V synchron- ization inputs can be driven by N 4 and N 3 respectively. The polarity of the sync signals can be selected with the aid of wire jumpers. Connections c and c’ result in VSYNC and HSYNC. The choice between jumper a or b depends on the type of dis- play used. Preset Pi is adjusted until the picture appears ver- tically synchronized: the adjust- ment is fairly critical when jumper a is used. The final results obtained with the circuit can be judged from looking at a few characters in the upper and lower area of the screen. The modest current consumption of the circuit, 10 mA, makes it possible to power it direct from the Electron computer. TW Extremely short, unwanted, pulses with a period in the nanosecond range are often referred to as glitches, and occur in most, if not all, digital circuits. Whilst the circuit in question can be designed and built with due attention paid to effective suppression of glitches, it is not always poss- ible to foresee the effects of external noise on, for instance, a clock signal. The filter pre- sented here effectively rules out the presence of glitches in a serial data link. Assuming that counter ICi is at state nought, and that the data input is logic high, IC 2 is con- figured as an AND gate. Output Q 4 of ICi, and hence the output of the deglitcher, goes high after 8 clock pulses. A short negative pulse at the data input merely results in a few more clock pulses being required before Q4 is activated. After another 8 clock pulses, the 87497 counter state is IS. This causes the Cl (CARRY IN) input of ICi to be driven high, so that the clock signal remains blocked as long as the data input is logic high. When it goes low, IC 2 is configured as a NOR gate, enabling the clock transitions to be counted down in ICi. Out- put Q 4 goes low again after 8 clock pulses, and the counter is blocked after another 8 pulses. Therefore, the filtered output data is delayed by 8 clock periods, but this is insignificant in the proposed application. The data frequency, foi, de- pends on the clock frequency, ftcLi: f[D] = f[CL]/16 The maximum usable clock fre- quency is about 8 MHz. The current consumption of this cir- cuit is less than 1 mA B 9-116 elektor india September 1987 NOISE BLANKER Kbl A noise blanker is indispens- able for improving the recep- tion of very weak signals on the SW bands. In most communi- cation receivers, the selectivity of intermediate frequency (IF) filters causes interfering pulses to be widened, blotting out the wanted signal. It is useful, therefore, to suppress inter- ference before this can wreak j havoc in the IF sections of the receiver. The 455 kHz IF signal is first buf- 1 fered in T 2 , and then processed separately in two circuits. The lower section of the circuit is a TCA440 based receiver for the interfering pulses. The TCA440 is in itself a virtually complete receiver, since it comprises an RF amplifier, a mixer, and an IF amplifier. All stages in the latter are used since pin 4 is grounded here. The pulse receiver has its own AGC (automatic gain control) to ensure effective suppression of relatively weak interference also. Preset Pi and poten- tiometer P 2 enable precise adjustment of the noise blanker for various levels of inter- ference. The circuit can be con- trolled digitally via R23; a logic high level renders the noise blanker ineffective. The inter- fering pulses are made logic compatible with the aid of opamp IC 2 . LED D 3 lights when noise is detected. In the upper section of the cir- cuit, the IF signal is first delayed in FLi to compensate for the processing time in the pulse receiver. ESi is opened when a sufficiently strong interfering pulse is recognized, so that the IF signal is no longer applied to output buffer T 2 . Also, the gate of this FET is then grounded for RF signals via ES3-C4, while ES2 is closed to maintain correct ter- mination of FLi. Properly constructed, this cir- cuit achieves noise suppression of the order of 85 dB. Alterations to suit operation at an IF other than 455 kHz merely involve Li and FLi, although due account should be taken of the parasitic capacitance of the electronic switches at relatively high fre- quencies. B elektor India September 1987 9-1 1 7 VIDEO DISTRIBUTION AMPLIFIER The Type TEA5114 from Thom- son-CSF comprises three elec- tronic switches followed by a buffer/amplifier. Normally the voltage amplification is 2 (6 dB). When the input voltage ex- ceeds 1.2 Vpp, or when the out- put voltage exceeds 1.5 V PP , an internal selector reduces the amplification to unity (0 dB). The threshold of 1.2 V PP is created with the aid of voltage divider R<-Rs, which also forms the input termination of 75 3. Series resistors Ri-Rj ensure 75 3 output impedance for driv- ing video equipment via stan- dard coax cable. The TEA5114 can be used as a video source selector also, provided each input has its own 75 Q termina- tion network. The non-connec- ted inputs should then be fitted with a coupling capacitor. Channel selection is effected by controlling the logic level at pins 10, 12 and 15. Note that the logic 1 (high) level corresponds to + 2.5 V here. D Resistors ( ±5%l: Ri Rs incl. = 15R R< -47R Rs = 27R 750 video outputs Capacitors: Ci . C« incl. = lOOn Semiconductor: ICi = TEA51 14* Miscellaneous: PCB Type 87466 (not available through the Readers Services). Thomson Components Limited • Ringway House Bell Road • Danneshill • Basingstoke • Hants RG24 OQG. Telephone: 10256) 29155. For distributors see Infocard 502 [EE February 1987). • OOOOl ©V 8 8 8 o rv i 6 ! 6 1 6 ] 9 9 9 o 9 1 18 elektor mdta September 1987 This two-key wiper delay cir- cuit is remarkable for its simplicity and ease of use. The wipe is started by pressing the set switch, which also serves to adjust the length of the wipe interval. The circuit is turned off by pressing the reset button. The wiper delay shown in Fig. 1 consists of three opamps and a monostable multivibrator (MMV). Opamp Ai is set up as a triangular wave generator, con- trolled by the output of the MMV. When this is low, a slowly rising sawtooth voltage appears at the output of Ai. The rise time of the sawtooth depends on R2-C3. Opamp A 3 compares the voltage across O, to the instantaneous sawtooth ampli- tude. The output of A 3 drops from 8 V to 0 V when the sawtooth voltage exceeds U(C4]. This change in the output volt- age of A 3 is delayed by R 6 -Ce and passed to A2, so that the MMV is triggered somewhat later. The wipers are switched on via Ti and Re when pin 3 on the S55 goes high. Also, C 3 is rapidly discharged via Di and Ri, while D 2 prevents the volt- age across C3 becoming positive. When the MMV output goes low, Ai generates a new sawtooth period. When the circuit is first switched on, C< is discharged, and the output of Ai is slightly higher than 0 V due to Vicei of the internal output transistor. This causes the outputs of A 3 and A2 to remain low, so that the wiper relay remains ener- gized initially. When reset is pressed, C4 is charged via Rs, causing the the output of A3 to go high, and the MMV to be stopped. The delay circuit around A2 is necessary to pre- vent C4 being discharged com- pletely after pressing the set button. The relay contacts should be wired such that the dashboard switch is by-passed when the relay is energized, and that the hold switch, H, for the wiper motor is opened— see Fig. 2. Due attention should be given to the correct connection of the hold switch on penalty of short- circuiting the car battery. R eleluor India seplembei 1987 9-1 1 9 DRIVER FOR BIPOLAR STEPPER MOTORS For some applications, the Universal control for stepper motors (see 1 1 f ) may be con- sidered too extensive a circuit. Many small motors with limited speed range can be equally well controlled by a relatively simple circuit, based on, for instance, the Type SAA1027 or TEA1012 |S1 . Most commercially available controllers are, how- ever, intended for driving uni- polar stepper motors, which are now gradually superseded by bipolar types of similar size. In different type of controller. The recently introduced Type MC3479P from Motorola re- quires a minimum of external components for controlling a bipolar stepper motor. The maximum quiescent stator cur- rent, L, depends on the value of resistor R between pin 6 and ground: above relation between Is and R is valid as long as the output transistors are not operated in the saturated area. The satu- ration point is reached sooner at low levels of the supply voltage, or when the ohmic resistance of the stator winding is fairly high. The manufac- turers state a maximum current of 350 mA per stator. The supply voltage for the motor (pin 16) depends on the ohmic resistance of the stator windings, and is allowed to vary between 7.2 and 16.5 V. When a high supply voltage is used, it must be remembered that the output transistors will not operate in the saturated area to prevent exceeding the set stator current, Is. The current control used here allows a fairly high step rate at the cost of an increase in the dissipation of the driver IC, particularly when the motor is held stationary. If necessary, the MC3479P can be cooled by connecting the 4 central ground terminals to a relatively large copper surface on the PCB. The integrated controller has 4 TTL and CMOS compatible inputs (see Fig. 1): supply at both ends (1). The lat- ter option improves the damp- ing of the motor in the half step mode, and will prove useful at relatively low step rates. Pin 11 of the driver IC is an open-collector output with a current capacity of 8 mA, activated during period A in Fig. 3. A LED connected to this output will flash rhytmically when the motor is running. Transistor Ti was added to obtain a reset function. No stator current flows, and the logic circuitry in the driver is reset, when the stand by input is driven low. When a logic 1 is applied, the motor is energized starting from state A. The ad- dition of R 2 makes it possible to switch the driver to the power- down state, rather than the reset state. The stator current is reduced to the value set with R 2 , as shown in the above formula. The motor driver is probably best controlled by a computer CLK (pin 7): every rising edge of the clock signal causes the motor to revolve one full or one half step, depending on the level at pin 9. The maximum step rate and the minimum pulse width are 50 kHz and 10 ys respectively. CW/CCW (pin 10): the logic level applied here determines the motor's direction of travel. F/H step (pin 9): this input allows selection between full (0) or half step (1) operation— see Fig. 3. OI (pin 8): this output impedance selection input is only effective in the half step mode. It determines whether the sta'or winding is effectively disconnected from the driver (0), or connected to the positive Is=(Ub-0.7)/0.86R [mA] where R is given in kQ. The B5 7V 2—16V2 k'onh-ffl CW/CCW CW/CCW CW/CCW Driver Driver output impedance BC547B 87504 87504 • 2 9-1 20 elektor mdia September 1987 output port. The circuit in Fig. 2 is intended for stand-alone applications. It is composed of a supply, R5-D3, an oscillator, N1-C3-R9-P2, and a re-trig- gerable monostable multivi- brator, N2-C2-R10-D2. When Si is opened, the oscillator is en- abled, and the motor will start running. The clock frequency, i.e.. the step rate, is adjustable with P2. The monostable will remain set via D2, and Ti will conduct, as long as clock pulses are applied to the motor driver. The amount of ever reversing stator current is limited by the stator inductance, but can be still be increased with the aid of Pi. When the motor stops, Ti is turned off, and the stationary stator current is reduced to the value set with R2. The above arangement keeps the dissipation of the motor and the driver within reasonable limits. The current consumption of the complete circuit is practically that of the motor alone (700 mA max.). The motor driver IC con- sumes about 70 mA. TW OUTPUT SEQUENCE Phase A B Phase A Output 1 High Impedance Logic 0" Don't Care (a) Full Step Mode '////// - High Impedance CW CCW = Logic "O' F HS = Logic 1 OIC = Logic 0 (b) Half Step Mode References: Phase A Output CW/CCW Logic 0 F/HS - Logic 1 OIC = Logic ’T [l > Universal control for stepper motors. Elektor Eelectronics, January 1987. 121 Stepper motor control. Elektor Electronics, July/August 1986. 87504-3 LOW CURRENT AMMETER adjusted for full scale deflec- tion of Mi at an input current of 1 «A. When it is intended to make a printed circuit board for the pico ammeter, it should be borne in mind that two 2.5 cm long, parallel running, copper tracks spaced 1.25 mm and etched on a high quality epoxy/ glass carrier represent a leak- age resistance of about 100 GS. This corresponds to a leakage current of 150 pA at a voltage difference of 15 V. Evidently, the PCB for the present ammeter should be thoroughly cleaned to rule out leakage cur- rent through residual moisture or resin. Also note that the insulation of standard test leads is likely to make reliable measuring of currents smaller than 1 pA impossible. The only way to overcome this diffi- culty is to use dry air or PTFE (Teflon). Sv This 7-range ammeter measures currents between a few pA to 100 pA without using precision resistors with very high values. The circuit is set up around a current mirror Tia-Tib. The input current is mirrorred in this transistor pair, and "the current through Tib is greater than the input current by a factor set with Si. Meter Mi is a 100 ixA fsd type for displaying the measured value. The effective series voltage drop at the input terminals of the instrument is only 500 because the voltage across the inputs of A i is forced to virtually nought. The accuracy of the ammeter depends mainly on the compo- nents used. Depending on the required precision, certain components may be replaced by types with a better specifi- cation. The Type LF411 opamp used in the Ai position, for example, can be replaced with the Type OP-41 to achieve a ten- fold reduction in the input bias current, and hence an improve- ment in the final accuracy of the instrument. Transistor pair Tu- Tib may be replaced by a Type MAT-02, and the voltage refer- ence set up with T 3 -T 4 by a Type LM313. These high-quality parts should ensure an accu- racy of 1% over most of the range. The meter is calibrated in the 1 >iA range. Preset Pi is Source: PMI Linear and Conver- sion Applications Handbook. elektor mdia September 1987 9-1 21 Cl II C 2 H ir If 2 p 2 u fe| - 1V2-.1V4 i* r R4 PI 1 Ml 1 2k2 I r~y T1...T4 = IC1 = CA3046 A 1 = IC2 = LF4 1 1 , 0P4 1 R5...R10 = 5110 , 1% D1...D2 = 1N4148 cal SI 11 a 100 pA b 1 nA c 10 nA d 100 nA e 1 pA 1 10 pA 9 100 pA 99 From an idea by M Schultz. A wiper delay is essentially a bistable multivibrator whose off-time is adjustable with a potentiometer. Many wiper delay circuits are based on the Type 55S timer in its standard application circuit, which has the disadvantage of introducing a delay of about 1.6 times the set interval before the first wiper action takes place. This is especially annoying when an interval of, say, ten or more seconds has been set. This cir- cuit is also 555 based, but is unique in that it arranges for the wipers to be activated immedi- ately at power-on. The circuit diagram of Fig. 1 shows the internal organization of the 555 timer to aid in clarify- ing the operation of the present circuit. When SI is closed, pin 6 is immediately pulled to + 12 V because Ci is discharged as yet (see also Fig. 2b). The bistable in the 555 is reset, the output goes low, and Rei is energized. This forms the basic difference with the standard application of the 555, where Ct, connected as shown in Fig. 2a, delays the relay action until charged to Vs of the supply voltage. Returning to Fig. 1, Ci is charged via R 2 and the 555's internal transistor when the output is activated. The bistable is reset when the voltage at pin 2 drops below ViVcc, causing the relay to be de-energized, and Ci to be discharged via R 1 -P 1 . The dis- FAST STARTING WIPER DELAY «3>-* ! charge time, and hence the wipe interval, is defined by the setting of Pi . When this is set to the shortest delay, the wiper motor is constantly powered via Rei, since C< is not charged via P 1 -R 2 only, but effectively via voltage divider P 1 -R 1 -R 2 also. The wiper delay is fed from the 12 V car battery, and its current consumption is practically that of the relay alone. Note that the coil current may not exceed 200 mA. TW 9-122 elektor mdia September 1 987 100 FISHING AID From an idea by C Trimbach This circuit provides audible and visible warning when a fish is nibbling the bait. Although this event is fairly easy to signal with electronic means, the cir- cuit is relatively extensive to ensure that it can be powered from a 9 V battery. The circuit is based on a slotted opto-coupler Type CNY37, and a home made notched wheel. Unfortunately, the current am- plification of slotted opto- couplers is very low (0.02 min.), requiring considerable current to be fed through the LED before a usable collector cur- rent flows in the phototransistor. To avoid rapidly exhausting the battery, MMVi pulses the LED at about 250 Hz and a duty factor of 0.05. MMV 2 detects the presence of these pulses. When a fish pulls at the bait, the notched wheel revolves in the slot, and intermittent pulse bursts are received at the trig- ger input of MMV 2 . Green LED Di lights, buzzer Bz beeps, and bistable N3-N4 is set, so that red LED D? flashes at a 1.5 Hz rate. Di and the buzzer are turned off when the fish gets off after nib- bling the bait, but D 2 continues to flash. The circuit around Ni, T 2 and C 3 then serves to keep the current consumption as low as possible. The circuit can be reset by pressing Si. Preset P\ enables adjusting the frequency of the buzzer oscil- lator between 600 and 2500 Hz. When several fishing-rods are being used, each can be as- signed a particular signal tone. The buzzer can be switched off by means of S 2 . A suggested construction of the light barrier and the notched wheel is shown in Fig. 2. A small shaft is used in combi- nation with a reel around which the fishing line revolves. The slots cut into the detection wheel should not be too wide: 1 mm is a good starting value. The detection sensitivity is determined by the number of slots in combination with the reel diameter. The light barrier should be screened from day- light. In the stand-by condition, the circuit consumes no more than 4 mA, which goes mainly on account of the LED in the opto- coupler. In the actuated state, the current consumption rises to about 12 mA. TW elektor inclia seplember 1987 9-1 23 BIDIRECTIONAL PARAL- LEL INTERFACE FOR C64 The so-called User Port on the Commodore C64 home micro is intended for connecting per- ipherals such as a modems, RS232 interfaces, and control circuits. In some applications, it is also used for ommunication with other C64s_ This circuit makes it possible to use port lines PB0-PB7 as inputs and out- puts. Software enables the com- puter to select between input and output by means of the PA2 line (terminal M). Examples: Data input: 10 POKE 56579,0 :REM user port is input, 20 POKE 56576,255 :REM interface is input. 30 A = PEEK(56577) :REM read variable A. Data output: 10 POKE 56579,255 :REM user port is output. 20 POKE 56576,251 :REM interface is output. 30 INPUT B :REM read dataword. 40 POKE 56577, B :REM and send to interface. green The circuit is essentially com- posed of 2 three-state octal bus drivers Type 8212. Via the logic level on PA2, each driver can be enabled individually so as to select between the input or out- put function of the interface, whose current state is indicated by a pair of LEDs. Switch Si selects between pull-up (a) or pull-down (b) termination of the input lines. Finally, an example for interac- tive data processing: o z o C - " - > > • 'CUMUi/>a.444U rj 2345 6769^1 B C O E f H J K ll on z o * — n>m«u»<0^r«Q .. asC6a)CB5Bocoa<* 10 POKE 56S67.2S5 110 IF A = 1 THEN B = 64 :REM interface is input. 111 IF A = 2 THEN B = 128 20 POKE 56579,0 112 IF A=4 THEN B = 192 :REM user port is input. 113 IF A-60- 1 THEN B = 32 30 A = 255-PEEK(56577) :REM read variable A. 100 :REM example of logic control: 300 POKE 56S77.B :REM load data register 310 POKE 56579,255 :REM user port is output 320 POKE 56576,251 :REM interface is output 330 GOTO 10 9-124 elektor india September 1987 Closed Circuit Open Circuit LD1 = least significant digit 1C 1. IC2 = 74(LS/HCT) 168 =» Dl 74 (LS/HCT) 169 ^ HI IC3 , IC4 = 9368 N1 , N2 = 1/3 IC5 = 74 (LS/HCT) 14 ERGONOMIC THUMBWHEEL Open Circuit Closed Circuit Channel B Industrial engineers and system operators need not be told of the often awkward problems that arise from having to change a thumbwheel setting. It is not surprising, therefore, that alternatives for the good old thumbwheel switch are cur- rently finding their way in new designs. The present circuit was designed to work with the Type ECW1J-B24-AC0024 digital contact encoder from Bourns. This device looks very much like an ordinary potentiometer, but essentially contains two switches. These are normally opened, and successively closed when the spindle is rotated one step. The order of closing is determined by the spindle’s direction of travel- see Fig. 1. The circuit diagram in Fig. 2 shows how the signal from the digital encoder is processed to obtain a two-digit display indi- cation. Two Schmitt triggers and associated R-C networks ensure sufficient debouncing of the switch pulses. An up/down counter keeps track of the actual switch position. The counters are reset by network R 1 -C 1 at power-on. Two counter types are stated in the circuit diagram to provide either a hexadecimal (0-FF) or a decimal FULL CYCLE PER DETENT (Normally Open in Detent Shown) Channel A CW etektor India September 1987 9“1 25 The speed of DC motors is rela- tively simple to control. For independently energized motors, the speed is, in prin- ciple, a linear function of the supply voltage. Motors with a permanent magnet are a sub- category of independently energized motors, and they are often used in toys and models. In this circuit, the motor supply voltage is varied by means of pulse width modulation (PWM), which ensures good efficiency as well as a relatively high tor- que at low motor speeds. A single control voltage be- tween 0 and +10 V enables the motor speed to be reversed and varied from nought to maximum in both directions. Astable multivibrator ICi is set up as an 80 Hz oscillator, and determines the frequency of the PWM signal. Current source Ti charges C 3 . The sawtooth voltage across this capacitor is compared with the control voltage in IC2, which outputs the PWM signal to buffer N1-N3 or Na-Ns. The darlington-based motor driver is a bridge circuit capable of driving loads up to 4 A, provided the run-in current stays below 6 A, and sufficient cooling is provided for the power transistors T2-T5. Diodes D 2 -D 5 afford protection against 9-126 elektor tndta September 1987 541T80VK HSS8618 A Y m. •-< 1 mu (0-99) display, The displays and associated drivers can be omit- ted if the Q0-Q3 outputs on the counters are used to drive a computer input port direct. The maximum current con- sumption of the 2-digit version is about 400 mA, which goes mainly on account of the 7-segment displays. TW Bourns Electronics Limited • Hodford House • 12/27 High Street • Houndslow • Middle- sex TW3 1TE. Telephone: (01 S72) 6531. PWM DRIVER FOR DC MOTORS inductive surges from the motor winding. Switch Si makes it possible to reverse the motor direction instantly. St 10 mA 1C 2 (7)555 ICI (7)555 BD 679 87462 y° 4 i ■ 2 x 1N4001 m ^ ^ A ^02 fC j) 05 i ^ 2 x 1 N 4001 ^ 1--- • ■ < bk° 3 ((- SAMPLE & HOLD FOR ANALOGUE SIGNALS Conventional analogue sample and hold circuits are notorious for their tendency to drift, a phenomenon unknown in digi- tal memories. It is, therefore, interesting to study the use of a digital memory element for storing an analogue signal. The present circuit is based on intermediate storage of digi- tized analogue information, and therefore requires an analogue- to-digital converter (ADC) at the input, and a digital-to-analogue converter (DAC) at the output. Unfortunately, DACs and ADCs are typically expensive compo- nents, and the present circuit is therefore set up with a DAC only, driven by an up/down counter— see Fig. 1. The coun- ter is essentially an ADC, since the output voltage of the R-2R based DAC is continuously compared to the input voltage with the aid of a window com- parator. The error signal pro- duced by the comparator ar- ranges for the counter to count up or down, depending on the magnitude of the difference I between the input and output voltage. The up/down counter is corrected until the input and output voltage are equal. The digitized result of the A-D con- version is available at the counter outputs. The extensions for converting the basic set-up into a sample & hold circuit are relatively simple. The current count is retaine d by activating the HOLD input, which enables halting the U/D counter. Evi- dently, the counter state is not subject to drift, so that the analogue output signal is available unaffected for as long as the circuit is powered. The converter used here is the Type ZN435 ADC/DAC from Ferranti. This chip contains everything shown in the dashed box of Fig. 1. With reference to the practical circuit diagram, Fig. 2, the internal voltage reference and the oscillator are adjusted with Ri-Ci and R?-C 2 respect- ively. The latter are dimen- sioned for 400 kHz, ie., nearly the maximum oscillator oper- ating frequency. The internal counter is controlled via inputs -Of A = Urel = 2.55V I = digital II - analogue up, down and mode. The logic level applied to the mode input determines whether the coun- ter continues or halts upon reaching state 0 or the maxi- mum value, 255. In the present application, the counter is halted. Gates Ni and N 2 are added to enable blocking the U/D counter. Opamps A 1 -A 2 form the window comparator. Current source T 1 -R 7 and Re arrange for the toggle threshold I of At to be 20 mV higher than that of A 2 . This off-set creates the window, or inactive span, needed to suppress oscillation of the counter’s LS bit, and to prevent unwanted effects aris- ing from the comparators' offset voltages. Decoupling capacitor C 3 is fitted for suppressing spikes that occur during state changes on the counter out- puts. The conversion time of this design is about 640 > 1 $, as determined by the oscillator frequency (400 kHz), the resol- ution (8 bits) and the input voltage change (2.55 V PP max.). This corresponds to a slew rate of 4 mV/^s at the input. Finally, bear in mind that the output impedance (ICi, pin 11) is rela- tively high at about 4 kQ. TW clektor India September 1987 9-1 27 105 ELECTRONIC SAND-GLASS This electronic version of the reversible sand-glass uses a set of LEDs to simulate the passing of sand grains from the upper to the lower bulb. The simple to build circuit is accurate enough for most domestic timing appli- cations. The circuit diagram appears in Fig. 1. On power-up, shift regis- ters IC3 and IC4 are reset by the low pulse from network R35-C7. A few seconds later, the sand- glass is started. The oscillator in ICj generates a clock signal for the shift registers. The clock frequency is adjustable with Pi. Switch S 2 enables selecting one of the three timing periods stated in the circuit diagram. Si is a small mercury or ball changeover switch mounted inside the sand-glass. When this is reversed, the switch toggles and so selects the odd or even numbered LEDs. As- suming that Si is set as shown in the circuit diagram, every clock pulse causes a logic high level to be shifted into IC3, for as long as pin 13 of IC 4 remains logic low. The MS bit of IC3 (output 07) is shifted into the second shift register, IC4. Controlled by the shift register outputs, tran- sistors Ti. . .Tie incl. switch off the odd numbered LEDs, and light the even numbered ones sequentially. When pin 13 of IC 4 goes high, counter IC 2 is reset via N 1 -N 2 , while oscillator N 4 is started. Buzzer Bzi is actuated and sounds for about 2 seconds (C4-R41). The pitch of the tone can be set with P 2 . When the sand-glass is re- versed, Si toggles, ending the reset state of IC2. Logic low levels are shifted into IC 3 because pin 13 of IC4 is logic high. The even numbered LEDs are extinguished one by one, and the odd numbered ones are illuminated, until pin 13 of IC4 goes low again. IC2 is reset, Bzi produces a short beep, and the sand-glass can be reversed for a new timing period. LED D33 indicates that the sand-glass is operative. The circuit is fed 8-128 elektoi india September 1987 Resistors ( ± 5%): Ri . . . Ri6 incl.;R«o = 330R R i 7 . . . R32 incl.;R 37 ;R 39 = 10 K R33 = 100 K R3n;R3i;R3»;R3i — 1M0 Rs«=1K0 Pi = 100K preset P 2 = 2K5 or 2K2 preset Capacitors: Ci = lOOp; 16 V; axial C2;C3;C$;Ce — lOOn Ci = 2p2; 16 V; radial C7;C* -- Ip Ce = 220n Semiconductors: Di . . . D33 incl. = red LED D 34 . . D 49 incl. - 1N4148 T 1 ...T 17 incl. = BC547 ICi = 4093 IC2 = 4060 IC 3 ;IC4 = 74HCT164 ICs = 7805 Miscellaneous: Bzi = PB2720 (Toko; Cirkit stock no. 43 27201). 51 = SPOT mercury, ball or tilt switch, e.g. Maplin order no. FE11M, or ElectroValue no. 339 881. 5 2 = single-pole, 3-position rotary switch plus knob. PCB Type 87406 (available through the Readers Services). Suitable ABS enclosure. DC power socket. from a small mains adaptor capable of supplying about 200 mA at an output voltage between 7.5 and 12 VDC. Construction of the sand-glass is straight-forward using PCB Type 87406— see Fig. 2. The position of the LEDs on the front panel of the enclosure is shown in Fig. 3. Make sure that each LED is connected to the corre- sponding soldering island on the PCB. SPOT Switch Si is made from two SPST mercury or ball switches, fitted together but mutually reversed at a suitable position in the enclosure. The action of the switches is tested by reversing the sand-glass and measuring the switch configuration with the aid of a continuity tester or an ohm meter. All parts in the sand-glass enclosure should be fitted securely in view of the reversibility of the enclosure. The socket for connecting the adaptor, and rotary switch S 2 , are fitted in one of the side panels. A prototype of the elec- tronic sand-glass is shown in Fig. 4. The detachable front panel that holds the LEDs was cut from perspex sheet. R elektor India September 1987 9-1 29 DIVIDER CASCADE N1...N3s V» IC1 = 74HC(T)04 FF 1, FF2 = IC2 = 74HC(T)73 750 mV 9-1 30 elektor india September 1987 This circuit can be driven either with an analogue, or a digital, precision 10 MHz signal for dividing down to a number of commonly used timebase periods. The oscillator pro- posed in hi is particularly suit- able for driving the present cascade, since it offers excel- lent stability thanks to the use of a 10 MHz quartz crystal fitted in an electronically controlled oven. It should be noted, how- ever, that its output is digitally compatible, so that components R 1 -C 1 and R 2 at the input of the circuit shown here can be omit- 107 Refrigerators dissipate the heat extracted from the inside via a grid structure mounted at the rear side. When a refrigerator is located in a confined space, the rear side can get fairly hot owing to the limited convec- tion. This problem derates the overall efficiency of the re- frigerator, since the motor is automatically switched on for longer periods when a con- siderable difference exists between the inside and outside temperature— notably on hot days it often seems as if the motor is running continuously. The ventilation control de- scribed here can help econ- omize on power consumption. The circuit is simple, and does not require a detailed descrip- tion. A simple DC supply is set up with Tri-Di-C,. Temperature is measured with the aid of bridge circuit R 1 -R 2 -P 1 and a NTC (negative temperature coefficient resistor). ICi is a comparator which converts the bridge output into a gate cur- rent for triac Tri, which controls extractor fan M. Some hyster- esis is provided by feedback resistor R3. The triac is con- trolled with a direct gate cur- rent to avoid triggering prob- lems arising from induced volt- age peaks. The circuit is uncritical as regards construction. Be sure to observe the correct connection of the al and a2 terminals on the T1C206, else it remains trig- ted, i.e., Ni is driven direct. Where an analogue, sinusoidal, 10 MHz signal is used, the amplitude must be 750 mVpp. Evidently, R 1 -C 1 and R 2 are then required to make the signal digitally compatible for clocking IC3. The circuit diagram shows that the cas- cade can be extended by adding further 74HC(T)390s and pairs of bistables. The Type 74HC(T)390 (ICs) holds two counters, the first of which divides by two (IQa), and by 5 (10c). Bistable FFi is driven with the IQd output, and out- gered permanently. It should be noted that the circuit is dangerous to touch, as it is con- nected direct to the mains. It is possible to reduce the stand-by puts the :10 signal, which is also applied to the CLK inputs of the second counter in IC3. This also divides by 2 and 5, while FF 2 gives a total division factor of 100 in the first block of the cascade. The use of decade counters results in output periods commonly used for an oscilloscope timebase. Coun- ters and bistables may be added to obtain relatively long, yet accurately defined, periods for specific applications. The current consumption of the cir- cuit as shown is about 12 mA. With two divider blocks added, current by omitting Trt, and powering the circuit in parallel with the refrigerator motor. The NTC should be fitted near the grid at the rear side of the the total current drain is expected to be approximately 25 mA, not 36 mA, since HCMOS circuits require less power at lower clock fre- quencies. D Reference: hi Oven-compensated oscil- lator. Elektor Electronics, January 1986. 87515 refrigerator. The tnac can do with without a heat-sink. TW FORCED COOLING FOR REFRIGERATOR elektor india September 1987 9"1 31 FRONT-END FOR SW RECEIVER There are many conflicting technical requirements for a good-quality front-end in an SW receiver. The noise figure and the intermodulation level should be low, the RF insulation between ports LO, RF and IF should be high, and some amplification is desirable. The Type SL6440 high level RF mixer from Plessey ensures a noise figure of around 10 dB, and offers sufficient sup- pression of the LO signal. The signal applied to the RF input (B) of the front-end is passed through a low-pass filter with a cut-off frequency of 32 MHz and an output impedance of 500 Q. The open collector output of mixer ICi has a relatively high impedance, which necessitates the use of Tri and Rs for correct matching to 48 MHz crystal filter FLi. The fixed impedance of this filter for signals outside its pass-band helps- to keep the intermodulation distortion low. Trimmers Cn and Cm are aligned for a maximum flat pass-band at minimum loss. The mixer’s intermodulation charac- teristics can be optimized by careful dimensioning of Ri and R 2 , provided the amplitude of the local oscillator signal is see text stable. A third-order intercept point of 33 dBm was achieved in a prototype. The mixer IC gets fairly warm, and should be cooled with a heat-sink. The RF transformers are wound as follows (use 30SWG enam- elled wire): Trc the primary winding is 10+10 bifilar turns, the second- ary is 10 turns, on a Type T50-12 ferrite core. Tr 2 : the primary winding is 2 turns, the secondary 18 turns, on a Type T50-12 ferrite core. Le: 6 turns through a ferrite bead. B COMPUTER CON- TROLLED ENLARGER This circuit is intended for anyone lacking a darkroom timer, or being less than satis- fied with it, but in possession of a computer. It is assumed that the computer has a built-in relay for controlling the motor in a cassette recorder. Where this relay output is not available, a different method needs to be adopted for driving the opto- coupler in the enlarger circuit described here. An MSX computer can actuate the relay in question with the aid of BASIC command MOTOR ON. This causes the phototransistor in opto-coupler ICI to conduct, so that Ti can trigger silicon-controlled recti- fier Trii. The enlarger lamp remains on until command MOTOR OFF is issued. Zener- diodes D 1 -D 2 ensure a safe operating voltage for Ti. Switch Si enables turning on the lamp for position adjustments. Writing a program for the en- larger control should not be too difficult, and is therefore left to your own ingenuity. The circuit should be fitted in a properly earthed metal case or an ABS enclosure to prevent accidental contact with points at mains potential. R 9 _ 1 32 eleklor India September 1987 SW PRODUCTS • NEW PRODUCTS • NEW MULTIPOINT SCANNER Multipoint Process Scanners are a logical extention of the Series 4500 high precision 4'A digit Process Monitors. When linked with transducers, the Autoscan 8000 finds its applications in the laboratory. Quality control, material test stations or in the field and serves as tool for Centralized Data Collection. The Autoscan 8000 allows upwards integration and when linked with PRT 8421 Numeric Printer the instruments established a Protocol with the Print Mechanism and serves as a complete Datalogging system. Standard features include Automatic/Manual Selection, skip-channel. Hold, channel select, scan rate selection, independent peak readout and peak value scanning. The AUTOSCAN is supported by a wide range of options and accessories that include PT -1 00/Thermocouple linearization and selectable digit alarms with two or more setpoints per channel. For further information please contact: ACCORD ELECTRONICS 201 Yashodham Enclave SA-1/56, Goregaon Mulund Link Road, Goregoan (East) Bombay 400 063. PANEL METERS In addition to their existing range of DIN size 72mm, 96mm and 144mm Square 240° Circular Scale Indicating Panemeters, MECO have now introduced another 240 Circular Scale Panel Meters with a clear transparent Acrylic Square front of 1 10 x 1 1 0mm and a round Phenolic moulded body of 100mm diameter. Ammeters & Voltmeters (both AC as well as DC) Frequency Meters, Wattmeters, Power Factor Meters & VAR Meters are available in the above Model. The above instruments have been developed primarily for defence use. These instruments are mainly used in the control panels installed i in Power Stations. For further information please contact: MECO INSTRUMENTS PVT. LTD. Bharat Industrial Estate T.J. Road, Sewree Bombay 400 015 Phone No.: 4137423 4132435,4137253 Telex: 11 71001 MECO IN Cable: STANCOR Bombay 15. TOOL KIT This is a new double zipped, padded wallet which contains a range of electronic aids to handle various jobs. The wallet has 23 fitted pouches to hold the items securely. The kit includes light weight cushion gripped 1C cutter. Flat Nose Plier, Micro probes-minipulators. Desoldering gun, knife with 4 replacable blades of different designs, Tong., Claw, Fibre grip heat resistant device, heat shunt, Component handling tweezer, Triangular file, PCB Brush, Hand held drill, Wire Stripper, Pin Vice, Doofer, Magnifying glass, Scissor, Retrievable Device and 1C Insertion And Extraction device. External dimension of the wallet when closed are: 12" x 9" x VA" and weighs 850gms only. For further information please contact: KOHL I TECOMA Stereo House, 46, 1st Marine Street, Dhobitalao Bombay 400 002. (India) Phone No. .319128, 8121167. MILLI-OHM METER The VRM22M with its four terminal measurements and 0.1 milli ohm resolution, makes low resistance measurement very simple and accurate. Being BATTERY/ MAINS operated and light weight it can be carried to the work spot, VRM22M has 3'A digit 0.5 inch bright LED display which is ideal for both factories and labs. In most cases it replaces the KELVIN DOUBLE BRIDGE. For further details please contact: VASA'JI ELECTRONICS ( Marketing Division) 630 Alkarim Trade Centre Ranigani Secunderabad 500 003. Phone No: 70995. MULTIMETER LEDTRON ELECTRONICS has introduced Pocket size analog Multimeter model 1015B in association with M/s. Hung Chang Products, Korea. It can measure DC/AC voltage upto 1000V DC current upto 250MA, Resistance upto 10 Megohm, decible from— 20 db to 62 db and battery check for 1 .5V and 9V batteries. Other features are a 10000 Ohms/ VDC sensitivity and 90° Arc mirrored scale for accurate reading. For further information please contact: LEDTRON ELECTRONICS 1 70 Lohar Chawl Bombay 400 002. 9-1 34 elektor intita September 1987 PRODUCTS • NEW PRODUCTS • NEW DATA SCANNER Advani-Oerlikon have developed a mini microprocess-based data scanner called UDS-30. This 30-point scanner is designed for scanning of temperature, voltage or any other parameters of water and steam boilers, windings of HP motors and high voltage transformers, distribution points in silos containing foodgrins, engine test and reaction vessels in chemicals and process industries. The system is field proven, versatile and compact. It is mounted in a standard RA 19 rack. It can accept multi-variable inputs such as Thermocouples, RTDs and Analogues. The system has built-in 24 columns, an alphanumeric 2 colous printer with re-rolling facility which gives out print out of scanned data and programmed parameters. The keyboard functions such as low level set point, control level set points, dwell time, high level set point, channel number, hysterisis, etc. are programmable individually for each channel. Display annunciation is provided for each channel. There are totally 90 LEDs. Each channel has a seperate indication for alarm, senor break and control status. The system also has the facility to scan alarm conditions on a priority basis. Output relay contacts are provided for each channel. One relay is provided for common alarm and one for senor break indication. EEPROM memory is used and hence no battery back up is required for the programme. A real time calendar is also provided which gives date, month, year, day of the week and time. Nickel cadmium battery is provided for the back-up of the calendar. The system uses a floating point arithmetics for linearisation and other mechanical calculati ons. Solid-state semiconductor switches are used for multiplexing, thus contributing to reliability and compactness. STD cards are used for flexibility of operation and ease of maintenance, thus ensuring minimal downtime. The plug-in PCB and the STD mother board have minimised wiring in the instrument. The unit has a hinged transparent unbreakable cover on the front space to avoid any accidental changes in the keyboard function. For further information, quote ref: PUB/2, contact: ADVANI-OERLIKON LIMITED Post Box No. 1546 Bombay 400 001 SPECTRUM ANALYSER ROFIN-SINAR LASER UK LTD, announce the introduction of the high speed RSO 6240 Spectral Processor to operate with the current line of Optical Spectrum Analyser equipment. The new instrument includes a more powerful processor, together with many system improvements such as dual double-density double-sided 3’/a" disc drives, an improved monitor, and digitising electronics. The entire system has been repackaged with an integral keyboard instead of the earlier seperate keyboard. In addition various accessories and software packages have been added to provide a very powerful package to measure transmission, absorption, reflection and colour, in addition spectroradiometric and software package. The system captures a complete spectrum in 5 msec and stores it in 80 msec in the processor. The wavelength range is 200-5000 nm which can be covered at one time using the "merge" software facility. For further information, contact: TOSHNI-TEK INTERNATIONAL 267 Kilpauk Garden Road - Madras 600 010. THERMOCOUPLE VACUUM METER The IBP Thermocouple Vacuum Meter is a simple, single head measuring device. SPECIFICATIONS Gauge Head: Chromium plated brass with octal socket. Vacuum Connections: Through standard 6 mm screwed union. Measuring Range: 1-1000 Microns. Calibration: Calibrated for dry air using a Mcleod gauge. Power Supply: 230 Volts, 50 Hertz, ± 10%. Dimensions: Small, compact construction with simple panel installation in Standard half module (H 135 mm x W 210 mm x D 145mm) Standard accessories supplied: Gauge head with cable of length 3 Metres. Applications: Used in Industrial Systems, Refrigeration Industry, Flask, Lamps, Capacitors and Condenser Industries etc. For further information please contact: IBP CO. LIMITED A Govt, of India Enterprise Engineering Division Sewri (East) Bombay 400 015. "ROCKER TOGGLE" SWITCH I EC has just introduced a range of "Rocker Toggle" switches with Black, Red, Blue, White, Yellow or Green colour knobs. These Rocker Toggle switches are available in 6A, 10A, 15A, 250V AC or 28V DC in single and double pole with on-off, changeover with or without centre off and momentary contact, to serve as Push Button. Special circuits are possible e.g. 1,2,3 or 1, 1+2, 2+3, etc., avoiding the need of 2 or 3 switches. Switches are supplied with screw terminals or push-in terminals (6.3mm). For further information please contact: INDIAN ENGINEERING COMPANY Post Box 16551, Worti Naka Bombay 400 018. 9-136 eleklor india September 1987 COLOUR TV SPEAKERS □ixco TWEETERS 6 LT 15 6 Lrio 5LT10F ''•OTlOs AVAILABLE IN 4. 8. & IB OHMS. OTHERS ON SPECIAL REQUEST LARGEST MANUFACTURER OF SPEAKER COVERING AUDIO. VIDEO. TOYS. TELECOMS & COMPUTERS O Manufactured by: LUXCO Electronics Allahabad-211 003 □ Sole Selling Agents: LUXMI ft CO. 56. Johnstonganj Allahabad-211 003 Phone: 54041. Telex 540-286 □ Distributors for Delhi & Haryana: Railton Electronics Radio Place. ChandniChowk Delhi-110 006, Phone: 239944, 233187. □ Distributors for Maharashtra. Gujarat and South India: precious® Electronics Corporation • Chotani Building. 52. Proctor Road. Grant Road (East). Bombqy-400 007. Phones: 367459. 369478 • 9. Athipattan. Street. Mount Road, Madras— 600 002. Phone: 566718 sound technology from a sound source B/W TV SPEAKERS etoktor india September 1987 9* 1 69 NEW Includes information on: • Home work -Table. • Hardware & Components • Checklists • Soldering • Component data & codes. 50I' • Send a 3 year Subscription for elektor India masazine and set this book free. • Existing subscribers can order this book @ 30% discount. Please mention Subscription number (or send latest address label from magazine envelope) I Please send amount by M.O7D.07P.O7 only to: eIeI