up-to-date electronics for lab and leisure s™""

LCD thermometer
short wave band shifting
ultrasonic yardstick
active antenna
electrolytics run dry

selektor

10-14

DSB demodulator

After our crash course in transmission and reception, we introduce another
type of modulation: DSSC or more commonly known as DSB.

L.C.D. thermometer

We present a digital circuit, which is inexpensive, accurate, precise, and has
a very low power consumption. The range of the instrument is such that it
is suitable for practically any application.

ultra sonic distance measurement

An electronic method for the measurement of distances, using ultra-sonic
sound, A very good starting point for experimentation.

electrolytics run dry

The article describes and discusses the merits of the different kinds of elec-
trolytics, arriving at a surprising conclusion.

darkroom computer - part 2

A continuation of the first part in the last issue. This article deals with the
accessories needed to make the computer fully operational.

short wave band shifting for SSB receivers

The article deals with and describes front ends which can be used with any
short wave receiver, effectively extending the coverage of the amateur
bands.

16 channels with only five ICs

A straightforward easily constructed infra-red remote control. Construc-
tion is made easy by using special ICs produced for the purpose.

pre-amp for the SSB receiver

An additional MOSFET pre-amp not only increasing the sensitivity, and
improving selectivity, but also extending the AGC range.

active aerial

An active aerial which is short and convenient to use, whilst enabling good
reception. What more would DXers wish.

transistor and 1C data

comprehensive information on a selection of CMOS ICs and transistors,
very useful aid.

missing links

10-17

10-20

10-24

10-28

10-30

10-36

10-40

10-44

1046

10-49

10-54

market

10-54

10-14 - elektor October 1982

Power from the invisible universe

A theory linking black holes, quasars
and radio jets in galaxies may explain
the basis of the universe and point to
its eventual end in a so-called Big
Crunch.

Only a few years ago it seemed that
astronomers were drawing near to a
comprehensive understanding of the
universe. Mysterious objects remained,
notably the quasars with their extra-
ordinary brightness, but a pattern was
emerging that promised to encompass
all the categories of stars and galaxies
that astronomers could see.

To some extent that is still true today.
But what is becoming apparent is that
the universe we can observe with any
kind of radiation, including radio
waves, infra-red, ultra-violet and X-rays,
is probably only a small fraction of the
entire universe. Most of it is invisible
because the matter it comprises pro-
duces no radiation for astronomers to
• detect with their telescopes.

But this unseen matter cannot be
ignored. Although it is invisible, it is
probably responsible for the most
violent events in the visible universe.
And its presence may be the factor
deciding the fate of the universe,
whether it will go on expanding for
all eternity or whether it will instead
stop expanding, contract and finally
disappear in a Big Crunch, the precise
opposite of the Big Bang in which it
came into being.

Quasars and Black Holes
The two most mysterious objects in the
known universe are quasars and black
holes. Quasars are stange super-stars,
each one shining more brightly than
thousands of complete galaxies of
stars, yet producing its radiation from
something with a diameter no greater
than that of the solar system. Because
quasars shine so brightly they are
visible at distances of several thousand
million light years. This means that we
are seeing them not as they are now but
as they were several thousand million
years ago, when the universe (which
began at some time between ten and
fifteen thousand million years ago) was
relatively young. This has suggested to
astronomers that, because no quasars
to be nearer at hand, in more recent
epochs, whatever provides the power
for them may be some force which is

Black holes the fates of large stars when
their nuclear fuel finally burns out.

When this happens there is no longer
outward explosive force to balance the
inward gravitational forces caused by
every particle of matter in the star
attracting every other particle of matter,
so the star collapses inwards and be-
comes more and more dense. It passes
through the stage where the electrons
orbiting atoms are squashed inwards
to fuse with protons, the negative and
positive charges cancelling each other,
thereby forming neutrons. Smaller stars
stop contracting at this stage and form
so-called neutron stars made of col-
lapsed atoms. They are dotted around
the universe and can be detected by
their gravitational effects on other stars.
But in larger stars the inward gravi-
tational forces are so strong that theory
predicts they will go on collapsing
inwards until all the matter in such a
star has been compressed into a dimen-
sionless point. A star which has reached
that stage can no longer be observed
because no radiation can leave it. For
this reason, and because events inside
it can no longer be understood through
conventional physics, it is called a
black hole.

Radio Jets

Black holes are no longer just theoreti-
cal notions. X-ray telescopes have
detected radiation coming from two or
three objects which are very probably
other stars in orbit around black holes.
More recently, there has been growing
suspicion that violent events in the
centres of galaxies may be caused by
black holes being formed there. In the

UK, Professor Martin Rees of the
Cambridge Institute of Astronomy and
his colleagues have put forward a theory
which not only explains quasars in
terms of black holes but explains
another mysterious phenomenon, too,
so-called jets in radio galaxies.

Radio galaxies, the main class of ob-
jects detected by radio telescopes,
appear usually to contain an ordinary
optical galaxy with a characteristic
disc shape. Protruding from the centre
of the disc in each direction perpen-
dicular to it, giving the appearance of
a wheel on an axle or of a spinning top,
are jets of radiation and charged par-
ticles shooting such colossal distances
into space that they dwarf the galaxy
from which they come.

To understand Professor Rees's theory,
it is necessary to go back in time at
least ninetenths of the way to the
origin of the universe, to a period
about one thousand million years
after the Big Bang. At that time, swirl-
ing clouds of gas were separating from
each other and forming spinning, proto-
galactic discs with solid stars condensing
out of the gas. As the stars formed, the
remaining gas sank towards the centre
of each galaxy, attracted by the pull of
the denser matter. There were no
forces, other than those which mutually
repel neighbouring atomic particles,
to slow down the inward collapse of
the gas into each galactic centre. So
collapse soon proceeded to the point
where a black hole formed. More and
more matter was pulled into the black
hole by its enormously powerful gravi-
tational field, so its pull grew still

October 1982

The double radio source 3C236. Contour lines map the intensity of the radio emission showing
the source to have radio lobes' (at the bottem left and top right of the map) separated by
20 million light-years.

stronger until it drew neighbouring
stars into itself, as well as dust and gas.

Synchrotron Radiation
This, says Professor Rees, is the stage of
galactic evolution represented by the
quasars. They are young galaxies in
which the central black holes are still
sucking in stars. While the stars are
being pulled into the black hole, before
they disappear for ever, they are accel-
erated to velocities near to the speed of
light. Matter accelerated to such speeds
gives off powerful radiation, called
synchroton radiation (see Spectrum
1291. The extraordinarily powerful
synchroton radiation we observe from
quasars is on the scale to be expected
from a black hole scouring the centre
of a galaxy cleaned of millions of
stars. When there are no more stars
left near enough to be swallowed, the
quasars revert to the radiative power of
ordinary galaxies. There are no quasars
today because there is no matter left
in the centres of galaxies to be swallowed
by the black holes that lurk there.
No other phenomenon is powerful
enough to produce the quasar radiation.
But, says Rees, that is not the end of
the story. Because the galaxies in which
they are formed rotate, the black holes
in their centres rotate, too. And because
they are made of infinitely dense matter
the black holes rotate virtually as fast as
it is possible to rotate. They spin at near
to the speed of light. Surrounding the
centre of each galaxy is a magnetic field
which is formed, predictably, in gas that
is too far away from the black hole
to have been sucked into it. The
magnetic field in the centre of such a
galaxy pulls on the spinning black hole,
acting as a persistent brake upon it.
When a brake is applied to any moving
object energy is lost; one example is
the heat given off when wheels are
braked. When a brake is applied to a
mass of a hundred million or so stars
in a black hole spinning at a speed of
near to 186 000 miles/second, we may
expect the energy produced to be on
an unfamiliar scale.

This energy, Rees proposes, is the
explanation of the jets in the radio
galaxies. The magnetic fields that
surround the spinning black holes have
narrow tunnels running through them
to each side, perpendicular to the
galactic plane, and the colossal energy
produced from the braking effect of
the magnetic field on the black hole
is flung out through the tunnels. It
emerges, shooting out through space.

as jets of particles and radiation
dwarfing the galaxy they come from.
Professor Rees believes that the evol-
ution of galaxies into the quasar stage
and onwards, with the formation of
spinning black holes and radio jets, is
probably a very common, perhaps
almost universal process. Most galaxies,
including our own, have gone through
it. Spinning black holes may form
from single stars as well as from millions
of stars in the centres of galaxies.
Single-star black holes would produce
miniature radio jets. Two objects with
structures that resemble or suggest the
existence of such jets are known,
namely the bright X-ray star
Scorpio XI and the optically visible
source SS433.

Violent Events

So this theory explains and links the
three strangest phenomena in the known
universe, black holes, quasars and radio
jets. But it implies that we must look
for the explanation of the most violent
events in the visible universe in matter
which can no longer be observed in
that universe, in black holes. And if
we are to predict the fate of the uni-
verse, say Rees and certain other as-
tronomers, we must again take into
account black holes and other categories
of invisible matter.

It is possible for astronomers to cal-
culate how much matter there would
need to be in the universe for the
gravitational forces pulling it together
eventually to overcome the outward
forces of expansion, so that the universe
will eventually stop expanding and

start to contract towards a Big Crunch.
For this to happen, the universe in its
present stage of expansion needs to
contain an average of about three atoms
rules out the crunch, because all the
matter in the universe that has been
observed by any kind of telescope
adds up to only about one-tenth of
one atom per cubic metre. But, says
Rees, there is a lot of matter in the
universe that we have not seen and can
never see. And there is growing evidence
that this invisible matter greatly out-
weighs the visible.

Some galaxies orbit around one another
in pairs, like double stars. Their masses,
calculated from their orbits, are es-
timated to be at least ten times that
of their visible matter. This is still only
one third of the mass needed to ensure
a Big Crunch, but it does indicate that
only about one-tenth of the matter in
galaxies can be detected, with any
kind of telescope. Some of the rest
may already have disappeared for ever
into black holes. Perhaps as much as
nine-tenths of the matter in the universe
has gone that way. More invisible matter
may be in the form of 'dark' stars,
which have burnt out without collapsing
into black holes but which no longer
emit enough radiation to make them
observable.

The amount of invisible matter which is
present in a third form, particles called
neutrinos, may dwarf that present in the
visible universe.

In the first fraction of a second after the
Big Bang, matter and its opposite, anti-
matter were probably created almost
equally. They promptly annihilated

IO- 16 -i

each other, leaving a relatively small
residuum, representing the margin of
matter over anti-matter, to become the
universe we see today. Particles called
neutrinos were produced by each
annihilation of particles.

Because neutrinos hardly ever interact
with other particles, those formed
just after the Big Bang have been
flying around the universe ever since.
Until recently it was thought that
neutrinos had so mass. Now some
astronomers think that neutrinos do
have mass and, although it is very small
compared with the mass of other
particles, there may be so many neu-
trinos that their combined mass dwarfs
that of all other matter in the universe.
It could be much more than enough
to ensure a Big Crunch. The problem

is how to detect neutrinos. Attempts
to trap them, by burying detectors
deep underground to screen out the
impacts of other particles, have so
far been unfruitful.

If these theories are correct, it is rather
sobering to think of how unimportant
our familiar universe of stars and
galaxies is, compared with the invisible
black holes which power all its most
violent events and which may have
already swallowed most of its matter,
and with the invisible sea of neutrinos
on which our familiar matter may
float like trivial froth on the oceans.

(813S)

Video recording at over 600 miles
per hour

Unique video coverage of the attempt
on the World Land Speed Record by
Richard Noble in his car Thrust 2' will
be provided by cameras and video
recorders from Sony Broadcast.

A lightweight video colour camera will

Structure of the radio source around the galaxy NGC 6251, with detail magnified in two

stages. The parsec, a unit of stellar distance, is equivalent to 3-26 light-years.

be installed in the spare cockpit of the
jet-powered car during the attemps to
break the world record of 622 mph
on the Bonneville Salt Flats in Utah,
U.S.A.

Sony equipment is being used by
Intervideo Productions Ltd. to record
a documentary covering the 7 year
development programme of Thrust 2',
the people involved in the project and
all the activities during the attempt in
Utah. Three main camera teams will
record the activities at base camp, the
refuelling point at the end of the
runway and from the air in a hot air
ballon or a light aircraft. In addition,
the camera installed in the car itself
will provide dramatic shots of both the
car's instrument panel and scenes of the
runway as viewed by the driver.

To break the record. Thrust 2' has to
make two passes over a measured mile,
in opposite directions, within an hour.
From standing start and acceleration
through the measured mile and then
deceleration will take something like
45 seconds over a distance of 10 miles.
At the point of braking considerable
G forces of up to 6.5 G will be ex-
perienced by the driver, but it is not
yet known how the video camera and
recorder will function under these
conditions.

The camera's weight factor combined
with the vibration from both the move-
ment of the car over the salt flats plus
high frequency vibrations from the
Rolls Avon 403 jet engine (which pro-
duces some 17 tons of thrust) creates
a very critical environment for both
camera and recorder in terms of micro-
phony.

The weight of any item of equipment
is multiplied by the G force applied.
This means that the Sony camera, which
weighs 10 lbs under normal gravity
conditions, will have an actual weight
of over 60 lbs on deceleration from
650 mph.

At present the camera is providing vital
telemetry information on the behaviour
of the car as well as an accurate record
of the instrumentation read-out from
the spare cockpit.

Richard Noble will be attempting to
break the World Land Speed Record
later this year at a speed in excess of
650 mph.

Sony Broadcast Limited

(806 S)

DSB demodi

ilektor October 1982 - 10-17

In the June 1982 issue we
presented an article entitled "The
principles behind an SSB receiver"
and we hope that it provided
answer to some questions. As a
practical complement to the
"Crash course in transmission and
reception" we then presented a
complete SSB short-wave receiver
for home construction.

With this article we would now
like to introduce another type of
modulation that was mentioned:
DSSC (double sideband, sup-
pressed carrier) or, as it is more
commonly known, DSB. An
appropriate demodulator in a
new circuit technique is provided
for those who like to become
more involved in DSB in future.

Conventional amplitude modulation
(AM) with carrier should by now be
quite familiar. Short-wave, medium-
wave and long-wave listening is very
common because all the problems of
this type of modulation appear to have
been solved by the invention of the
detector receiver. Nevertheless, for
widely varying reasons, the field of
telecommunications has provided the
impetus to 'invent' many other analogue
and digital modulation methods; these
are no doubt all justified but make
quite different demands on the receiver.

with respect to frequency stability of
the demodulator.

At this stage we would like to present
another example of DSB which we
encounter daily whilst hardly noticing
it. We are referring to the stereo signal
from the VHF receiver. Advanced radio
enthusiasts will immediately point out
that VHF radio operates with FM. This
is true, but let us examine the stereo
signal. It consists of the frequency
band L+R, the pilot tone and two
L-R bands around the 38 kHz sub-
carrier. The carrier is modulated with

DSB demodulator

carrier regeneration using the audio-frequency method

Theory

Nobody really knows why DSB was not
able to establish itself successfully. Some
cynics tell us that vested interests in
SSB carried more weight at the decisive
moment. It is possible that the SSB
technique was better developed at that
time than the DSB technique. In any
case, with respect to efficient use of
transmitter power, the DSB technique
represents a middle route between AM
and SSB modulation (figure 1). If, for
example, a sinusoidal carrier of a fre-
quency of 4 MHz is modulated with a
sinusoidal 'information signal' of a fre-
quency of 1 kHz, two 'side frequencies'
are produced in addition to the carrier
frequency (3999 kHz and 4001 kHz).
This can be proved mathematically but
is beyond the scope of this article.

Figure 1 shows a modulated signal as
displayed on a spectrum analyser. A
spectrum analyser displays the ampli-
tude or, in this case the power level of
a signal, as a function of frequency and
not as a function of time, as is the
case with an oscilloscope. Assuming that
the information signal does not consist
of one frequency but of a mixture of
frequencies, bands are obtained instead
of lines in the spectrum of the modu-
lated signal, one to the left and one to
the right of the carrier. Both bands
contain precisely the same information.
The carrier contains no information but
requires the most power, as can be seen
in figure 1. If the carrier is then sup-
pressed during transmission and the
energy it contains is transferred to the
information-carrying sidebands, the
result is the two types of modulation
DSB (DSSC) and SSB. The advantages
and disadvantages of SSB were already
discussed in the June 1 982 issue. With
DSB the effective, information-carrying
power is doubled with respect to
conventional AM. As with SSB, however,
the carrier must be regenerated at the
receive end. This is quite a challenge

the L— R signal in DSB technique. Thus
the carrier is initially missing in the
receiver. The entire signal is finally
frequency-modulated with the RF
carrier. To avoid any misunderstandings,
we are not designing a new type of
stereo decoder here, but merely wanted
to provide an example of DSB!

Block diagram

Figure 2 shows the block diagram of
a DSB superhet receiver. The input
signal is mixed with an oscillator signal.
The output of this oscillator is some-
what higher in frequency than the
input signal and is tuned simultaneously
with the input signal. In this way the
frequency difference between input
signal and oscillator signal remains
constant over the entire tuning range of
the receiver (455 kHz in this example).
The 'difference signal' is known as the
IF (intermediate frequency).

The output signal of the IF amplifier
is applied to the input of the DSB
demodulator. Mixing then takes place
here too, but this time with a square-
wave signal from the sampler. The result
is an audio-frequency signal from which
the carrier (now an AF carrier) is
regenerated with relatively simple means.
If this carrier is then mixed with the
IF signal previously obtained by sub-
tractive mixing, the AF signal is pre-
sented at the output. The frequencies
given in figure 2 relate to the example
in figure 1 .

The term 'sampler' that appears twice
instead of oscillator is not really rel-
evant in our case. We are more con-
cerned with carrier regeneration than
with the details of AM reception.

The mathematics of the method are
complex and involve many trigonometric
formulae. The result is straight-forward,
however: if two sinusoidal signals, like
the 'side frequencies' that we have
already met, are multiplied, the carrier
is produced with twice the frequency

10-18 — elektor October 1982

together with some other frequencies
which are filtered out.

Practice

In contrast, the method is clearer in
practice. Multiplication of two sinus-
oidal signals is normally performed
using a 4-quadrant multiplier (ring
modulator). We will use a simpler
method, however. Since a perfectly
sinusoidal signal is not needed at the
output, digital multiplication can also
be employed. This merely requires an
exclusive-OR gate; the original signal
and the out-of-phase signal are applied
to its inputs. With a phase shift of
90° a square-wave signal appears at its
output with twice the frequency of the
input signals. This type of phase-
comparator, exclusive-OR gate can
therefore also be described as a digital
4-quadrant multiplier.

Figure 3 shows the circuit of the DSB
demodulator which we shall now exam-

ine step-by-step. T2 performs the
function of the second mixer in the
block diagram. The corresponding
sampler consists of the oscillator with
Fil and T3 and switch T4. All types
of filter which can be tuned to
455 kHz can be used.

Although the first stage of the demodu-
lator still operates at high frequency,
it is followed by audio-frequency
stages. The IF signal obtained by
subtractive mixing is applied to a
potent amplifier via buffer A1. The
signal is then amplified until a 'clean'
square-wave signal is present at the
output of comparator A3. The phase-
shift mentioned is performed by
integrator A4. It is configured so that
this phase-shift takes place between
approximately 10 and 30 kHz. The
'shifted' signal is shaped into a square-
wave signal by comparator A5. Exclus-
ive-OR gate N1 forms the digital 4-
quadrant multiplier. Ignoring for a
moment the PLL circuit IC5, the fre-

DSB demodulator

quency of the output signal of N1 is
then divided down by FF1 to the
frequency of the carrier. The low-pass
filter consisting of R20/P3 and Cl 9
compensates for the 90° phase-shift
caused by the PLL circuit (45 at
fvco/2). Comparator A6 forms a square-
wave signal from the audio-frequency
carrier.

The third mixer mainly consists of T5.
Two signals are applied to it: the audio-
frequency IF signal via low-pass net-
work R1 2/C10 and the signal from the
sampler consisting of FF2 and N2 . . . N3.
This circuit may appear a little strange
at first sight, but it is really not compli-
cated. It consists of a monostable that
is triggered by the audio-frequency
carrier. If a positive pulse appears at
pin 11 of FF2, output Q becomes a
logic 1. After passing through the delay
circuit of N2 . . . N4 a pulse appears
at differentiating networks C20/R21,
but the flip-flop is simultaneously
reset and waits for the next triggering

chiefly

pulse. During this period T5 conducts;
the weighted signal is fed to the active
low-pass circuit of A8 via buffer A7
and the audio-frequency signal is
present at the output (wiper of P5).

The PLL circuit of IC5 performs two
functions. Firstly, of course, it allows
a frequency to be 'locked on' very
precisely. Secondly, this frequency is
also retained when the control voltage
for the VCO has become very low. In
our case this signifies the following:
when viewed on an oscilloscope the
DSB signal looks like beads on a string.
One would expect the carrier to be at
the points of contact between the
beads, but there are none. Since the
amplitudes of the signals from the
mixture of frequencies of the sidebands
are very low in this range, the demodu-
lator does not really 'know' whether a
useful signal is present. In could simply
be noise. Thus, in a manner of speaking,
the PLL circuit buffers the generated
carrier under conditions of fading field
strength, so that the carrier does not
disappear again.

Application and alignment

The DSB demodulator can be utilised

in any AM superhet receiver. Since the

second mixer is a 'harmonic' type, an
IF signal in the range of 455 kHz to
approximately 20 MHz can be processed.
Figure 2 shows the general configur-
ation. The only stage still required at
the output of the demodulator is an
AF amplifier. Clearly the demodulator
also provides advantages in conventional
AM reception.

Alignment does not involve any prob-
lems, because the audible method is
used. First connect the AF amplifier
to the output of A1. Tune the receiver
to a conventional AM transmission
(with carrier) so that a signal is present
at the output of the IF amplifier. Adjust
the core of Fil so that the whistle in
the loudspeaker is just at the audible
frequency limit (about 15 kHz). If
this tone sounds distorted, the mixer is
being overdriven. In that case PI should
be adjusted so that the whistle is barely
undistorted.

In the second alignment step, connect
the AF amplifier to its real location in
the circuit, i.e. to the wiper of P5. Set
P2 . . . P5 to their middle positions.
Then adjust P4 so that the PLL circuit
'locks on'. When a transmission is
received, i.e. when modulation is
present, (the whistle should cease
when the circuit locks on). It may be

necessary to shift the range by adjusting
P2.

In the third alignment step, adjust P3 |
so that the AF signal reaches its maxi-
mum. This is precisely the case when
the carrier is subjected to a phase-shift
of 45° by low-pass network P3/C19.
Finally, the output level can be matched
to the following amplification stage by
adjusting P5. The entire alignment
process should be repeated several
times. In particular, the distortion
caused by selective fading should
cease. Instead a kind of phasing effect
appears.

One thing becomes clear from the
description of the alignment process:
if the demodulator does not precisely
'lock on' to the transmission, an un-
pleasant howling and whistling is
heard from the loudspeaker. For this
reason, one should tune twice: first
in the normal manner with the existing
detector and then precisely with the
demodulator described here. The only
additional controls required on the
receiver are a changeover switch and a
potentiometer. We hope that this
circuit will contribute to an improve-
ment in your short-wave reception. M

10-20 -ele

1982

■C.D. thermometer

The 7106 is a well known 1C in the
world of A/D converters, and was
chosen for three main reasons. Firstly
this 1C is a 'jack of all trades' and is
widely used in all forms of voltage or
temperature measuring instruments.
Secondly, because it is universally
available and relatively inexpensive.
Last but not least, the 7106 and its
big brother (7116), have so many
functions already integrated within
themselves that only a few passive
components and a LCD are needed
to complete a good circuit.

The 7106 contains an A/D converter,
clock generator, reference voltage

dependant upon the time. In turn the
contents of the counter are then dis-
played on the LCD. The advantage
of using this method is that a relatively
simple and straigthforward oscillator
can be applied. The oscillator frequency
of the 1C is in fact determined by the
values of R2 and C3. This frequency
also determines the number of 'samples'
taken in every second. As a matter of
interest, using the values as indicated
in the circuit diagram, three samples
are taken every second.

The 1C ensures a zero setting before
each 'sample', or measurement, auto-
matically. Quite simply, the inputs are

L.CJ). thermometer

source, BCD-to-seven-segment decoders,
and latch and display drivers! Quite a
bundle of energy! And even if this
array of goodies was not enough, it is
also equipped with an automatic zero
correction, and polarity indication.

The 7116 (believe it or not), not only
has everything the 7106 has to offer,
but also includes a hold facility enabling
the read-out to be frozen, if required.
The circuit described here is

designed
to accept either
1C, allowing the con-
structor to decide which
of the two he prefers to use.

The circuit diagram
The circuit as shown in figure 1 is really
nothing more than a digital voltmeter,
which in turn measures the voltage drop
across a temperature sensor.

The dual slope conversion principle is
applied for the voltage measurement.
Basically the input voltage from the
sensor charges capacitor C4 for a fixed
period of time. The capacitor then
discharges, the rate at which the capaci-
tor is discharged being determined by
the reference voltage. The actual time
it takes for the capacitor to discharge
fully (return to zero) is then pro-
portional to the input voltage level.
During the discharge period, pulses
from an oscillator are stored in a
counter, obviously the number of pulses

first of all decoupled internally from the
actual input pins and then short cir-
cuited. The automatic zero capacitor
(C5 in this case) is charged via a separate
feedback loop, so that the offset
voltages of the buffer amplifier, inte-
grator, and comparator are compensated
for, inside the 1C. This guarantees any
measurement really does start from 0 V,
and that when the display reads 000, it
does denote a 0 input voltage.

The temperature measurement stage is
straightforward if somewhat sophisti-
cated. It contains three voltage dividers:
R10 and R11; R8/P1; R9/P2. The
junction of the first divider containing
the sensor R11 is connected to the
'IN HI' input of the 1C. The wiper of
is linked to the
jt and the wiper
the 'REF HI' in-
effect the circuit
measures the differential
voltage between one side of
the sensor and the wiper of
PI. Any measurement is com-
pletely independant of the supply
voltage level, because the reference
voltage of the 1C is also derived from
the supply (via the divider R9/P2).
Keep in mind that a full scale readout
will be equal to twice the reference
voltage. Any decrease in supply voltage
will not change the readout, because
the reference voltage will decrease
by the same amount (when compared
with the measuring voltage that is).
Resistor R4 and capacitor C6 act as
a input smoothing filter.

The display is driven directly by the
1C. The EXOR gate N2 ensures that
the decimal point is activated, by
supplying the inverted backplane signal
to the corresponding LCD points.

The circuit also has a low battery
indication function. The display denotes
this by either an arrow or the term
'Low Bat'. An EXOR gate also controls
this function!

Transistor T1 is used as a supply voltage
level detector. The emitter is connected
to the junction of R5 and R7, and its
base to the test connection of the 1C.
This pin not only allows the display

. . . accurate to 0.1
of a degree

During the past few months, the
Elektor offices have been inun-
dated with requests for a digital
thermometer.

In answer to all these requests,
and to relieve the pressure on our
technical queries department, we
present a digital circuit using a
special 1C and a LCD display.

The design is inexpensive, but,
nevertheless accurate, precise, and
has a very low power consump-
tion! The range of the instrument
is from — 50° C to +150°C. The
temperature is displayed 0.1
degree at a time, therefore making
it suitable for practically any appli-
cation.

itself to be tested (by connecting it to
a +5V supply), but, moreover can
provide us with a positive stabilised
d.c. voltage! By choosing the right
ratio between R5 and R7, T1 will
cutoff the moment the supply voltage
drops below 7.2 V. As a result the
collector voltage of T2 increases,
causing N1 to activate the correct
notation on the display.

A 9 V battery such as a PP3 is quite
sufficient, since the circuit consumes
only a few milli-amps. A mains supply
is also possible, and it is for this reason
that R1 and the zener D1 are added to
the circuit.

The temperature sensor

There are various types of sensors on
the market, and the only reason we have
picked two particular ones, is that they
are inexpensive.

Original tests showed the KTY 10 from
Siemens to be ideal, but, as this can be
difficult to get hold of, we also tried

the TSP102 manufactured by Texas
Instruments which worked well. Most
of the types looked at consisted of a
silicon plate, whose resistance depended
on the temperature. The only real
difference between types was their
temperature range. The KTY10, for
instance ranged from -50°C to +1 50 C,
whereas the TSP was effectiv| over a
range from — 55°C to 125 C. The
first version has a nominal resistance
of 2000 ft at 25°C and the TSP 1 000 ft
again at 25°C. The temperature co-
efficient was 0.75%/°C and 0.7%/°C
respectively. These last figures denoting
the resistance increase, per degree
celcius, as a percentage over the nominal
value.

The accuracy of the circuit is mainly
dependant on the width of the
measuring range. Which type to use is
left to the discretion of the constructor.
A serial resistor (R10) is applied (in
series with the sensor) in order to
stabilise the linearity of the sensor,
especially when small measuring ranges

are required. Table 2 provides a sum-
mary of several ranges, with the
linearity error, and serial resistor values
needed. Table 3 describes, in detail, the
differing sensors, together with their
housing dimensions and type numbers.

Construction

Figure 2 illustrates the specially de-
signed printed circuit board of the
circuit.

The dimensions of the board and the
way that the components have been
grouped together allow the com-
pleted circuit to fit into a case manu-
factured by Vero (type Nr. 65-2996H).
Provision has been made for all the
components to be mounted onto the
printed circuit board. Constructors
should make sure that low profile
sockets are used for I C 1 , IC2 and the
display. The display can be inserted
into a 40 pin socket which has been
sawn in half. We also advise the use
of good quality multi-turn presets.
As with anything made of glass, great

care should be taken when handling
the display, especially when inserting
it into its socket. Too much pressure
on the glass plates may cause the
display to appear internally smudged,
12v permanently!

When using the circuit as a normal
thermometer, the decimal point DPI
should be connected to point Y by
means of a wire link. Obviously,
depending on the application, the
decimal point can be moved around,
by using a rotary or slide switch, if
required.

As already stated earlier, the circuit
!2V is designed to take either the 7106 or
the 7116. For the 7106 wire links
are required across points A and B and
06, as shown on the component overlay
illustration of the printed circuit board.
In the case of the 7116, link 06 is
be removed and replaced with a link on
points '16' Should you then require
the ability to freeze (hold) the display
reading, link AB has to be replaced with
__ a simple on/off press button switch.

Keep in mind that this facility is not
available when using the 7106.

The sensor can be connected to the
circuit by means of ordinary insulated
wire, the length of which is not critical.
In fact anything up to 30 metres is
possible without difficulty. For re-
liability we suggest encapsulating the
soldered connections of the sensor
with epoxy resin or glue.

A PP3 type 9 V battery is ideal for
the power supply, as it has the advan-
tage of fitting nicely into the battery
compartment of the Vero case.
Constructors wishing to feed the circuit
from the mains, can install a miniature
supply socket next to the battery, to
cater for a 9 V mains adapter. Figure 3
clearly illustrates how this should be
wired. The battery supply will be
automatically cut off immediately a
power plug is inserted.

A single bolt or screw with a spacer
ensures the circuit is firmly fixed into
the case. A piece of clear perspex in
the window of the case will protect the

+ 40°C
+100°C
+140°C
+13(fC
+150°C

+0.08

+0,03

+0.07

+0.6

. — 0,04°C
. — 0.02°C
. — 0.04 o C
. — 0.6°C
,-1®C

jr TS . . . 1 02 sensors

.+ 45°C
,+100°C
. + 1 25°C

display. The switches, sockets and so
forth can be mounted in the power part
of the housing.

The current consumption of the circuit
when using the most commonly avail-
able sensor (TSP102) is only 2 mA.
Several sensors, which are activated
consecutively by a separate switch can
also be used. To do this correctly,
sensors have to be selected for equality,
otherwise errors in measurement
readings will occur.

Calibration

Perhaps we have been a little too quick
to explain how to install the circuit
into the case, because first of all it has
to be calibrated.

Initially the sensor has to be placed into
a small cup of chopped melting ice. The
cup should contain more ice than water,
and the water must cover the ice com-
pletely. Give the sensor time to react
(about 5 minutes), and turn PI until
the display reads 00.0. P2 sets the scale
factor. How this is adjusted depends on
the measuring range required. For lower
temperatures (— 25°C to +45°C), P2 can
best be calibrated using a normal
thermometer. Insert both thermometers
into a bowl of water having a tempera-
ture of around 36 ... 38 C, give the
sensor a little time to react, and then
set P2 so that the reading on the display
corresponds.

Higher measuring ranges can be cali-
brated by suspending the sensor
boiling water, and then adjusting P2
until the readout is 100°C. The only
critical aspects of this procedure are to
ensure that the water really is boiling
and that the sensor does not touch the
sides, or bottom of the kettle.

Finally as you have completed the
circuit, why waste the hot water. Make
a nice cup of tea and relax. H

10-24 - elektor

Jber 1982

Measuring distances is not difficult,
especially when the right equipment
is available. Modern technology has
certainly done away with the old
comical approach of measuring the
shelf size by spreading your arms out.
For short distances a simple ruler will
do, but, in surveying, everything from,
chains, theodolites, and sonic equip-
ment is used.

The main advantage of using ultrasonic
sound is, that the need for any mechan-
ical parts is completely eliminated,
simplifying construction considerably.

In practice we found the circuit to be
accurate upto about 10 metres, which
is very good considering the circuit is

ultrasonic

distance

measurement

a good starting point for experimentation

There are several ways to measure distances. The method adopted
depends not only on what is being measured, but also on your
occupation.

The circuit described utilises ultrasonic sound, working on the principle
that as, sound travels through the air at a known speed, the time taken
and therefore the distance travelled between two points, can be easily
determined.

only a starting point for further
experimentation.

Before going on to describe the circuit
in any great detail, it is interesting to
know a little about the definition of
a 'metre', and in the various ways the
standard has been arrived at over the
last 300 years.

Exactly one metre
The 'metre' started life around 1792.
At sometime during that year it was
decided to define the standard as: one
millionth part of one quarter of the
earth’s peripheral circumference. Fine
in theory, but, totally inaccurate in
practice. Scientists soon found out that
the circumference of the globe was
constantly changing.

A new standard was determined in
1799, and seemingly quickly forgotten,
as far as the history books are con-
cerned. The next one to be recorded

anu iu appeared in 100 a, a mere

90 years later. This was made of a
mixture of platinum and iridium. It is
this reference standard which can be
seen in Sevres, near Paris. Rumour, at
the time, gave the idea (unjustifiably)
that the standard was based upon the
hight of Napoleon II and as his stature
was diminishing with every military
defeat, it could not be accepted. For
whatever reason (perhaps best left
unsaid) the rest of the world continued
to search for a more accurate standard.
As most of you are aware one particular
country in europe (remaining nameless),
took nearly 300 years to even come to
terms with the fact that a metre even
existed.

At the beginning of the twentieth
century, scientists started to look at
the possibility of using the wavelength
of light to define the metre. Conse-
quently the cadmium lamp became the
international standard for spectroscopy
in 1927. For the uninitiated, this means
the study, measurement, and analysing,
of rays, light and other phenomena by
optical means. The actual unit of
length was defined as the Angstrom
(1 A = 10 10 m). Even this was not
good enough for certain applications,
although it is still used as a secondary
definition.

The modern standard was established
in 1960 using the wavelength of a
krypton lamp, which as a matter of
interest was not supplied by the famous
strip cartoon gentleman.

The metre is defined as equaling
1 650763,73 wavelengths of radiation
(measured in a vacuum) released by
Krypton isotope 86KV during its
transition between 2p 10 and 5d s .
The multiplication factor remained,
because scientists still wished to relate
the new standard to the old original

A new way of defining the standard is
now emerging, using the helium-neon-
laser, and so it will not be long before
a more complex result will be ,
accepted.

It is a fact of life that the more ad-
vanced our technology, the more
accurate any standard measure has to
be.

Measuring distances
To summarise, the ways that distances
are normally measured can be placed
into three main catagories.

• mechanically

• optically

• electronically

The mechanical method does not need
explaining, as the tools are rather
obvious.

Optically, distances are measured in a
trigonometric way (triangles).

Last but not least we have the electronic
system of measurement. Nearly all
methods use some form of radiation
like radio waves, light, sonic or infra-
red rays. As the propagation speeds
of all of these are known, it is a matter

jber 1982 -

of determining the time taken for a
waveform to travel between two points.
Infra-red radiation is mainly used for
great distances (a number of miles), as
it is relatively simply to modulate.
Electronic equipment has been used to
measure distances of 60 miles and
above, but the effectiveness of such
systems depends on a number of factors,
such as atmospheric conditions, visi-
bility and so on.

With the advent of space technology,
lasers are used in combined electro-
optical systems to determine the hight
at which satellites are orbiting the
earth.

In practice

All the methods so far described are
mainly used to measure great distances.
The average man in the street, certainly
does not require sophisticated equip-
ment to decide the size of his living-
room carpet, unless of course he lives
in a mansion house or castle.

There is one popular hobby in which
the accurate measurement of distances
is very important; photography! As we
all know, it is essential to determine the
exact distance between the subject
being photographed, and the camera,
otherwise the lens cannot be adjusted
correctly for focus. The industry
supplies quite a lot of aids and equip-
ment which overcomes this problem.
Most cameras utilise some form of
optical trigonometric system, with two
or more indicators within the view
finder which have to be aligned by
rotating the range adjustment of the
camera. Reflex cameras for instance,
use a complex network consisting of
a fronted glass, a wedge frame and
tiny prisms.

During the last few years, a number
of manufacturers have introduced the
automatic focusing camera. Quite a
few employ a system of mirrors and
LL prisms, with an electric servo motor

moving the lens accordingly. Some are
even equipped with an infra-red LED
and lens enabling the camera to be
automatically focused at night.

A recent development, worth a closer
look is the new sophisticated system
introduced by Polaroid.

The Polaroid system

When considering automatic focusing
cameras, the Polaroid system is some-
thing really special, in as much as it is
the only one using ultrasonic sound.
A large honey-comb patterned gold
coloured disc on the outside of the
camera casing acts as the transducer
(transmitter/receiver) for the sonic
pulses.

Figure 1 illustrates the actual distance
meter contained within the camera.

The transducer sends out a 'burst' of
1 ms duration. This consists of a series
of pulses each having a different fre-
quency, four to be precise (60, 57,
53 and 50 kHz). The reason for so
many frequencies is that it is possible
for a particular frequency to be ab-
sorbed, rather than reflected, by the
subject being photographed. The chance
of this happening depends on the shape
and material of the topic. So, rather
than putting all the eggs in one basket,
by transmitting four frequencies, the
reflection of at least one is ensured.

The transducer then switches over to
reception immediately the burst has
been transmitted. The reflected signal .
it receives is then amplified and fed to a
digital circuit which determines the
time interval between transmission and
reception. The circuit processes the
signal and in turn controls a servo
mechanism, which adjusts the lens to
the correct focus setting. The gain
of the receiving amplifier can be varied
(in 16 steps), dependant on the distance
the signal (burst), has had to travel.
Obviously the greater the distance
between camera and subject, the weaker
the signal.

The system is proven and therefore
functions well and accurately. It is
effective up to a maximum of 1 0 metres,
which is more than sufficient for
normal photographic purposes.

The ultrasonic distance meter
The Elektor design team have combined
their thoughts, together with the ideas
contained in the Polaroid innovation to
come up with an ultrasonic distance
meter.

As mentioned earlier sound, ultrasonic
or otherwise has a known speed through
air. Therefore the time taken to travel
from transmitter to subject and back
again can be used to determine the
distance. The transmitted 'burst'
supplies a start pulse to a counter
which operates at the same frequency
as the propagation speed of sound
in cms per second. The received re-
flected signal provides the stop pulse.
The counter would therefore give the
distance over which the Burst has
travelled. This value would obviously
be twice the distance between the
object and the transmitter, so, a simple
division by two would give us the
correct answer.

Figure 2 illustrates in block diagram
form what we have just described;
transmitter, receiver, counter with
readout, and an oscillator switched
on and off by the transmitted and
received pulses.

The circuit diagram
The circuit diagram of the complete
circuit is shown in figure 3. The trans-
mitter consists of gates N1 and N2,
which together form a bridge circuit.
The ultrasonic transducer US1 is con-

nected between the two gate outputs
to ensure an a.c. voltage 1 8 Vpn exists
across it (with a supply of 9 V). N1
also acts as an oscillator, which is
switched on and off via N3. Its fre-
quency, set by PI , depends on the type
of transducer used. This particular
design has a 40 kHz type from TOKO,
but, there are others that will do the
job. The oscillator frequency is adjusted
by PI to as near 40 kHz as possible
as this is the point at which the trans-
ducer reaches its maximum efficiency.
The receiver has been kept simple
because of the experimental character
of the circuit. Two consecutive emitter
circuits (T5 and T6), amplify the signal
received by US2. T7 operates as a
threshold detector, as it only conducts
when its base voltage is lower than
the supply (-6V). To put it another
way; T7 conducts when the a.c. voltage
(measured at the wiper of P2) exceeds
1.2 Vpp. A further oscillator is con-
structed using IC3 and its surrounding
components (R17, R18, P3 and C9).
IC3 is in fact a 2 14 divider with built
in oscillator.

The frequency is set to 17190 Hz
using P3, since the speed of sound is
343.8 m/s at a temperature of 20°C;

The 214 digit DVM published in our
February 1981 issue is used as a counter
with readout. I Cl (counter, latch and
display control) directly drives the
displays Dp2 . . . Dp4, which are
multiplexed by I Cl via transistors
T2 . . . T4. IC2 feeds a stabilised 5 V
supply to the counter and display
stage of the circuit. IC1 is able to drive
4 displays, but Dpi and T1 (from
original DVM circuit), are omitted, as
only three are required here.

Nearly all the other components in
the circuit are needed to synchronise
the various stages. The importance of
correct timing is illustrated in figure 4.
This denotes the differing pulses and
frequencies present at various points
in the circuit.

With an oscillator frequency of
17190 Hz, output Q14 of IC3 will
have a signal frequency of approxi-
mately 1 Hz, (17190/2 14 ). This output
is connected to the latch input of IC1
via a monoflop (N6, R19, CIO) and also
to the reset input via inverter N7 and a
second monoflop (N8, R20, Cl 1 ). With
the arrival of a negative going edge at
Q14, a short pulse is fed to the latch
input. A positive going edge at Q14
supplies the reset input with a pulse
instead. The signal from Q14 is inverted
by N7 and fed to 2 further monoflops;
one (N3, RIO, C5) driving the trans-
mitter and another (N4, R 1 1 , C6)
connected to the reset input of flip-flop
FF1. The clock input of FF1 is connec-
ted to T7 and its 0 output to N5.
Therefore I Cl gets a reset pulse with
every arrival of a positive going edge
at the Q1 4 output of IC3, automatically
zeroing the counter. At the same time

the monoflop around N3 is activated
(with a negative going edge at the
output of N7), thus releasing a signal
from the transmitter/oscillator for
0.3 ms. During this time period US1
transmits about 12 (40 Hz) pulses,
which are then reflected by the subject,
and received by US2. Simultaneously,
the moment the ultrasonic signal is
transmitted, FF1 is reset and held by
monoflop N4 (almost 2 ms). Conse-
quently output 0 becomes logic T
and the signal from the 17190 Hz
oscillator is fed to the counter (IC1)
via N5. Once the received and amplified
sonic signal (burst) reaches the clock
input of FF1, output 0 becomes
logic '0' with N5 blocking the counter
input of I Cl . The counter now contains
the actual distance measured in cms.
N6 activates the latch moving the
contents of the counter into the latch,
which is then displayed. The counter is
reset by the next positive going edge
at Q14 allowing a new measurement
to be taken. The previous readout
remains on display until the information
from a new measured distance arrives.
New readings can be taken every
second.

There are some further aspects of the
circuit which need a further expla-
nation. US2 will of course pick up the
transmitted signal immediately, unless
we do something about it. If we do not
avoid this the counter will be cut-off
straight away defeating the whole
exercise. We get over this problem by
ensuring that the mono-time of N4 is
considerably longer than the time it
takes to transmit the 'burst' (2 ms).
During this time frame the flipflop
remains in the reset position, not
caring whether a signal is present at
the clock input or not. After the 2 ms
FF1 is released so that the circuit does
not confuse a reflected signal with a
direct one. The only drawback of this
inbuilt delay is that distances less than
35 cms cannot be measured, so you
will have to rely on a ruler.

The circuit does not include an AGC
in the receiver stage or an automatic
error compensator (comparing a number
of consecutive readings for the same
distance), in order to keep it as simple
as possible.

Constructional points to consider
The counter and display stages can be
mounted onto one of our ready made
printed circuit boards, as listed in the
EPS lists as number 81105-1, (first
issued in February 1981). Remember
that Dpi, T1, C2a and C2b illustrated
on the component overlay of this board
are omitted. One side of R8 is fed to the
decimal point (Dp2) the other to
ground. Pin 6 of I Cl must also be
connected to ground.

We suggest using veroboard for the rest
of the circuit. Keep wiring as short as
possible and the transmitting/receiving
stages separated from each other. The
two transducers are mounted side by

4

elektor October 1982 — 10-27

-«'»*■* '<=' n #

i-— r

I“~" j.

y. direct signal

BB| r-

Figure 4. This figure denote! the different pulses and frequ.
in the circuit.

mcies present at various [

to be placed further apart, and a higher

side, (not touching) both facing exactly
the same direction. We advise the use
of flat 4.5 V batteries as a mains supply
may cause stability problems. Power
consumption is rather high at 250 mA,
but this cannot be avoided when using
a LED display. The use of an LCD was
discounted as being too expensive for
just an experimental circuit. Even so,
the batteries should still have a long
life simply because the circuit would
normally be used a few seconds at a

The correct operation of several com-
ponents and stages can be checked
without the need for a scope. Interrupt
the connection between N5 and the
clock input and connect the latter to
pin 4 of IC3 (output Q8). The display
should then read 128. When the clock
input is shorted to ground the display
should read 000. This is a good way
to check both the display and oscillator
stage around IC3. The transmitting
function can be checked quite easily,
just by listening to US1 . Although the
actual 40 Hz signal cannot be heard,
the 'burst' will be heard as a soft click,
(one every second). Should you happen
to hear Radio 3, then something is
certainly wrong. In that case write to us
and tell us how you did it! Testing the
receiver is not easy, but you can assume
all is well when the d.c. voltage at the
collectors of T5 and T6 is approxi-
mately 4.5 V.

Once all this is done then the complete
circuit can be tested and calibrated.

Turn the wiper of P2 to maximum, and
take a note of the reading. This is
produced by the counter between the
reset and latch pulses, which are always
half a second apart. It is important to
remember that this reading will always
be displayed when the receiver does not
pick up a reflected signal. Now aim the
circuit at an object like a closet, which
is one metre away and has a vertical area
of at least one metre square. Slowly
rotate P2 backwards, until a certain
point is reached when the display reads
approximately 1 metre. Should this not
occur and the read out is in the range of
40 to 60, then the transducers will have

value capacitor used for C6. Once a
setting of P2 is arrived at, which corre-
sponds to a reading of 1 metre, we can
go on to the next stage, which is to set
the 40 Hz frequency.

Keeping the circuit in the same position,
rotate P2, clockwise until the display is
blank. Now rotate PI, until a reading is
once again displayed. This procedure is
repeated until it is no longer possible to
blank out the display by any further
adjustment of P2. Reposition the
circuit a measured distance of let us
say 5 metres from the subject and reset
P2 only until the correct reading is
displayed.

Finally position the circuit an exact
distance (3 metres) always from the
same subject, and adjust P3 until the
display indicates the correct reading,
and that is it!

We obtained very good results with
the prototype. The accuracy was
± 2 cm at a maximum distance of
7 ... 8 metres. The accuracy is de-
pendant on the ambient temperature,
air pressure and humidity, as these
factors, influence the speed of sound.
The range of the instrument can be
extended by increasing the gain of
the receiver and by increasing the
transmitting voltage. Equiping the
meter with an offset adjustment (to
correspond to the length of the meter
housing), will allow wall to wall
measurements.

This particular design can also be
used in a car, so that the driver is
always aware of the distance between
himself and a wall or other vehicle.
Very useful when trying to parkl
In fact this idea has already been put
into practice by certain manufacturers.
Any reader wishing to do this can
modify the circuit quite easily by
substituting the display for an acous-
tic indication. A series of rapid bleeps,
which increase in number as the
distance decreases, until a continuous
tone is heard, advising an immediate
halt, (unless you want to pay the
repair bill that is).

10-28 - elektor October 1982

When deciding which capacitors to use,
consideration should be given to re-
liability, the permissible range of oper-
ating conditions, size and so on. Size is
important especially when building high
density circuits, and last but not least
price. Keep in mind, that, any need for
special current limiting resistors is going
to increase the overall cost of using
tantalums. Even so, tantalum capacitors
are used widely, where the operating
characteristics of the capacitor is critical.
Quite a few Elektor circuits specify the
use of tantalums, and not just because
they are small and good to look at.
They have a stable capacitive value, and
a long shelf life. The impedance is vir-
tually unaffected by frequency changes.
So, on the face of it tantalums are ideal.
However, they do have one major draw-
back; price!

applications, and certainly fell down,
for high frequency designs.

From the offset for many applications
the tantalums did not have too much
competition, and with the craze for
miniaturisation they filled a need
straight away. The biggest factor
initially in favour of tantalums was this
question of shelf life. Even after years
of storage, their current leakage, and
value remain unchanged. In fact their
shelf stability is about 100 times better
than the wet aluminium type.

Apart from everything so far explained
the tantalums were not embarrassed by
temperature. The wet electrolytics are
affected quite considerably by tempera-
ture, causing large increases in the
leakage current level. It is this failure to
dissipate heat that causes their mediocre
conductance.

elecirolytics run dry

everything you wished
to know!

So far for many industrial and
professional applications wet
electrolytics and tantalums have
ruled the roost. With new
innovations and technological
advancements, it is now possible
to use alternatives which can be
cheaper and more reliable. Many
factors need consideration when
making the right choice, and a
good knowledge of the merits and
limitations of the different types
available is useful.

In fact a comparison shows that
the new solid aluminium
electrolytics, can be used as
alternatives for tantalums, and in
many respects can be seen to be
better.

With such an incentive as this, some
type of alternative was needed.

With the constant improvement in tech-
nology, coupled with energy saving and
the conservation of natural resources,
many experts started to question the
sanity of using tantalum for capacitors.
Tantalum is now in limited supply, as
with everything else, and its price is
increasing by leaps and bounds. So, why
use tantalums?

We must first of all keep in mind that at
the time that they emerged onto the
market, the only other type, with which
they could be compared, were the wet
aluminium electrolytics. These were and
still are inexpensive, but are generally
larger than any solid counterpart and
suffered from a relatively short shelf
life. By this we mean, that after long
storage, their leakage current increases,
and can only be restored to its original
value by post-forming. Also, if they are
used close to their maximum operating
temperature, their life expectancy,
which is normally far less than solid
types, is curtailed even more. The wet
type could not be used in certain new

By comparison the tantalums have a
wide operating range as far as tempera-
ture is concerned, making them suitable
for filters and oscillators. Hence the
reason why they are widely used in
Elektor designs.

Most of you by now must probably
think that the writer must be com-
pletely sold on tantalums. Not so! They
do have, what can be termed as incon-
veniences, rather than faults or disad-
vantages:

• The voltage level they can sustain
when connected the wrong way

round is extremely small, even for a
very short interval. They breakdown
rapidly and can explode easily.

• Their a.c. voltage performance is
poor and further diminished at high

frequencies and temperatures.

• The charge/discharge rate resistance
is 3 S2/V, making it necessary to use

series resistors.

• A surcharge, whether it is of a
thermo, current, or voltage nature,

will cause immediate breakdown, short
circuiting and a possible explosion.

• The price of each item is quickly
approaching prohibitive levels.

All in all tantalums are certainly not
perfect, mind you what is these days!
Should series resistors not be used in
order to limit the charge/discharge rate,
then the results are always fatal. This
is because field crystallisation will
occur, causing short circuiting.

At the beginning of the article we ex-
plained all the advantages of using
tantalums; impedance, heat dissipation,
life span, high frequency performance
and so on. But, it seems that it did not
take us very long to arrive at the con-
clusion that even these are not as good
as we would wish. As with many things,
it is a fact of life that the more you use
something the less appealing it becomes.
Notice that we do not say everything.

jber 1982 - 10-29

Figure 3. Structure of a solid aluminium electrolytic.

since the good things in life are always
welcome.

Luckily the capacitor manufacturers
have not stood still and have come up
with a relatively new development.

Using deeply-etched foil an axial-lead
solid aluminium electrolytic has been
created, achieving a high CV density
making them less expensive replace-
ments for tantalums. Although they are
not going to depose the latter com-
pletely, they will be very widely used in
a variety of industrial and professional
equipment.

Solid aluminium capacitors

The solid aluminium electrolytic has a
comparable performance with the tan-
talum type, but not only is it cheaper,
but it does have a few advantages.

Figure 1 shows the different com-

Figure 5. Stability of the main electrical
parameters of the three types of electrolytic

as a function of time. The curves are for a

0.1 pF item measure at 85°C.

ponents which go to make a tantalum.
There are a lot of similarities in con-
struction with the solid aluminium type
(SAL). Looking at figure 1, you will
note that the former has layers of silver,
graphite and manganese dioxide (MNOj)
which form the cathode. Then comes a
dielectric layer and finally the anode
made of tantalum. This is sintered to
the tantalum oxide (dielectric layer).
Figure 3 shows the make up of a SAL.
The cathode is composed of the same
materials as the tantalum. The real
difference between the two lies in the
fact that, the anode is composed of
deeply etched aluminium and that the
dielectric layer is aluminium oxide
AL2O3. Hence the remarkable conduc-
tivity of the solid aluminium electro-
lytic!

These SALs, to coin a phrase, are very
robust to say the least. They can operate
near their maximum temperature ratings

without shortening their life span, and
do not have any catastrophic failure
mechanism. In other words they are not
going to blow-up in your face at the
wrong moment. An added bonus is the
fact that series resistors are not needed.
The values already available are in the
range 47...1000pF and one major
manufacturer has proposed to make
smaller ones with 0.22 ... 47 pF, but,
it may take some time before these are
available.

They are slightly larger than their
equivalent counterparts, and although
being less expensive than tantalums they
are marginally more costly than the wet
type.

Present applications include telecom-
munications, space programs, and power
stations. Their small size and robustness
make ideal for the automobile industry.
Because they are being improved upon
all the time, a rosy future lies ahead of

them.

To summarise the main characteristics

of the SAL are:

• Lower price.

• Their voltage rating remains un-
changed throughout the operating

range, even at high temperatures (-80

to 1 75°C) .

• The allowed d.c. voltage (reversed) is
around 33% of their rated voltage.

• Does not require current limiting.

• a.c. voltages (up to limits) can be
handled and do not adversely affect

their performance.

• Their impedance fall more steeply
with increasing frequency than any

other type.

• They can withstand 50/100 Hz a.c.
voltages up to a level which is 80%

of their d.c. rating.

• Temperature stable, and low failure
rate coupled with longevity.

darkroom

computer

photometer,
temperature meter
and process timer

darkroom computer

Readers who have already read the
'instructions for use' in the previous
issue, will have a pretty good idea of the
capabilities of the darkroom computer.
However, before all the available
features can be used the necessary
accessories have to be constructed. In
this case there are three such circuits; to
measure light, temperature and a time/
sequence indicator.

The processor timer is little more than
a box containing a series of LEDs in
a line giving an optical indication of
the passing of time. After all, it is nice
to know how much longer a particular
photograph has to remain in the devel-
opment tank of bath, rather than panic
at the last moment when the acoustic
alarm is heard. The rows of LEDs, in
efect, indicate the amount of time that
has elapsed since the start button was
pressed. How long each LED remains
lit and at which point the buzzer
sounds is determined by the computer.
The alarm can sound after 15 or 25
LEDs have consecutively lighted, de-
pending on the time frame decided

The light meter is used to establish the
correct exposure of the paper and
check the contrast of a negative. A
special colour corrected diode converts
the quantity of light it detects into a
pulse width modulated (PWM) signal,
which is then fed to the computer.

The electronic thermometer keeps a
continuous check on the temperature
of tanks and baths, and is accurate to
± 0.1 C. This kind of accuracy is
achieved by using a sensor with ex-
cellent linearity properties.

Each individual circuit has its own
specially designed printed circuit board,
with provision for mounting all com-
ponents. The circuits are all separately
housed and are connected to the com-
puter with multi-way ribbon cable.

The process timer

The circuit of the process timer, as
illustrated in figure 1, consists of
25 LEDs and two four-to-sixteen
decoders. The address inputs of
IC1 and IC2 are connected to lines
PB0 . . . PB4, whereas, the LEDs are
connected to outputs 1 ... 15 of
IC1 and 0 ... 9 of IC2. The operation
is straightforward. The binary code
(which LED lights when) is fed to lines
PB0 . . . PB4. PB4 is logic 'O', enabling
IC1 , that is to say as long as the binary
number is below 16. When the binary
number is between 16 and 25, then
PB4 will be logic 'T and IC2 will be
enabled.

The LED display is multiplexed in the
interests of the power supply. During
2.5 ms the first code for the first LED
is put onto lines PB0 . . . PB4, and
during the next 2.5 ms the code for the
second and so on. If only one LED is
'running', during the second time
period a '0' will be on lines PB0 . . .PB4,
and as output o of IC1 is disconnected
nothing will happen.

jber 1982 - 10-31

The set-reset flip-flop, constructed
around N1 and N2, is on the left-hand 1
side of the circuit diagram. This flip-flop
really acts as a noise suppressor for
switch SI, and is linked to the NMI
connection of the processor. D epre ssing
SI, the flip-flop supplies the NMI line
with a negative going edge, which starts
the process timer program and lights
the first LED. Pressing SI a second time
starts a second LED 'running' at a time
interval from the first, enabling two
processes to be timed. If SI is then
depressed a third time, nothing will
happen. Programming the timer is
described in the constructions for use
published in the previous article
(part 1). The oscillator and piezo buzzer
is activated when a logic 'V appears on
line PB6. Potentiometer PI adjusts the
output volume of the buzzer.

One aspect of the total process timer
is important. After having read both
articles it becomes apparant that there
are in fact two timers. One, let us call
it the second one is included in the
computer, whereas, the first ( a separate
one) forms part of the remote circuit.
This first timer works completely
independantly of the second and
therefore the computer, which allows
all the other facilities to be utilised
without interruption; ligth and tem-
perature measurements, the running of
the second timer and so on.

The light meter

As can be seen by looking at the circuit
diagram in figure 2, the light meter is
the smallest accessory. However, it
would be a mistake to assume that it
is also the simplest. After all the dif-
ferences in light levels to be measured in
any darkroom are going to be very small
indeed! It is therefore essential to have
a good quality sensor such as the
BPW21. This particular photo diode
has almost the same light and colour
sensitivity as the human eye, which
makes it also suitable for colour defi-
nition measurements. Further more its
conversion of light to current is
extremely good, as far as linearity is
concerned (range of 10' 2 . . . 10 s lux).

A photo diode reacts very quickly to
changing light conditions, in contrast
to light dependant resistors (LDRs)
which need a considerable amount of
time to adjust especially in low levels
of light.

The current flowing through the photo
diode is proportional to the intensity
of light. So that the processor can
handle the information, the current has
to be converted into some form of
digital signal. The simplest solution was
to use a current to PWM converter. A
closer look at figure 2 may give the idea
that the photo diode is short circuited,
after all it is connected between the
ground and the inverting input of IC1.
This is correct, because this forms a
virtual earth connection due to the
feedback loop via IC1. This technique
is used to ensure that the operation of

Figure 1 . The circuit diagram for the process timer. The passage of time is displayed by
of 25 LEDs.

2

10-32 -ele

1982

the diode is independant (as much
as possible) of ambient temperature
changes. The current supplied by the
photo diode to IC1 ensures an increase
in the output voltage of the 1C. The
speed at which the voltage increases is
directly proportional to the rise in
current, and therefore to the intensity
of light. The output of IC1 is fed to a
7555 timer (a CMOS 555 timer). The
input of the timer remains logic '1'
as long as the input voltage of IC2 does
not exceed 3.33 V. At this level or
higher, the output is pulled low ('0'),
only returning to logic '1', when the
input voltage becomes 1.66 V or lower.
Supposing an output voltage from IC1
slightly below 1 .66 V, then the output
of IC2 will be logic 'V. In this case FET
T2 will conduct and FET T1, which is
connected as a diode will cut off.
However, the voltage level at the output
of the integrator (IC1 and Cl) increases,
due to the current supplied by the
photo diode. As soon as the output level
reaches 3.33 V, the output of IC2 will
change (to logic '0'), T2 will switch off
causing a current to be fed to Cl via R2,
T1 and R1 . This current not only flows
in the opposite direction, but, is also
much stronger that that supplied by the
photo diode. As a result the output
voltage level of IC1 will reduce, very

quickly. Consequently when the output
of IC1 reaches 1.66 V, the output of
IC2 will return to a logic '1', causing T2
to conduct and T1 to cut off again. As
the current from 01 is integrated, the
output voltage increases again starting
the complete sequence all over from the
beginning. The events just described are
repeated continuously. The point of all
this, is that it establishes the fact that
the output of IC2 only remains as
logic '1' during the integration of the
photo diode current. Therefore the
length of time during which a logic 'V
situation exists is proportional to the
intensity of light detected by the
photo diode. The greater the intensity
the shorter the time!

The only thing left for the processor
to do is to decide upon the width of
the pulses in order to calculate the
right exposure time. The processor
determines the average value by sam-
pling the pulses supplied during 2
seconds. In this way any measurement
cannot be influenced by any 100 Hz
modulated light source, such as the
lamp of the enlarger. Remember that
this lamp is controlled by a 50 Hz a.c
voltage supply.

Some readers may still be puzzled as
to why a FET (T1) is used as a diode.
The voltage drop between the drain and

darkroom computer

source of T2 is still a few millivolts
when it is conducting. Therefore T1
prevents any current flow through R1,
no matter how small. Should current
flow via R1 any measurement will be
cancelled out, since the current through
the diode is still about 100 pA, with
low levels of light. At low voltages, a
FET connected in this way has a much
lower voltage leakage than any normal
diode. For example at 200 mV, a
BF256A used in this way has a current
leakage of 20 pA, whereas a 1N4148
would have about 12 nA! Quite a
a difference!

With such low current levels a very low
leakage integration capacitor is also
necessary. A type which has a time
constant of more than 100 s should be
used for Cl . In other words the internal
leakage resistance multiplied by the
capacitance must exceed 100 s. The
input current of IC1 is typically 2 pA,
with a supply voltage of 5 V, which
means it does not need to be taken
into consideration (negligible).

The thermometer

The electronic thermometer also works
under the same principle of pulse
widths. The sensor in this circuit is an
LM 335, which is in effect a temperature
sensitive zener diode. The voltage

Figure 4. The printed circuit board for the process timer has been designed to fit the BOC 430
case from West Hyde.

supplied by this diode in mV is equiv-
alent to ten times the temperature in
degrees kelvin, so that at 0 C (273 K)
the diode voltage is 2.73 V. The sensor
is accurate over the measuring range
15 to 50°C, which is ideal for use in
darkrooms.

Figure 3 illustrates the circuit diagram
of the temperature meter or ther-
mometer. The voltage drop across the
sensor is applied to the positive input
of A2. Under normal conditions the
gain of A2 is set to approximately
8 X. Opamp A1 supplies a reference
voltage, determined by PI, to the
inverted input of A2. Preset PI is
adjusted so that the output of A2 is
0 V at a temperature of 10°C, and
800 mV at 20 C. This output is then
fed to the inverted input of a comparator
A4. The other input of A4 is connected
to a linearising capacitor C3. This
capacitor is charged with a constant
current source, fed via A3, T1, R4, R5,
R9 and P2. When the voltage level
across C3 exceeds that from the output
of A2, the comparator A4 causes T3
to conduct. In effect, the time it takes
for C3 to be fully charged is what we
are interested in as this is proportional
to the temperature.

Once T3 conducts, C3 discharges via
T2, which you will notice is connected
in parallel to C3. The base of T2 is in
turn connected to the X terminal of
the darkroom computer. The processor
supplies, via this line, 10 pulses per
second, during the measuring period.
This pulse train causes T2 to conduct
and C3 to discharge ten times a second.

With a logic '0' on line X, T2 is not
conducting and C3 charges. The time
taken between a logic '0' on line X, and
when '0' appears at the output of T3
on line PB0, is used to determine the
temperature. It is in effect this infor-
mation that is digested and processed
by the computer, which then shows it
as a temperature reading on the display.

Construction

Process-Timer

Figure 4 shows the printed circuit board
and component overlay for the process
timer. This has been designed so that it
can be mounted into a BOC 430 case
from West Hyde. A slot is needed in the
top of the case to accommodate the
LEDs (25). Do not forget to make the
necessary holes for the press-switches.
For a graduated scale and indicator we
suggest two strips of card fitted on
either side of the LEDs. On the one side
a series of numbers (for the LEDs), and
process periods on the other. The photo-
graph in figure 5, gives good idea as to
how this proposal will look. For the
prototype we used, the first block to
denote the development, the second for
the stop bath and the last for the
fixer. In order to keep to this time
frame, the alarm must be programmed
to sound when the 10th, 13th, and 19th
LED is on.

Temperature meter

Figure 6 illustrates the printed circuit
board for the temperature meter, or.

elektor October 1982 - 10-33

Parts list for the process timer

Resistors:

R1 ,R2 = 5k6
R3= 10 k
R4 = 220 O
PI = 1 k preset

Capacitors:

Cl - 180 n
C2- 10 m/10 V

Semiconductors:

D1 . . . D25 * LED
IC1.IC2-74LS154
IC3 = 74LS132

Miscellaneous:

SI - digitast switch

PB = piezo buzzer Toko PB 2720 (Ambit)

to give it its correct title, the digital
thermometer. The sensor is remote from
the circuit. Care should be taken to en-
sure that the LM 335 is wired correctly.
Only the centre (+) and the negative
pins are used. The ADJ connection is
not required and can be cut off. The
terminals and soldered joints of the
sensor should be encapsulated in epoxy
resin. This ensures against short circuits
and increases the reliability factor, (see
photograph in figure 7).

The printed circuit can be mounted into
its own case, in the main computer
housing, or even within the process
timer! The choice is left to the con-
structor, but, we suggest the latter. In
this case the amount of external wiring
is greatly reduced, because points
PB0 and 0 can be combined with con-
nections +10 V, X, PB0, NMI, +5 V and
0 of the process timer into one length of
multi-way ribbon cable fed to the com-
puter. Try and keep the cable as short as
possible, although, anything up to 2
metres will not cause problems.

The light meter

This is again a completely separate
section and is the last accessory we will
deal with in this article. The printed
circuit board as illustrated in figure 8,
is assembled in a rather unconventional
manner. The components are in fact
mounted on the track side of the board
while the other side completely covered
in copper. This acts as the earth plane
and the front face of the housing.
Screening from outside interference is
of paramount importance for the circuit

10-34 - elektor October 1982

darkrc

compute

5

u 4

Figure 5. This photograph of the process
timer illustrates one example of the timing

the case have the copper surface facing
outwards while on the ends and sides
it is facing inwards. Construction may
be aided by the use of a vice to hold the
parts together while soldering.

If your finished case is an excellent
example of the art of air-tight boxes it
is your own fault . . . the connecting
wires to the microprocessor should
have been fed through a hole in one
end before completing the box 1 1

Calibration

The only circuit requiring calibration is
that of the temperature meter. The
link next to C3 on the printed circuit
board of the temperature meter must
first be removed. A power supply of
between 3 and 10 volts (a 4.5 V battery
if necessary) is connected to pin 1 1 of
IC3 and earth (negative to pin 11). At
this stage the temperature meter is con-
nected to the computer but the latter is
switched off. Firther requirements for
the calibration procedure are a volt-
meter, a thermometer and a developing
tray of water.

The voltmeter is connected across
resistor R8 and the sensor is placed in

the tray of water. This is where the
thermometer comes in. The water must
be at a temperature of precisely 10°C.
This will not be particularly easy but
persistance will pay off in the long run.
A thermometer that is very clearly
readable will make the job a lot easier.
Unfortunately, the majority of domestic
thermometers do not fall into this
category. The sensor must be suspended
in the water without touching the sides
or the bottom of the tray. After giving
the sensor a little time to settle down,
adjust PI until the reading across R1
is 0 V. Two or three attempts may be
needed before results are considered
satisfactory.

The tempory power supply can now be
removed and the wire link replaced.
Switch on the computer and enter
MEAS,— 2 on the keyboard. With the
sensor suspended in free air, the com-
puter reading should display between
10 and 50° C.

The sensor is now placed in water at a
temperature of about 35-40°C. Allow
time for the sensor to adjust and then
set PI so that the computer reading
corresponds to that of the thermometer.

to operate correctly and, for this reason,
the entire housing is constructed of
copperclad board. The sides, ends and
top and bottom of the case are simply
soldered together. As the top (or front
face) of the housing is the printed cir-
cuit board, this must be assembled
before the rest of the case is fitted
together. Care must be taken when
mounting the components on the board
as the risk of short circuits is high with
this type of construction. However, the
efforts are well worth while as the end
result is very effectively screened and,
incidentally, can also look very neat if
care is taken.

Remember that, although the sensor is
mounted inside the case on the same
side of the board as the rest of the
components, it must protude slightly
through the board (sensor opening)
enough to prevent it from being
shielded. It can be fixed in place by the
use of epoxy resin (or similar) but, be
warned, an overzealous application of
the 'sticky-stuff' can affect the sensi-
tivity enough to prevent the circuit
from operating at all. This was learned
by experience on one of the proto-
types.

It will be noticed that there is just one
small hole through the printed circuit
board. This may appear as though some-
body started to drill all the holes
through and then thought better of itl
Not true of course. An off-cut of wire
is fed through the hole and soldered
both sides in order to earth the earth
plane! Don't overlook this step or
it will effectively nullify the whole
exercise.

If the construction seems a little com-
plex the illustration in figure 9 will
clarify matters. The top and bottom of

Resistors:

R1 = 4k7

R2 = 6k8 1% metal film
R3 = 5k6 1% metal film
R4 = 3k9 1% metal film
R5 = 82 k 1% metal film
R6 = 10 k 1% metal film
R7 - 68 k 1% metal film
R8 = 3k3

R9 = 4k7 1% metal film
R10 = 68 n
R11 = 10k
R12 = 8k2

PI = 1 k multiturn preset
P2 = 2 k multiturn preset

Capacitors:
Cl = 220 n
C2= 100 n
C3 - 270 n

Semiconductors:

T1 = BC557B
T2 = BC 547B
T3 = TUN

IC1 = LM335Z (National)
IC2 ■ 723
IC3 - 324

6

Figure 6. All the components for the temperature meter, with the exception of the sensor, ere
mounted on this printed circuit board.

alektor October 1982 - 10-35

6

9

Cl = 56 p ceramic
C2 - 560 p
C3 ■ 10 p/10 V

Semiconductors:

T1 = BF 256A
T2 = BS170
D1 = BPW21
IC1 =3130
IC2= 7555

Figure 9. The construction of the light meter case is illustrated here. The printed circuit board
forms the top face.

8

Figure 8. Care should be taken with the
assembly of the light meter printed circuit
board. As explained in the text, all the
components are mounted on the same side
as the track pattern.

In practise . . .

Before dashing off to the darkroom and
locking yourself in, it must be decided
what times the different processes are
going to take so that a process timer
card can be drawn up. As an example
we are going to use time factors associ-
ated with black and white photography.
These can be 1.5 minutes for develop-
ment, half a minute for the stop bath
and one minute for the fixing bath. This
is in fact the timing shown in the photo-
graph in figure 5. Keeping in mind that
the time from one LED to the next is
10 seconds, then the 'alarm' LEDs will
be numbers 10, 13 and 19.

Pressing the START PR. T key will now
begin the process time. However, it may
be that a second timing procedure is
required if, for example, a black and
white film is to be developed. The
timing for this can be 6 minutes for
development, 1 minute for wash, 3
minutes for the fixing bath and then a
30 minute wash. In order to carry out
this sequence the memory in the main
computer will have to be programmed
accordingly. Remember that up to 10
different time periods are available. The
programming procedure for this has
been covered in detail in the user
instructions in part 1 . Keep in mind the
fact that it is also possible to program
acoustic signals during the processing
sequence. These will be a short bleep
for each LED with a longer buzz for the
end of each period.

A point in part 1 that may need further
clarification is the multiplication factor
relating to the light meter. Moving the
sensor to different areas of the enlarged
image will obviously change the light
level falling on the sensor. If a number
of positions are checked the computer
can provide an average reading.

Another method that may be used
requires a sheet of opaque tracing paper
placed between the enlarger lens and the
sensor. The image will of course be out
of focus but a good idea of the average
light level can be found in this manner.
An 'average' negative must be used in
order to arrive at a correction factor.

The paper in use is also an important
factor when determining the exposure
time. Therefore, by experience and the
information of the materials used, an
error adjustment (or multiplication
factor) can be arrived at and entered
into the computer which will then
compensate accordingly. This factor
need only be entered once as long as
the same materials are used. This is
really no different to any normal
manual procedure only that, in this
case, once the factor is found and
entered into the computer it can be
forgotten (after making a note of it!).
The method of entering the multipli-
cation factor into the computer was
described in detail in part 1.

1 0-36 — elektor

>ber 1982

Since we published the SSB receiver
article in June of this year, it is appar-
ent from all the letters received, that
quite a few enthusiasts in general
electronics have developed a taste for
short wave listening. As the popularity
grows, then the need to cover a larger
number of bands increases. The SSB
receiver ideally fits the bill, in as much
as it is capable of covering the whole of
the amateur bands, obviously together
with the necessary converters. Basically
each circuit acts as a wave band shifter,
converting the aerial signal which is
either above or below the 20 metre
band, into the band which the SSB
receiver can receive without modifi-
cations. Each converter is connected
to the aerial input of the receiver
eliminating any need for changes to
the actual receiver. This in effect means
that the circuits described can be used
with virtually any short wave receiver.

is then mixed together with the fre-
quency from a fixed crystal oscillator to
give a number of frequencies at the
output (of the mixing stage). Since we
are only interested in signals lower than
14 MHz and because the first filter only
very roughly separated out the particu-
lar band in question, a second filtering
stage is necessary. This now extracts the
product of frequencies (in the right
band) which are required. The basic
reason why we use a crystal for this
purpose is because they are easily
available and relatively cheap. For the
very low frequency bands (VLF), such
as 10 . . . 140 kHz, which are also
possible using this technique, things
are slightly different. In this case we
would use a crystal which gives an
oscillator frequency slightly below the
14 MHz band, and then the result is
that the summed up frequency comes
into the desired band. In this case the

short wav<“ band shifting
for SSB receivers

from 14 MHz to 14 metres!

The article deals with and
describes front ends which can be
used with any short wave receiver,
specifically the SSB described in
our June issue, effectively
extending the coverage of the
amateur bands. One circuit is
designed to convert the band
below the 14 MHz 'up' into the
required receiving range with the
second converting down from
higher frequencies again into the
14 MHz band. It is also possible to
cover the 2 metre band using this
technique. The circuits, as their
name implies (front ends) can
simply be connected to the input
of the SSB receiver. The number
of circuits required will only
depend on the number of bands
constructors wish to cover.
Component values are given
enabling up to 13 converters to be
built, plus of course the original
20 metre band already in the SSB!
This is a good way to increase
your band coverage in nice easy
stages.

Lower than 14 MHz

One of the simplest solutions for these
wave lenths, is to use a band-pass filter,
followed by a mixing stage which is in
turn followed by another band-pass
filter. The first filter is used to extract
only the required wave band. This signal

first stage becomes a simple low-pass
filter.

Figure 1 shows how a double 'deck'
wafer switch is used to select the
required wave-band, assuming all the
different converters (one for each band)
have been built.

Figure 1. Building five 'up' conveners, allowes the reception of five extra amateur wave bands
(above 14 MHz).

Figure 2 shows the complete circuit
diagram for a lower than 14 MHz
converter as just described, which is
in other words an 'up' converter. The
part of the circuit in the bottom left
hand corner which includes Cl . . . C6,
LI and L2 is the band-pass filter which
caters for 1.8, 3.5, 7 and 10 MHz.
Directly above this section is a stage
consisting of C7...C10, L3 . . . L5,
which is the low-pass filter for the VLF.
The component values needed for
differing bands are shown in table 1.
This is followed by the simple passive
mixing stage constructed around the
FET BF256C. This FET acts as a
switch, which is controlled by the
crystal oscillator built around T2.
The sum and difference of the product
of the filtered input frequency and the
oscillator frequency, appears at the
output of the mixer. The frequency
from the crystal for the 1.8, 3.5, 7 and
10 MHz is chosen so that the difference
of the product from the mixer falls
into the 14 MHz band (which can be
dealt with by the SSB receiver). For the
very low frequency band 10... 1 40 KHz,
it is the sum of the frequencies which
comes into the correct band. The value
required for the crystal is for each case
listed in table 1. The output of the
mixer is then fed to a band-pass filter
which ensures that only the 14 MHz
band is brought to the input of the SSB
receiver.

The input impedance of the converter
with band-pass filter is 50 £2 and that of
The VLF-one, 1 to 2 kS2. The latter
higher value ensures that a simple aerial
(single piece of wire) will still provide
adequate reception for that band. The
loss in signal strength as a result of using
a converter is 6 dBs. with a loss in the
filter stage of 2 dB, which is very small.

Higher than 14 MHz

The converters for anything higher than
14 MHz can be used right up to the
2 metre band! Figure 2 once again
shows the circuit in block diagram form.
The input stage is the same as before; a
band-pass filter. But, then instead of
going straight to a mixing stage an
amplifier is included. From then on the
sequence is identical, that is, as far as
the block diagram is concerned. When
the circuit itself is described in greater
detail you will see that this is not the
case. Again a crystal oscillator is used,
but, in this case a further buffer stage
has been included. The circuit diagram
is shown in figure 4. The component
values for the band-pass stage depend
on the wave band required, and are
shown in table 2. After the filter there
is the amplifier T1 which is followed by
a further filter set up in the same man-
ner as the first (L3, C8. C9). A mixer
follows this, with T2 being controlled
by a crystal oscillator. Finally there is a
buffer stage with T4. For all the wave
bands listed in Table 2 the values have
been calculated on the basis of the

Figure 3. Block diagram for the down converter (above 14 MHz). Although this seems simpler
than the circuit shown in figure 1, it does illustrate the principles used.

Again a separate converter is required for each band and a multi-way switch can be used to

Table 1. The component values for converters for lower than 14 MHz.

Band L1.L2 Cl C2.C4 C3 X

MHz pH nF pF pF kHz

0.01... 0.14 - 14000

1.8 (160 m) 27 3.3 180 33 16200

3.5 (80 ml 8.2 3.3 180 15 18000

7 (40 m) 2.2 2.2 180 10 21300

10 110m) 1 1.5 150 6.8 24300

difference between the oscillator and In the 2 metre version the gain of the
the aerial input frequencies. After the converter is between 6 and 12 dB, and
buffer stage comes the last band-pass in the others approximately 4 dBs. In
filter, ensuring only signals within the the latter versions the gain can be
14 MHz band are fed to the receiver. boosted by increasing the value of R3,
At the foot of table 2 there is a separate but, then the value of L3 has to be
box giving the component values needed lowered and C8 also increased,
for the 2 metre band. In this case a
65 MHz crystal is used, and the buffer Construction

stage also operates as a frequency Figures 5 and 6 show the printed circuit
doubler. boards for all converters. The board in

figure 5 is for bands below 14 MHz
(circuit in figure 2) and in figure 6 the
board for wave bands above 14 MHz is
shown.

The only item worthy of note when
building the circuit onto the printed
circuit board shown in figure 5, is that
a screen must be positioned across the
board exactly where the dotted lines
are shown.

Both boards are double sided, meaning
that the component side is a copper
plane which must be earthed.

The only really difficult circuit to build
is the 2 metre version in as much as,
the coils (LI, L2, L3) have to be wound
by hand. L3 is the simplest coil to make,
as It has only one turn, but L2 and L3
both have 4 windings each, so please
take care. These coils must also be
inductively coupled, which means they
have to be mounted side by side and
end to end, and not at right angles,
(as shown on the layout). There is
plenty of room allowing all this as some
of the components, namely C2 and C4,
are not needed for the 2 metre version.
Figure 7 gives a clear illustration of this.
A screen is also required on this board,
to separate the input stage from the rest.
The dotted lines as shown in figure 6
indicate where this is positioned. We
suggest that all, the completed circuits
are mounted into screened boxes.
Although this involves a considerable
amount of work the final results fully
justify it.

Parts list for below 1‘

Resistors:
R1 - 100k
R2 = 39 k
R3* 1k2

Capacitors:

C1,C2.C3,C4 = see table 1

C5.C6 = 60 p trimmer

C7 = 6n8

C8.C10 “ 1 n

C9 = 2n2

Cl 1 = 270 p

Cl 2 = 27 p

C13- 120 n
Cl 4= 1 n ceramic
C15 = 20 p trimmer
C16- 56 p

Coils:

L1,L2 = see table 1
L3 = 33 mH
L4.L5 = 4,7 mH
L6 = 100pH
L7 = 6.8 pH

Semiconductors:
T1 = BF 256C
T2 = BF 494

X = crystal ( see ta

Alignment

High frequency designs ( R F) need
great care when aligning. However, the

alignment of the circuits so far de-
scribed is not critical and is certainly
straightforward. It is just a matter of
adjusting all the trimmers until the
maximum signal-to-noise ratio is
achieved. Obviously as with any normal
aligning procedure, every time one
trimmer is set, it will influence the
other, therefore each should be adjusted
in turn (several times) until the correct
alignment is achieved.

One final comment concerns the
trimmer C13 in circuits catering for
frequencies above 14 MHz (figure 4).
This is used to ensure that the oscillator
frequency is exactly what is required,
getting rid of any discrepances in the
crystal frequency. 14

10-40 - elektor October 1982

16 channels

ily five ICs

Figure 1 illustrates the circuit diagram
of the transmitter. It contains; a key-
board with 16 on and off keys, the
transmitting 1C, and final stage. A 9 V
battery such as the PP3 is ideal for the
supply.

A command given by depressing a key
is immediately converted into a corre-
sponding E-D-C-B-A 5-bit binary code.
We have purposely left the detailed
explanation of how the code is allocated
till later, as any reference to it as this
stage could confuse the issue.

The 5 bit code, which is nothing more
than a pulse sequence of 6 identical
signals, is transmitted by modulating
the infra-red diodes D1 and D2. The
actual information, is hidden in the

operational switch, ensuring that the
power consumption does not exceed
6 pA, when the circuit is in a stand-by
situation.

Any even number of keys can be used
as long as they do not exceed 32. This
is because an 'off' as well as an 'on' key
is required for each function.

The receiver shown in figure 2, consists
of; a preamp (IC1), and the pulse pause
modulation (PPM) decoder, IC2 . . . IC4.
The input transistor of IC1, and the
receiver diode D 1 , are biased in the
same way, by receiving their base
current setting from T1 . The input stage
of IC1 is followed by three differential
amplifiers, with the output (pin 2)
feeding out the received PPM signal.

16 channels witfi only
five ICs

There have been many circuits
published for infra-red remote
control systems of varying com-
plexity. The design here however,
while still providing 16 channels,
keeps construction fairly simple
by utilising special ICs produced
for the purpose by Plessey. In
fact, the circuit is similar to that
found in many domestic television
sets.

Control is by means of push
buttons, and both the transmitter
and receiver can be made very

gaps, or breaks between pulses. A short
pause denotes logic 'V, a long one
a logic 'O'. Pulse and pause duration can
be set with the aid of trimming poten-
tiometer PI. Keep in mind that a logic
'0' pause is about 1.5 times longer than
a logic '1', and that each pulse is ap-
proximately 3 ms. There is preset delay
period of 54 ms between any two
different commands. Infra-red radiation
is only possible when pin 3 of IC1 is
pulled low. This is the only way that a
current 'pulse' or surge (lasting about
15 /is), will flow through T2 and the
diodes. As a matter of interest this can
be as high as 8A!

The 1C contains an internal electronic

PPM decoder

ICs ML 928 and 929 can be found in
various kinds of circuits, although they
were originally designed for use in
TV sets. They each contain; a PPM
demodulator, time base generator
(together with an oscillator), and a
shifting register with following latches/
memories. The binary information is in
fact, located at the output of the
latches. Apart from all this, there is
an integrated comparator taking care
of the error correction, automatically!
So, under normal conditions nothing can
go wrong, what so ever! Each 1C is able
to digest and process 16 commands
converting the received information into

compact. -y

jf; a keyboard, transmitter 1C. final stage and a 9 V battery.

binary code. The way Pin 2 is con-
nected , together with the setting of PI,
determines the oscillator frequency.
ML928 reacts to codes from 00000 to
01111, with ML929 reacting to codes
starting at 10000 and ending at 11111.
This is ideal for our purposes.

In order to control a total of 16 func-
tions both ICs are needed. Only one
decoder is required when only 8 func-
tions are sufficient. In this case, care
must be taken to ensure that the correct
codes are allocated which correspond
to the keys being used. An allocation
can easily be derived from the keyboard
matrix shown in figure 1.

The 5-bit code is transmitted in the
sequence E-D-C-B-A and interpreted by
the receiver accordingly. Column E
denotes which of the two ICs it is meant
for ('O', ML928, 'V ML929); D supplies
the 'on' or 'off' information; C, B and A

contain the information for which of
the 8 functions is to be connected.

The decoding circuit constructed around
IC5 converts the code into switching
pulses for T2 . . . T9. IC2 and IC3,
produce the 'write disable' (WD) pulse
for IC5. The EXOR gates register any
level changes at the inputs; DATA, A2,
A1 and AO. The NOR gate, situates a
logic '0' at the WD input. The allocation
of the binary codes to the outputs of
IC5 and to the switching outputs, is
infact, mirrored! This is simply because,
IC4 is supplied with a negative operating
voltage, enabling it to also function with
negative logic. That means, output Q6
and not Q1 is used when the data '001'
is situated at AO . . . A2. Consequently,
the switching signal at the DATA input
will reach T8 via output Q6. The WD
input will be logic 'V and the data input
will be blocked when no further switch-

ing information is on the inputs of
IC5. The outputs are not affected.

Construction, calibration,
application

We advise constructors to build the
circuits with printed circuit boards, the
design of which are shown in figure 3
and 4. Unfortunately, due to unforseen
circumstances we are unable to supply
ready made ones from our EPS service.
However, we feel certain that this will
not prove to be a stumbling block for
our readers.

The transmitter diodes are equipped
with reflectors, which improve the light
beam intensity, making it possible to
control devices from a range of 8 to 10
metres.

Miscellaneous:

Tr = 16 V/0.1 A mains transformer
fuse 63 mA

two-pole mains switch Figure 6. Design and component layout for the power supply board.

A keyboard can be built up on vero- inside the apparatus being controlled,
board quite easily, and then inserted, as this means the use of low capacity
together with the battery and circuit connection wire, rather than thick
into a plastic case. obtrusive mains cable. Mechanical relays

The receiver needs a 15 V supply. This will require a protection diode con-
can either be obtained from the equip- nected the reverse way round. Any relay
ment being controlled, such as the can be used which has a maximum coil
stereo system and so on, or from the voltage of 12 V and a resistance of
power supply illustrated in figure 5. around 150 fi.

Where you install the receiver will The calibration procedure is quite short,
depend on what is to be controlled. Set PI of the transmitter to its mid
Obviously, anyone wishing to control position and depress a key (to switch
more than one piece of equipment something on). Now adjust PI of the
should house the receiver in a separate receiver until the relay is activated,
plastic case, and install separate or one Repeat the procedure a few times
multi-way socket. until you are satisfied that the relay is

The output signals from the unit are activated every time the key is de-
suitable for use with relays. As a matter pressed and that is it! M

of interest our June 1982 issue de-
scribed a solid state relay which would
be ideal for this application. We suggest
the relay is placed near to, or actually

« SSB « SSB » SSB « SSB « SSB » SSB

Readers may be excused for jumping to
the conclusion that the preamp is only
useful for increasing the sensitivity.
However this is only partially true.
After all the SSB receiver already has
an exceptional signal-to-noise ratio
(0.15 /zV at 10 dB ) . Nevertheless the
RF preamp does supply an extra
lOdBs, which, you must agree is useful
and in some cases desirable. This im-
provement in sensitivity is certainly
going to be welcomed by SSB owners
with a small or compact aerial.

Apart from the better selectivity and
sensitivity, this RF stage supplies
additional gain in order to overcome
some of the traditional SSB problems.

pre-amp for die

SSB receiver

boost the selectivity and sensitivity
of your SSB.

In the normal course of events, it
is a fact of life that anything good
can always be improved upon. It
is therefore only natural that we
should try to improve a good and
successful design such as the SSB
receiver published in the June
1982 issue.

An additional MOSFET preamp
not only increases the sensitivity,
and betters the selectivity but also
extends the AGC range.
Improvement just for the sake of
change is not a philosophy
followed by us. We therefore feel
certain that SSB users, especially
ones without access to large
sophisticated aerial systems, are
going to appreciate the little extra
that the circuit introduced here
gives.

Practise has shown that 'strong' trans-
mitters operating within the 19 metre
band can, under certain conditions,
'swamp', or interfere with, other
weaker stations. Despite the extensive
filtering included in our SSB receiver,
this will still occur, if only from time to
time. Trying to overcome this problem
by simply twiddleing the controls is
time consuming, aggravating and to
some extent futile. When you consider
the fact that some of these offending
stations have a transmitting power of
around 2 mega watts, fighting them off

is harder than standing on the beach
and trying to stop the tide. And every-
one remembers what happened to the
last man or king who tried that!

As David did with the proverbial
Goliath, we have supplied a subtle and
highly effective weapon in the form of
an additional band-pass filter at the
input of the RF stage. The band-width
of this filter is 5000 kHz and together
with the filters originally included in
the SSB receiver, it provides adequate
protection against the 'giants'. In
effect the circuit becomes highly selec-
tive, thereby muting and damping the
overbearing 19 metre transmitters, even
when using large, sensitive aerials.

A further advantage of using the RF
preamp relates to the ability of the
receiver to handle and control high
level input signals, (eliminate cross-
modulation). Basically we are adapting
the principle that the more amplifi-
cation stages which are controlled by
the AGC voltage are used the more
effective the AGC will be.

As the AGC voltage controls the gain
of the MOSFET, the result is to con-
siderably extend the effective range of
the AGC. In practice this increase is
about 20dBs! Strong signals adjacent
to weak ones are now 'squeezed' a little
more allowing the receiver to handle
them much more easily. Consequently
the receiver produces less noise when
being tuned and has good station
separation.

To summarise, the additional RF
preamp enable the user to achieve;

• higher sensitivity.

• better selectivity.

• an extended AGC range.

The circuit diagram

Looking at figure 1 will show that a

wind coil, L3, can be replaced by a resistor R3.

1982 - 1045

re 2. L3 is constructed as follows;

nd the result 10 times onto the core end
Ider the two uncommon wires together t

dual gate MOSFET specifically type
BF 900 is used as the active element of
the preamp. Elektor readers who have
already built or are thinking of building
the SSB published in our June issue,
will probably wonder why we are using
the same type of MOSFET as we did
before (for the RF, oscillator and mixer
stage of the SSB). After all there are
several other semiconductors which can
be used to construct an excellent RF
preamp for 14 MHz applications! Well
the answer is quite simple. The BF 900
is easily available, inexpensive as far as
MOSFET devices go, and experience has
shown us that it is ideal for RF appli-
cations.

Returning to the circuit diagram,
readers will note that a dual band-pass
filter network is positioned at the
input. This is made up of LI, L2, and
Cl . . . C5. The dual gate MOSFET
follows the filter enabling a 'classical'
amplifier design to be constructed.

Gate 1 of T 1 is connected to the voltage
source via R1. The source voltage level
is set by R2 and D1 to +0.6 V. The gain
of T1 is varied by connecting the AGC
voltage to gate 2. This is a positive
voltage, the level of which depends on
the strength of the input signal. The
stronger the input signal, the lower the
gain factor. Therefore with a higher
voltage across gate 1 than gate 2 a con-
siderable reduction in gain is achieved.
With weak signals, maximum gain is
effected (about lOdBs), thus increasing
the sensitivity from 0.15 /tV to 0.05 pV
with a signal-to-noise ratio of lOdBs.
The amplified signal is taken from the
drain of T1 via a double wound (bifilar)

» SSB « SSB « SSB « SSB « SSB » SSB » SSB « SSB » SSB « SSB « SSB * S

MOSFET are as short as possible. This

H) - £/D will ensure the good operation of T1.

' Coils L1 and L2 are quite straight
forward to wind. They both consist of

it.. Ip 18 turns of 0,6 n ? m diameter c °PP er

m wr if) core f° rmers - having an outer diameter

of 0.5 ins. The prototype used toroids
JzSm manufactured by Amidon Associates,
but in case constructors have difficulty
XJpb finding them, a complete specification

Wst is included in f'9 ure 3 t0 enable equival-

” ents to be found. Apart from the

£|j physical dimensions please keep in mind

^ (ft that the completed coil should conform

y -T, A Q to the same 'Q' factor as the prototype,

0 otherwise the performance will be

‘ » When making the coils, you should

ensure that the windings are evenly
spaced to cover all of the core. In
contrast to L2, LI is tapped two wind-
ing up from the ground connection.

The construction of L3 is not straight-
forward and that is why we have
devoted a separate section for it at the
nd any two uncommon wires. end of this article. Anyone not wishing

realise the centre tap. The other two ends are to get j nv0 | vec | with L3 can replace it

with a drain resistor R3, but as already

explained some of the performance is

o then lost.

° We suggest inserting the completed

Typ*c*i curve* f-om mu, -.nd.^ on m« »n,« co,«. circuit into the case of the SSB receiver,

220 1 1 | [ | | 1 I 1 ! as any available space is suitable. Try

210 Q j ‘ 1 ^ T50-6 and put it as close to the aerial con-

I 1 1 I /M I t ~ t J-'* " ; ' ' ~~ nection of the RF section (of the SSB)

2oo • ; ~ -■ _• • - as possible. The output of the preamp is

' / I tty.. /]■■'' l 19-1 ^ connected to the aerial input on the

190 1 ' / n [ i/- 4 ■ SSB printed circuit board by means of

iso | t jLi-J -l— j j | {■ 22 25 2*64 coaxial cable. The link between the

nU, I I ^ ^ aerial bus and the preamp input should

170 Vm ,' ( -L “ “ also be made with coax,

iso -j-}- Frequency (Me.) -j- 31 92 3140 The AGC connection point on the SSB

1 3 1 5 1 7 ' » ’ Vi ' 13 ' is ‘ it ‘ i‘» receiver board is clearly indicated and

82164 3 should not present problems. The

supply voltage can be derived from the
... . . . junction of L11 and L12 on the RF

Figure 3. Toroidal core specifications of the ’ .

original prototype, which used Amidon cores, seemun.

together with the q chan. Double windings (bifilary) of L3

type T50-6 permeability factor 8. Thj$ coj , cons j sts 0 f -JO double windings

type T50-2 P™«»>iiity i°- with cen tre tap, on a toroidal core (core

hL hainht m«an innohr specification see figure 3).

.3031™ .18* 3.20ol” ' First ol all two equal lengths of enam '

elled copper wire are twisted together
as shown in figure 2a. The result should
coil L3. Constructors not fond of | 00 k like the old fashion two way flex,
winding coils can replace L3 by a drain f his double wire is now wound onto the
resistor. However this will effectively, r j n g core (io windings), ensuring that
if only, slightly reduce the sensitivity. t h e windings are evenly spaced over the
The dotted line section below the main whole core. Figure 2b clearly illustrates
circuit diagram illustrates how the drain t hj s procedure.

resistor is connected. The next stage is to trim any excessive

length of wire. Now, with the aid of an
Construction ohmmeter, or continuity tester, find

Taking the simplicity of the circuit any two uncommon wires (see figure
into consideration, constructing the 2b), an solder them together as shown in
preamplifier onto veroboard is rela- figure 2c. This in effect is the centre
tively easy. The actual method adopted tap. The other two remaining wires are
is not critical, but, you should take connections a and b of the coil as
care that the connections to the denoted in figure 1. H

coil L3. Constructors not fond of
winding coils can replace L3 by a drain
resistor. However this will effectively,
if only, slightly reduce the sensitivity.
The dotted line section below the main
circuit diagram illustrates how the drain
resistor is connected.

Construction

Taking the simplicity of the circuit
into consideration, constructing the
preamplifier onto veroboard is rela-
tively easy. The actual method adopted
is not critical, but, you should take
care that the connections to the

10-46 - elektor October 1982

An active antenna certainly cannot
perform miracles. If, for example, we
really do want to receive the "Voice of
the Andes' on 17790 kHz, we really
need a resonant X/2 dipole aerial of
about 8 m in length. An active aerial
with a rod length of about 1.5 m can
only be substituted for the dipole as a
physical compromise. The following will
show how we arrive at this compromise.

adiw aerial

A short, active aerial for DXers

There it is, the new communications receiver. But how do we get it
to pick up the "Voice of the Andes"? As the saying goes: a good
aerial is still the best RF amplifier. However, the landlord will probably
object to metres of wire and suspending the wire around the inside
of the home will provoke the wrath of the family. A tangle of wire
under the sofa is no solution either. What we need is an active aerial:
it should be short and convenient to use whilst enabling good reception.
An active antenna certainly cannot perform miracles. If, for example,
we really do want to receive the "Voice of the Andes" on 17790 kHz,
we really need a resonant 2/2 dipole aerial of about 8 m in length. An
active aerial with a rod length of about 1 .5 m can only be substituted
fot the dipole as a physical compromise. The following will show how
we arrive at this compromise.

Some radio principles
The problem involves an 'electrically
short’ receiving aerial for frequencies
below 30 MHz, the hunting grounds of
the short-wave DXer. But here is the
paradox: how can an active aerial with
a rod length of about 1 .5 m be used over
the frequency range of 1.5 — 30 MHz,
in which we normally have to utilize

half-wave dipoles of up to 95 m in
length?

Here we have to go into a little more
detail. Atmospheric noise is the factor
governing the design of receiving aerials.
In the case of our half-wave dipole, the
atmospheric and industrial noise level is
high compared to the noise level of
commercially available receivers. Thus,
reception quality depends only on the
signal itself and the interference
received.

If the aerial is shortened, the signal-to-
noise ratio is initially constant because,
although the signal level is reduced, so
is the level of received noise. However,
there is a limit to this shortening in
length - the point at which the 'elec-
tronic noise' of the receiver, which is
independent of the aerial, becomes
greater than atmospheric noise. Figure 1
shows the relationship between signal-
to-noise ratio and aerial length in a
graph. In region b, an aerial which is
considerably shorter than a 'normal'
one, can still be utilized. In this case
the received noise level is just as high as
that of electronic noise.

Short aerials of this type are con-
structed as vertical aerials (rods or
whips) and horizontal aerials (dipoles)
for reception over the range 10 kHz to
30 MHz.

Matching

So far so good. But why can't we simply
connect an aerial to the receiver in the
form of a short rod? This can be best
explained by figure 1. First of all, the
signal level received by the aerial is not
greatly reduced when the aerial is
shortened. For example, a dipole which
is short with respect to wavelength
receives only 10% less signal level than
the half-wave dipole. The problem is in
the question of matching.

In figure 2, the aerial is represented as
an AC voltage source with the charac-
teristics Ra (= radiation resistance) and
Xa (= reactance) . At a constant fre-
quency, the radiation resistance is
proportional to the square of the
length of the dipole. The reactance is
inversely proportional to the length.
This means that the shorter the aerial,
the greater is the reactance. With a short
dipole of 10 m overall length, for
example, we have the following values
at 1.5 MHz: Ra approximately 0.5
and Xa a few kilohms. With proper
matching for the transfer of power,
however, this total impedance must be
exactly the same as the input impedance
of the receiver, (50 S2). Considering the
aerial itself as a voltage source, and as
the impedance increases as it is
shortened, then the consequences are
severe. Feeding a high impedance
unloaded voltage to a low impedance
input of a receiver simply means that
you will get absolutely nothing out!
What is required is a proper match!
With passive aerials, transformers are
employed to correct the miss-match.

Using this technique on active aerials
would also work, but, only over a
narrow frequency range.

The solution to our problem is really
quite simple! The short high impedance
aerial is first of all connected to an
amplifier which also has a high im-
pedance input. Thus the unloaded
voltage from the source (aerial), is not
destroyed. Matching the receiver to the
amplifier is achieved by providing the
amplifier with a low impedance output.
To summarize, therefore: the secret of
the active aerial is that when the short
aerial (short with respect to wavelength)
is properly matched to a receiver (using
an amplifier stage), it delivers exactly
the same reception results as its big
brother. An added benefit is that it
provides advantages in DX reception.
The theoretical explanation would be
beyond the scope of this article, but it is
true to say that from a technical view-
point, active aerials are a good compro-
mise between high sensitivity and short
dimensions.

The active aerial

The active aerial consists of three parts:
impedance transformer and amplifier,
power supply, attenuator (see figure 3).
The RF part of the active aerial is
designed around transistors T1-T3. The
passive part, the aerial rod, is applied
directly to the gate of field effect
transistor T1 via coupling capacitor Cl.
T1 is configured as a source follower,
resulting in the desired performance as
an impedance transformer (high input
impedance, low output impedance).
T2/T3 form a two-stage RF amplifier
whose amplification is adjusted by R7
and R9. The amplification can be
increased is necessary by varying R7 and
R9. In this case, the values in parenth-
eses apply.

The circuit is powered by the remote
power supply consisting of Trl, the
bridge rectifier and C5 (see figure 4) .
The DC voltage is applied to the output
of the amplifier via L1/L2/C6. The
DC voltage reaches the amplification
stages via L3.

The attenuation stages which are selec-
ted with S2 and S3 form the third part

of the active aerial. Thus the output
signal from the amplifier can be atten-
uated by -6 dB, -12 dB or -18 dB or
not at all, depending on switch settings.
This avoids overdriving the receiver
input.

We have chosen a wide-band version for
the active aerial so that it can be erected
as far as possible from sources of inter-
ference. We shall examine this aspect
later. For this reason, band selection
using switched capacitors and/or a
variable capacitor is not provided.

The quality of the aerial is in no way
inferior to commercially available types.
The so-called intercept point IP3, a
measure of intermodulation perform-
ance of the circuit, is at 30 dBm. By
way of comparison, a commercially
available aerial (AD-270/370) exhibits
the same value. The frequency range
extends from 3 kHz to 100 MHz
(—3 dB) with T2/T3 providing a gain of
11 dB!

Practice

Figures 3 and 4 show the printed circuit
boards for the two sections. T3 should
be fitted with a star-shaped heatsink.
Once we have soldered in the com-
ponents, we must decide where to
position the aerial. In any case, the
aerial rod must be directly connected to
the appropriate terminals on board 1.
The optimum location for the aerial is
at a distance of at least 1.5 m beyond
the building's interference field. In this
case we need an aerial rod of about
30 cm in length which is accommodated
in a waterproof housing together with
the amplifier. Aerial and output socket
joints must be properly sealed. The
output stage of the amplifier is designed
to be able to 'drive' up to 100 m of
coaxial cable. At the receiver end,
board 2 with the power supply and
attenuator is directly located at the
aerial input.

The second-best application for the
active aerial is indoors. In this case, both
printed circuit boards and the aerial rod
(now 1 m in length) can be installed in a
housing.

Now all that is left is to try it out. Have
fun and good DXing. K

3 4 5

TO-92Z TO-126 TO-3

(SOT-32)

F I 9

elll. JIL

- 10-53

elektor October 1982

Linear ICs

Voltage Regulators

£>]

301

318

709

> 741

CA3130
| CA3140

LF 355/356/357

TL 071/081

5o? 7805

^3"“ 7806

18 7808

7815

oil]* 7824

1°A

CTl „ 7905

7906

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=ji ’ R LM 387

VV § NE542

[ol 78M05

m-- d ss

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MOmA

[OL* 79M05

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-500 mA

®S So TL 074

§- r r ,-- 7 *^ TL084

jf^l

Si [f© RC4136

|A|

HI 78L05

PS 78L06

78L08

78L12

78L15

#111 0 78L18

78L24

100 mA

«■ 79L05

1 S 79L06

79L08

79L12

79L15

- 11# 79L18

A 79L24

-100 mA

SH

LMioc

u out* 5v LM309K

*,U, 'ouflA

LM323K

•0 l out = 3 A

C A 3080

LM 13600

U ou t- 1.2 V... 37 V

LM317K

LM 723

l 0 ut " 200 mA

(pesdeU*) 37Vmax

L . . ... . A ]

U ou t‘ 2.85 V... 40 V

'i o Lx L 200

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Input Q

Output $

All ICs shown top view

1

IliilBilMiIlIll

I information appearing on the CRT, can Portable oscilloscope

»di^ e ,fon qU Both Wi ‘at°^ N ° W available ,rom Ma,ron is ,ha BS - 310!

picture (normal resolution) o
picture (high resolution).
Thandar Electronics Limited,
London Road,

Huntingdon Cambs.
Telephone: 0480.64646

Home telephone system (Elektor 89)

Extensive tests have shown that improved
performance can be realised with a few minor
modifications regarding the power supply
of the home telephone system. In short, it
means lowering the power supply output to
5 V. This really only applies to those systems
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The modifications to the circuit consist
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The changes are as follows:

(2483 M)

Inexpensive LCD multi-testers

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Video printer

The TP55 VIDEO PRINTER, now available
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Selective-call module

Frequency counter

T ^ The model 86 1 0B : 600 I
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(2443 M)

1982 - 10-57

True RMS 4% digit multimeter

The newly introduced 1504 multimeter from
Thurlby Electronics combines true RMS ac
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The instrument is designed for bench or field

designed for bench or field peg mounting inductive
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rower consump Memory mapped VDU card

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A display of up to 8.000 characters or 256 x
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