D 71883 87,88 July | August 1982 130 p. up-to-date electronics for lab and leisure summer circuits ’62 more than iOO ractical projects slektor july /august 1982 - 7-03 I I 54 a 'MID-FI' receiver 55 low cost temperature indicator 56 duty cycle meter 57 AC motor control 58 pulse generator 59 oscilloscope aid 60 mini EPROMmer 61 5 V super power supply 62 short wave converter 63 a simple window comparator 64 symmetrical opamp supply - J. Wallaert .... 65 monoflop with a CMOS gate 66 electronic thermometer 67 fluid level detector 68 voltage controlled waveform generator 69 economical battery tester 70 telephone bell 71 CMOS switch Schmitt-trigger 72 universal VCF 73 keyless lock 74 digital logarithmic sweep generator - J. Meijer 75 car lock defroster 76 LED tuning indicator 77 calling Junior vectors - R. Matyssek 78 RTTY converter 79 single cycle mode for the Junior Computer - E. Kytzia 80 super low noise pre-amp 81 crystal oscillator 82 infra-red remote control receiver 83 voltage controlled TTL oscillator - N. Rohde . 84 RS 232 interface 85 magic running lights 86 stable start/stop oscillator 87 sound effects generator 88 VCOTA . . 89 biomedical interface 90 dissipation limiter - H. Burke 91 stereo power amplifier 92 power failure protection 93 12 dB VCF 94 voltage controlled filter 95 simple frequency converter - R. van den Brink 96 high performance video mixer 97 rear light monitor 98 connection tester - P. Verhoosel 99 AC/DC converter 100 high speed printer routine - F. de Bruijn 101 phase sequence indicator - F. op 't Eynde . . . 102 the Elekterminal with a printer - E. Francois . 1 light sensitive switch 2 DC motor speed control 3 polystyrene cutter 4 summer circuits power supply 5 simple AGC 6 high voltage converter 7 slave flash trigger - G. Kleinnibbelink 8 temperature to frequency converter 9 frequency generator 10 signal strength meter 1 1 inverter oscillator 12 serial keyboard interface 13 R F amplifier for the 1 0 metre amateur band 1 4 active attenuator 15 executive decision maker 16 automatic outdoor light - J. Bodewes ; 17 slave flash - G. Konig 1 8 555 pulse generator 19 pushbutton interface - J. Ritchie 20 true RMS converter 21 miniature amplifier 22 OTA Schmitt-trigger v/23 TRS 80 cassette interface rediscovered .... 24 mixing console 25 low voltage stabiliser 26 overvoltage protection for meters 27 stable amplitude low frequency oscillator . . 28 positive triangular waveform generator - « R. Storn 29 smoke detector 30 reciprocal amplifier 31 blinky - J. Meijer 32 double alarm - M. Prins 33 automatic delay switch - M. Prins 34 dynamic RAM for SC/MP - D. Paulsen 35 economical crystal time base 36 FET field strength meter for output amplifiers - 37 automatic switch W. Wehl 38 mini high performance voltage regulator 40 converter for varicaps 41 low octave switch 42 program EPROM 43 infra-red remote control transmitter . . . 44 high-speed NiCad charger 45 logic probe 46 high quality tape playback pre-amp . . . 47 square triangle VCO 48 graphic oscillator 49 analogue monoflop 50 the simplest PDM amplifier 51 class AB amplifier 52 omnivore LED L 53 EX(N)OR opamp - A. Rochat We regret to inform readers that the technical queries telephone service will not be operating during the months of July and August. elsktor july/august 1982 - 7-13 ... no matter what it says on the cover. Each year, we produce ten issues of Elektor: January to June, and September to December. For July and August, we print something completely different: the Summer Circuits issue! It's not really a magazine, because there are too many circuits and too many pages. But it's not really a book, either, mainly because the size is wrong (although lots of people use it as a kind of quick reference book for new circuit ideas). So what? As one philosophically-minded genius proclaimed, after several hours of deep and profound thought: "It is what it is". Even if we don't know what it is, at least we know what to put into it: 'more than 1 00 circuits'. Each year, we try to do better than the year before. Last year we stated, truthfully, that 'nearly all circuits have been built and tested' — the exception being a few simple application notes and some straightforward ideas from external authors. This year, we built and tested the lot! Although ... no, that's not quite true. Another of our traditions is to include one 'joke circuit'. Last year (for the few readers who didn't spot it!) we published a solar-powered torch. This year ... no, work it out yourself. We can give one clue: we think the circuit should work, but we can't see how to actually test it in the situation for which it is intended. Tradition and progress. This issue is 'traditional' — we've been doing it for years — and the quality and diversity of the circuits is even better than last year (we think), so that's 'progress'. Next year, maybe, we'll try to make the text better: cram even more valid information into the number of lines available. Maybe even improve the grammar? You never know: ten years from now this issue may be required reading for 'Arts' students. So, what's new? An 'editorial introduction', in the best tradition (and that eliminates a large number of editorial introductions . . . ) should contain something more than light reading. Let me think ... we must have something . . . Electronics in the future? Difficult ... we try to convert our futuristic ideas into something practical, and simply publish it as a circuit. Next month's ideas, maybe? No let's surprise our readers with that dark-room computer, way-out hifi system and 16-bit-microcomputer . . . Talking about computers, there's one point: 'hardware', 'software' and even 'firmware' are known — but have you ever heard of 'paperware'? No?! Well then, that's new! Take a look on page 90. What else? Oh yes! I almost forgot. Our front panels! We've had a 'front panel service' for quite some time, but it never really satisfied us. Either a panel is expensive, or else you can see that it is not so expensive. Now - at last! — we think we've solved it. Professional front panels at a price that came as a pleasant surprise to us. As a first shot, we've got a panel for the Elektor 'Artist'. If that one works out as we expect, our 'printed circuit board service' may well become a relatively traditional side-line. The new 'front panel service' could well lick it hollow, as regards 'uniqueness' (or should that be 'uniquity' or 'unicitude'? As stated earlier, grammar is scheduled ten years from now.). Now — forgive me! — I intend to stop. There's a hot soldering iron beside me, and I want to use it. That no-l-won't-tell-you-which circuit intrigues me . . . Your editor. 7-14 - elektor july/a 1982 There is a wide range of applications for light sensitive switches: staircase light timers, outdoor illumination, automatic door openers by means of a light beam, alarm systems and so on. Many of our readers will be familiar with the single transistor opto-switch where a LDR is placed between the base and either ground or supply depending whether a 'normally on' or 'normally off' function is required. This simple circuit gave way to more complex arrangements involving the use of opamps with the advent of the supercheap 741 ! Another, not so well- known, method of opto-detection uses a bridge circuit operating on the principle that current flow across the bridge will be zero when the four impedances have been calculated correctly. The 'bridge is in balance' when this occurs. The latter principle is used in the circuit here. The opto-detector is situated in a bridge circuit and a comparator is used as a 'bridge is in balance' indicator. The comparator output fires a thyristor via a transistor. Caution must be used with this circuit, since it is not isolated from the mains supply. Power to the circuit is derived via the bridge rectifier D1 ... D4 and is smoothed and stabilised by means of R1 , Cl and D5. The bridge circuit may be difficult to see in the circuit diagram, but it consists of R2 . . . R4, PI and the light dependent resistor (LDR). IC1 is connected as compara- tor and its output voltage level will become approximately 1 .8 V when the potential at the inverting (negative) input exceeds that of the non-inverting input. Resistor R5 creates an 'hysteresis' of about 1 V to prevent T1 and the thyristor from switching 'on' and 'off' (flickering) in marginal light conditions. The switching point of the comparator is adjustable by means of PI . With this potentiometer set to minimum resistance, the lamp will switch on at twilight. Readers who require greater flexibility can replace PI by a 1 Mft type. The LDR can be exchanged with the P1/R4 combi- nation to provide the circuit with 'inverse law'. The lamp Lai will be extinguished at the onset of darkness. Some practical considerations: For switching higher power lamps D1 . . . D4 must be replaced by 1N5404 types and a heat sink must be used for Thl. With these modifications the circuit will cater for current levels up to 3 amperes. The maximum gate current available for Thl is 250 /iA, which means that a fairly 'sensitive' thyristor should be elektor july/a 1982 - 7-15 Resistors: R1 = 100 k/1 W R2,R3 = ICO k R8= 33 k R9 = LDR 03. 05 or 07 100 k preset potentiometer Capacitor: Cl = IOOp/16 V Semiconductors: D1 . . . D4- 1N4004 (1N5404I 05 = zener diode 10 V/400 mW T1 = BC547B IC1 = 741 Thl “ TIC 106D Any LDR should be suitable. There is no apology for repeating the cautions regarding the lack of isolation from the mains supply. With this in mind it is essential that the completed circuit is housed securely in some form of plastic box. A hole can be made in the top of the box for the LDR to 'see' through. Make sure that both the input and output cables are fitted securely. These precautions will ensure that prying fingers will not come to grief. H The LM 1014 1C from National Semiconductor can be used to provide a constant speed control for small DC motors. A well known trick is used here. This takes into consideration the fact that when the motor current rises ' (due to an increase in load) the voltage across the motor will follow suit. The reason for this is that if the motor speed drops slightly the back EMF decreases which means that the motor current (given the same supply voltage) is going to increase. It follows that raising the voltage across the motor will increase the speed. Theoretically then, it is possible to hold the motor speed virtually .constant in this way. 'However, in practice this system has a Vref AV ret MT (VI (mV/°C) 0.95 -1.0 1.15 -0.3 1.35 +0.3 1.55 +1.0 2 gnd P 3 open 2 open, 3 gnd 2/3 gnd tendency to be unstable and the only way to keep it within acceptable limits is to allow slight speed variations in the order of a few percent (depending on load conditions). A disadvantage of the circuit is that the value of the components required cannot be given as hard and fast. It is a circuit then that does require some experimenting with in order to obtain the best results. The values of resistors R1 , R2 and R3 should be selected so that is equal to the dynamic impedance of the motor. How do you find this? A good start for the calcu- lation is to simply measure the resistance of the motor with a multi- meter and start with this value. Choose R1 to be slightly on the low side from the formula and check whether or not the motor is still controllable. As long as it doesn't run wild (run up to maximum speed and stay there) or start hunting, R1 can be increased in The output voltage, and with it the speed, can be adjusted by means of PI . The formula for the output voltage is given in the diagram. Before calcu- lations are begun, a reference voltage must be selected via pins 2 and 3. Each reference voltage has a different temperature coefficient (see table). This parameter of the motor will rarely be known and so the choice will come down to personal taste. The value of PI is not really critical. This potentiometer at minimum value will certainly give maximum volts supply but using too small a value will only render it impossible to slow the motor very much. The choice of R1 not only determines the dynamic characteristic of the circuit but also limits the maximum motor current. With the value shown in the diagram (1 S2) the maximum current will be 1 .4 A. The values given were actually used with a motor that was measured as follows: Dynamic resistance: 16.3 fi Reverse EMF: 3.25 V at 2000 rpm Torque: 5.9 mA per mnm National Semiconductor Applications july/august 1982 polystyrene cutter hot wiring for beginners Have you ever tried to cut polystyrene panels or blocks with a conventional saw? Messy is it not? Little bits of the stuff everywhere and you still have not achieved what you set out to do. The only way to cut polystyrene ef- ficiently is by the hot wire method. The wire has to be kept at just the right temperature otherwise it will either not cut or it will burn the ma- terial into horrible little black bits. A low voltage transformer delivering a reasonable current of approximately 2 A is sufficient for the circuit. By controlling the current flow through the wire the actual temperature can also be regulated. In order to reduce the consumption and power dissipation the current is switched on and off intermittently by a triac. One side of the 'hot wire’ (represented by R L) is connected directly to the secondary winding of the transformer. N1 and N2 ensure that the sine wave (A.C. voltage) supplied by the transformer is con- verted into a square wave. In order for this to happen the values of R2 and R3 are calculated so that N2 switches on and off in phase with the A.C. supply. The RC network R4, C2 differentiates the positive pulse, the internal clamping diode of N3 suppressing the negative pulse. N3 and its surrounding components form a time switch which in turn controls the triac. The switching periods are determined by C3. This capacitor is charged by way of PI , and discharged by way of R5 and D3, to the output of N3. The charge and discharge levels of C3 are within the threshold levels of the Schmitt-trigger N3. It therefore follows that the voltage across C3 will either be logic 1 or 0. With a logic 1 , N3 receives a positive pulse front N2 resulting in a short negative pulse at its output. This triggers N4 and in turn T1 , which switches on the triac. The RC network R6/C4 ensures that the triac conducts for one complete mains cycle. The negative pulse also causes the voltage across C3 to drop below the level of the trigger threshold of N3. Keep in mind that the time frame for all this to happen can be varied, by adjusting PI . N3 now no longer reacts to the pulses from N2,so its output remains at logic 1 . C3 can no longer discharge via R5 and D3 and therefore the triac will switch off. After a defined period of time (set by PI ) the voltage across C3 is logic 1 once more and the pro- cedure starts all over again. The wave- form across the triac is shown in the illustration. As already mentioned R6 and C4 ensure the triac conducts for one complete mains cycle. By doing so SH : ,3 the loading of the transformer is symmetric, reducing the need for high DC currents. It should be noted that the total resistance of the cutting wire should not exceed 5 ft. Construction can be similar to the drawing where a fret saw frame has been used (with insulation!). ihsfcrrna slektor juty/august 1982 - 7-17 The title means what it says! A power supply specially designed for use with our summer circuits. The novelty of this design is that it has a variable out- put from 0 V up, without using a transformer with two secondary windings. The circuit can either be constructed using the well known 723 1C, or for higher output voltages the L 146, which although less popular, is still easily available. The choice is left to the constructor. The output current limitation is also variable, but once set it is continuously effective. Table 1 shows all the different component values needed to make three different versions (30, 40 and 60 V maximum output). The circuit diagram actually illustrates the 40 V/0.8 A type. The L 146 1C was used because this can handle the higher output voltages far better than the 723. Normally speaking 2 V is the minimum regulated voltage which either 1C can provide. The resistor networks R3, R4 and R5, R6 get over this restriction allowing the output to be adjusted right down to practically 0 V (with the aid of P2). these resistors ensure, that sufficient voltage is present at pins 4 and 5 of the regulator (thereby keeping it stable), even when voltages lower than their tolerated input level are required. ^Another aspect of the design which R1 R4.R5 R9 0-25-30 V 1.3 A 0.47 H (MOV 0.8 A 0.82 Jl 0-60 V 0.6 A 1.2 n 2k7 24 V 2 A 5k6 33 V 1.5 A 10k 48V 1 A C1/C5 IC1 T2 T3 / 723 BD 242 2N3055 / LI 46 BD242A 2N3055 / L146 BD242B 2N3442 'strikes the eye' is the unusual way in which T3 is driven. As a result, a closer look at the way the circuit works is called for. When the required output voltage is below the tolerated minimum of the regulator, the actual voltage potential at pin 4 is below that of pin 5. This results in the 1C trying to compensate for this by attempting to increase the output voltage from pin 9. This, however, will not work simply because pin 9 is earthed via R7 and D2, thereby limiting the voltage increase. Although the voltage cannot increase, the current certainly can, so R7 is also used to limit this to 6 mA. The current flowing through the 1C (in at pin 1 1 and out at pin 9), causes a voltage drop across PI . This in turn drives T3 open (by way of T2), therefore increasing the voltage. As the wiper of PI is connected to T1, it can be used to control the current limitation. 1N4001 When the voltage drop across R 1 exceeds 0.6 V, PI is shorted out by T 1 , and T3 is cut-off. During a normal operation (without current limiting), the voltage drop across PI is a constant 1.2 V, made up of the flow voltage of D1 and the Ube of T2. A part of this voltage can be used to drive T 1 before 0.6 V is reached across R 1 . This is possible because the base voltage of T 1 is composed of the drop across R1, and the divided voltage value at the wiper of PI . In the way just described the output current can be controlled from 0 to the maximum available, quite easily. Keep in mind that a 723 can only handle a maximum of 36 V. An L 146 should be used with any transformer supplying more than 24 V. As the L 146 can safely handle up to 80 V, the maximum size of transformer that can be used is one with secondary windings supplying 48 V. Whatever output requirements the constructor decides upon, must also determine the type of capacitors and semiconductors to be used. Remember that a 2N3055 is only rated to 60 V, therefore for 80 V a 4041 1 or 2N3442 should be used, and so on. T able 1 indicates the component values needed to construct three different power supplies dependant on the voltage range required. The most important factor to bear in mind is to limit the output current sufficiently to keep the power dissipation of T3 under 40 W. The maximum output of a 40 V version is 0.8 A. It is possible to connect two 2N3055's in parallel (with emitter resistors) , to double the output current, but, then a 2 A transformer is necessary. 1982 This circuit will provide an output with a fairly constant amplitude of 4 V peak to peak, from an input that may vary between 1 00 mV to 2 V. There was no intention of achieving 'hi-fi' performance as the distortion figures are not exactly in that league. Nevertheless, this automatic gain con- trol is ideal for use when recording computer programs onto cassette tape where a constant amplitude is more important than low distortion. Opamp A1 provides an output impedance that is sufficiently low to drive the attenuator formed by diodes D1 and D2. Opamp A2 is a straight- forward amplifier with a gain of lOOx but its DC setting is a little unusual in that it is derived from the average of the input signal via R5 and C4. The off -set voltage of A2 cannot escape being modified to some degree but, simply a cascade rectifier out of an old T.V. set. Version 2 delivers a voltage three times higher than version 1 because the cascade rectifier acts as a voltage multiplier (3X). IC2 regulates the output voltage. The opamp compares the voltage across PI with that at the junction of the voltage dividers R6/R8 or R7/R8. If the out- put exceeds the preset voltage level, IC2 will reduce the supply voltage to the output via T3. The most important part of the circuit is the transformer. Even though it is rather essential, its construction is not that critical. A variety of E, El or ferrite cores having a diameter of 30 mm can be used quite easily. The core should not have any air gap; an AL value of 2000 nH is about right. The primary winding consists of 25 turns of 0.7 mm ... 1 mm enamelled copper wire and the secondary is 500 turns of 0.2 . . . 0.3 mm wire. The primary and secondary windings must be properly insulated from each other! With respect to the high voltages the constructor should pay special atten- tion to the following points: • Capacitor C6 must be able to cope with at least 3 kV. • R6 in version 1 consists of six 10 Mn resistors in series. R7 is made up by using 10MS2 resistors, also in series. This is done in order to avoid spikes at the output. Either circuit consumes approximately 50 mA without a load and 350 mA when delivering 2 ... 3 W into a load. Transistors T2 and T3will require heat sinks. slave flash trigger A triggering circuit for slave flash guns ensures that the 'slave', flashes simul- taneously with the main or master gun. Apart from the commercially available units, there are quite a few circuit designs published in electronic magazines. Unfortunately most of these have one major drawback. They all need some form of power supply, such as normal batteries etc. The circuit design described in this article uses a virtually in-exhaustable supply! Solar cells are applied here in an ingenious way I The flash of light amitted by the master gun will trigger the slave. The small delay which occurs is so small (in the order of 1/1 000th of a second), that it is virtually undetectable, by the human eye. The circuit consists of a sensitive low powered thyristor, in this case the TIC 106D (Thl ), and a choke. The solar cells (which should have a minimum surface area of 1 00 mm 2 ) are connected in series. They generate the ignition pulse for the thyristor immediately the master flash is fired. A 68 mH choke ensures that the circuit is insensitive to ambient light. The prototype achieved an operating distance of 50 metres, between the slave and a master flash gun with a power figure of 28! temperature to frequency converter Although a temperature to voltage converter may be more common, a temperature to frequency converter is much more useful when digital circuits are used for temperature measure- ment. This type of converter can be connected to either a frequency counter or even a microprocessor, without the need for an additional A/D converter. The circuit described here is remarkably accurate. A 10 Hz/°C conversion factor is maintained within 3 Hz, throughout the 5° to 1 00°C range. A 'pseudo' zener diode, the tempera- ture dependent LM 335, is used as the temperature sensor. The 1C comes in a plastic transistor package. The ADJ pin is not used in this application. The voltage across this zener diode is directly related to the absolute temperature in degrees Kelvin: U|_M 335 = 10 • T (mV) Therefore, at 0°C the voltage will be exactly 2.73 V. In order that the voltage to frequency converter can b calibrated in degrees centigrade, this 7-20 - elektor july /august 1982 2.73 V input can be cancelled by an equal and opposite (negative) voltage. Instead of using a negative supply voltage for this, a little trick is employed. A +5 V regulator, IC3, boosts the GND connection of IC1 to +5 V with respect to supply common. The input offset can now be taken from preset PI . At the other end, the LM 335 is fed by the current source around T1. The output of the LM 331 (IC1) is a square wave, swinging from +5 V (GND for this 1C!) to positive supply. It is not difficult to relate this signal to the actual 0 V rail: two switching transistors, T2 and T3, take care of this level conversion. T3 has an open collector output, so that it can easily be used to drive TTL or CMOS logic circuitry. Alternatively, frequency counters with an AC input can be connected direct to pin 3 of IC1 and T2 and T3 can be omitted. To calibrate the circuit, a mixture of crushed ice and water gives a good 0°C reference. With the sense- in this slush, the voltage between the positive end of IC2 and pin 4 of IC1 (GNDi can be set toOVby means of PI. A further reference is now required at approxi- mately mid-scale — warm water at 50°C, as measured with a good thermometer. (Alternatively: approxi- mately 37°C — there are very accurate thermometers in this range . . . ). The output frequency is then set, with P2, to correspond: 370 Hz at 37°C, say. For good temperature stability of the circuit, metal film resistors should be used for R5 . . . R7,and a poly- carbonate capacitor for C4. Preferably, PI and P2 should be Cermet helical potentiometers. One final point. If the circuit is used to measure air temperature, this will invariably imply that the circuit itself will also be warmed up. In this case, the output may drift up to +0.5°C off mark. The solution is to . . . recalibrate the thermometer! Alternatively, try and keep the circuit as cool as possible, using plenty of heatsinks. M One 1C, a quartz crystal, three Table: resistors and two switches are all that is required to obtain 16 different frequencies! Can it be more versatile than that? Motorola calls its 1C MCI 41 1 a 'bit rate generator' which can be used as a frequency source for numerous applications within the area of data transfer, such as teleprinters, video terminals and microprocessor systems. A quartz controlled oscillator r~ pt Pin Output Number Number 1 FI 17 F2 2 F3 16 F4 3 F5 15 F6 4 F7 5 F8 7 F9 6 F10 8 F11 14 F12 13 F13 9 F14 18 F15 19 F16* Output Rates (Hz) X16 X8 X64 614.4 k* 460.8 k* 307.2 k* 230.4 k 153.6 k 115.2 k 76.8 k 38.4 k 19.2 k 12.8 k 9600 8613.2 7035.5 4800 921.6 k 1.843 M‘ 153.6 k 115.2 k 76.8 k 57.6 k 38.4 k 28.8 k 19.2 k 9600 4800 3200 2400 2153.3 1 758.8 1200 921.6 k 1.843 M 76.8 k 57.6 k 38.4 k 28.8 k 19.2 k 14.4 k 9600 4800 2400 1600 1200 1076.6 879.4 600 921 .6 k 1.843 M *F16 is buffered oscillator ol XI 9600 7200 4800 3600 2400 1800 1200 600 300 200 150 134.5 109.9 75 921.6 k 1.843 M july /august 1982-7-21 forms the 'master frequency source'. The oscillator signal is buffered at pin 19. Moreover, the signal reaches a divider that produces five different output signals: The oscillator signal divided by two is always present at pin 18, the other four signals (:1, :4, :8, :64) can be fed to a 14 stage divider, as desired. So, with the two switches (SI , S2) in the open position it already supplies 4 different signals. In addition there are 14 + 2 signals simultaneously available. The table shows all the possible combinations. The output pins of the 1C are not indicated in the circuit diagram, but are found in the table. One final remark: The 1C can be 'fed' with an external clock signal via pin 21, so that the various division factors can be used to the full! \ r signal strength meter ) K. yv yv y with audio output A meter of this kind is very useful for determining the radiation characteristics of directional beam transceiver aerials. It allows the user to trim the aerial accurately for an optimum transmitting radiation pattern. An auxiliary aerial should be positioned a short way from the main transmitting one. The signal received by this is then fed to a resonance circuit formed by LI, L2 and the L varicap C2. This enables the meter to be accurately tuned to the particular transmitting frequency to be measured. With the coil values shown the circuit diagram the 'band width' of the meter is between 6 ... 60 MHz. The R F signal is then fed to the diode D1 , which constitutes a rectifier/demodulation stage. Finally the signal is routed to the non- inverting input of opamp 1C 1. The gain of this opamp and therefore the sensitivity of the 1 mA meter is adjusted by PI . The prototype was found to be extremely sensitive, and highly selective. A pair of headphones can be connected to the output of the opamp allowing the actual transmission to be monitored. The overall resistance of these should not be less than 2k2 otherwise an extra amplification stage will be required. inverter oscillator ' can be crystal controlled Not another TTL squarewave gen- erator?! Surely, there are plenty of them in other issues of Elektor? Yes, but this is an oscillator with a difference: unlike most of its counterparts its frequency is vari- able. In fact it may be adjusted over a wide range. The circuit shown here consists of two inverters with one or two ex- ternal components. Resistors R1 and R2 and the trimming capacitor Cl set the frequency. With the ven component values, the oscil- lator frequency may be adjusted from 800 kHz to 1 2 MHz. The resistors set the frequency in just about the right region, whereas Cl provides the fine ad- justment. The resistor values are not really critical; just make sure that they are both the same. The circuit is also suitable as a stable crystal oscillator. All you have to do is replace the trimming capacitor with a crystal with the corresponding frequency. Sup- posing, for instance, the oscillator frequency is to be 1 MHz, then the crystal will have to be a 1 MHz type. st 1982 With a bit of luck it is sometimes possible to purchase a high quality keyboard without having to pay too much for it. Most of these keyboards have a parallel output that supplies an ASCII or Baudot code. Trying to connect it to a personal computer will cause some problems because most computers are equipped with a serial RS 232 interface. The circuit described in this article will provide the solution to this problem; It converts a parallel ASCII or Baudot code into a serial signal. The signal conversion is performed by a UART of which only the transmitter is used. The Baud rate is produced by a clock generator which is constructed using the well-known 555 timer. The clock frequency must be 16 times the Baud rate. The serial data signal is situated at pin 25 of the UART and is boosted to the RS 232 level by way of transistor T1 . The lenght of the serial 'word' can be set with the aid of the logic levels at pins 37 and 38. The logic level at pin 35 of the UART determins the setting transmitted parity or 'no parity'. With the circuit diagram shown in figure 1 the data word will be 7 bit long and will not contain a parity bit. (As a result pin 39 is not used.) Literature: 'Elekterminal': Elektor December 1978, p. 12-16.. . 12-25 m RF amplifier for the 10 meter amateur band The VN66AF manufactured by Siliconix has quite a few advantages, over its rivals; good value for money, in terms of price per watt, high dielectric strength and exceptional gain. It also has a low tendency to oscillate. The most common appli- cation for VMOS FETs is in power amplifiers, but, that is not a reason to discount them for any other use. They have been used successfully in pre- amps, and RF amplifiers. In this 12. 15V particular case it is used as an RF amplifier for the 1 0 metre amateur band (26 . . . 30 MHz). Small transmitters of around 200 mW can be transformed into reasonably powerful ones delivering between 2 and 3 W by using the circuit described here. The design is fairly straightforward. The fixed filter network positioned at the output, suppresses noise by as much as 55 dB. If the coils are constructed to the specifications outlined in the parts list, then the filter will not require calibration. Obviously experienced hands may wish to change the specifi- cation and the design is sufficiently flexible to allow this. The amplifier is suitable for most types of transmission mainly because the drain current from the F ET can be varied, by PI . For linear applications (AM and SSB), the drain should be set to 20 mA. When used for FM and CW, PI should be adjusted so that no quiescent current is flowing. For the application that the original design is meant for the quiescent current should be between 200 mA and 300 mA. The ready made printed circuit board ensures speedy and accurate construc- tion. The coils should be wound onto |®o%o|K>3 ®| l or w * r 'o „ aerial coil formers with a diameter of 9 mm. Care should be taken to lay the windings close together without any apparent gaps. It is advisable to use a heat sink for the FET. Capacitors: C1.C2- 1 n ceramic C3.C4 ■ 1 50 p cararr C5 “ 47 p C6 = 1 0 u/35 V tant C7 = 22 n ceramic ' juty/august 1982 — 7-25 Needless to say this source needs to be close by. Please remember that the removal or repositioning of lamp posts needs the authority of the local coun- cil, so we do not recommend this circuit to anyone who has to extensively remodel the landscape. The LDR is mounted into a tube, behind a lens, and aimed at the light source. This structure is positioned, so that the person approaching the front door, causes a shadow to fall onto the lens. Do not forget to ensure that the tube containing the LDR is water tight. Immediately the LDR is in shadow, its resistance will increase. This results in T1 applying a negative pulse to T2 via Cl and R6. T2 con- tinues to conduct until this negative pulse arrives. As soon as T2 cuts-off, C2 starts to charge. When the voltage across C2 rises above 2 V, the schmitt-trigger formed by T3, T4, T5 (and their surrounding components), switches on transistor T6. T6 conducts and triggers the relay, which switches on the outside light. The rate at which C2 discharges is adjusted by PI . When the voltage across C2 falls below 1 .5 V the schmitt-trigger returns to a quiescent state. T6 will cut-off switching off the relay and therefore the light. The light will remain on for a maxi- mum of one minute. Longer periods are possible, but then C2 will have to be substituted with a larger capacitor. Switch SI and R3 are connected in parallel to R2. SI can be a make/break contact mounted on the front door. When the door is opened the light will switch on, going out immediately it is shut. In order for the circuit to work effec- tively, the tube containing the LDR (and lens), must be positioned, relative to the light source, so that the voltage ^measured at the junction of R1, R2, is not less than 3 V, and not more than 20 V. 7-26 - elektor july/august 1982 fast, sensitive and reliable Electronics have been making signifi- cant inroads into photography for some time now and, judging by the number of requests we receive, many of our readers want to push the frontiers even further. However, there are a few things that even we dare not do and dabbling with the insides of an electronic camera is one of them. One of the most repeated requests is for a flash slave unit and the super fast, super sensitive (and super insensitive) circuit here takes care of that. This can be used for any application of flash photography indoors as well as outdoors. The apparent confusion between super sensitive and at the same time super insensitive is easily explained. The slave unit is super sensitive to the master flash gun, but super insensitive to the ambient light conditions. It will react within about 10 /js depending on the light power of the master flash gun. This means that when using a computer controlled flash gun with a flash duration of 1 ms, 99% of the slave flash is included in the computer's calculation. This makes it especially ideal for use with automatic flash/camera systems. The total range of the slave is set by means of T1, R1, R2 and D1. The setting is to achieve maximum sensi- tivity in low and average light levels. A special shield for difficult light conditions is not normally required. However, if the slave is to be used for daylight fill-in flash photography then a certain amount of protection from sunlight will be advantageous. On the other hand, switching a normal incandescent lamp on and off in the same room will not trigger the slave. parts list R2. R6 - 100 k R3, R8 * 10 k R4 = 22 k R5, R9 - 1 k R7 - 33 k RIO - 390 n Capacitors: Cl - 10 p/16 V tantalum C2 * 10 n ceramic Semiconductors: 01 - Z-Diode 3V9/0.4 W 02, D3= 1N4148 T1 = BPY 61/11, FPT 100 T2, T3 = BC 557 C Thl -TIC106D There is very little to be said about the circuit itself and photographers with sufficient electronic know-how will be satisfied with the following information. A brief flash from the master reaches photo transistor T1 and causes a pulse at the base of T2. This pulse is boosted and passed via T3 to the gate of the thyristor. When the thyristor fires, it effectively shorts the contacts of the flash gun which is connected at this piont. For the electronics enthusiast with an interest in photography we can say a little more. The slave flash gun is connected in parallel with the thyristor. Apart from this a 9 V compact battery is required and should last for quite a long time. The resistors are mounted vertically on the printed circuit board in order to keep the board as small as possible. One further tip, for the connection to the slave flash gun use ... a flash gun extension cable) M 555 pulse generator . with variable duty cycle This circuit may look familiar to many between PI and P2: n = 1 + P2/P1 . readers since it is one of the many variations of circuits on the 555 timer theme. This does not detract from its usefulness however since a versatile pulse generator with a variable duty cycle is an excellent aid for the workshop. Unlike the standard circuit usually adopted (see infocard 19), the ’resistance between pins 6 and 7 consists of P 1 , P2, R 2, D 1 and 02. A closely defined charging time for capacitor Cl is obtained by diodes D1 and D2. This would normally lead to a i duty cycle of 50%, if it were not for V P2. In this case the duty cycle ^^depends on the relationship For example, if P2 = 0 (n = 1 00%), the frequency will then be: 069 ~ (2 • PI + P2 + 4.7 kS2) - Cl pushbutton interface ' combined debounce and latch fuction The circuit here extends the effec- tiveness of the simple push-to-make tewitch by enabling it to be used as either a 'one-shot', with clean, de- bounced edges, or as a push-on/push- off latch. These functions remove the problems associated with any switch, that of electronic 'noise'. Resistor R1 and capacitor Cl 'de- bounce' the switch and provide a positive edge to trigger the monostable FF1 . This generates pulses (in anti- phase) at its Q and Q outputs. The pulse width is determined by R1 , R3 and C2. The positive pulse (Q) is fed to an 'OR' gate consisting of D2, D3 and R5. The trailing edge of the negative pulse is used to trigger flip- flop FF2. The normal (or stable) state of FF1 is with its Q output low and (logically enough) its 0 output high. In this condition, if the switch is closed briefly and then released, the J and K inputs will be low when the triggering edge arrives. In this case FF1 will ignore it and stay reset. If, however, the switch is held closed until the monostable 'times out' and FF2 is clocked, the J input is taken S,oo. FF1 ici high, K low and the flip-flop 'flops'. Now Q and the output (via the OR gate) are high, and consequently, so is K. If the flip-flop is triggered with K high and J low it will revert to its reset state. Holding the switch closed will not affect the circuit action, for with both J and K high, FF2 will change state on the arrival of a clock edge. true RMS converter .. . requiring no special components A true RMS converter can be a very complex circuit requiring high tolerance components and precision calibration. It is fair to say that such a circuit would give a very high performance. The RMS converter here, however, consists entirely of readily available components and yet provides a very acceptable performance. The circuit diagram shows that the RMS converter really is an automatic gain control (AGC) amplifier circuit, which is constructed around 2 ICs, the well-known XR 13600 (A 1,A2) and the XR 1458 (A3, A4). The circuit adjusts its gain so that the A.C. power of amplifier A 1 remains constant. This output level is monitored by the squaring amplifier formed by A2 and the average value is compared to a reference voltage with the aid of A3. The output of this amplifier provides the diodes of A1 with bias current, via a 2 kfi resistor and transistor T 1 , in order to attenuate the input signal. As mentioned before, the output power of A1 is held constant, therefore the RMS value remains constant as well. Obviously the attenuation is directly proportional to the RMS value of the input voltage and the diode bias current. This leaves only the function of A4 to be discussed: This amplifier adjusts the ratio of current flow through the diodes, so that they are equal, Con- sequently the output voltage of A4 corresponds to the RMS value of the input voltage. Last, but not least, the potentiometer situated at the input of this circuit, must be set so that the Vo reads directly in RMS volts and can be calibrated by direct comparison with another RMS voltmeter. miniature amplifier . . with active tone control There are many ICs available today that contain all the circuitry required for various versions of power output stages. The 1C presented here goes further than that. It can be used complete amplifier. Obviously it is not super hi-fi but for a second (or third) amplifier it is quite good enough. The 1C LM 389 was used in the Summer Circuits issue last year. In that case it formed the basis for a small siren. The resemblance between a siren and an amplifier is quite obvious and the natural progression is published here. The 1C contains a small power output stage and three further transistors on the same chip. This means that no elektor july/s 1982-7-29 further active components are required for the amplifier. The gain of the output stage is simply set by means of a capacitor and a resistor. In the cir- cuit diagram the gain is set at 20x (26 dB) which means that pins 4 and 1 2 are simply left floating. If a 1 0 pE capacitor is connected between these pins, the gain increases to 200x (46 dB) and 50x if a 1k2 resistor is inserted in series with the capacitor. Transistor T1 is used as an emitter follower (high input impedance/low output impedance). This sets the input impedance of the circuit to approximately 50 kft. The so-called Baxandall tone control is formed by the networks R5 . . . R8, C4 . . . C7 and PI and P2. Transistors T2 and T3 are the active part of the tone control circuit and ensure a gain of 1 to 1 in this stage. The signal is then fed to the power amplifier via the volume control P3. The output stage is not given in detail here, but simply as a block, IC1. The maximum output power into a 4 n load is about 300 mW with a dis- tortion figure of 1 0%. With an 8 El load this becomes 600 mW again with 10% distortion. If the maximum out- put power is required with a 12 V supply, it is advisable to use a heat sink for I Cl. Readers who would pre- fer a lower distortion figure can achieve this by limiting the power output to 1 20 mW. This presents a reasonable distortion figure of 0.2%. The minimum input voltage for maxi- mum output is approximately 100 mV for a 4 n load and 1 50 mV for an 8 El load. Obviously modifying the gain is going to alter the input sensitivity up to a factor of 1 0. When constructing the circuit a few points must be watched. Pin 18 of the 1C is connected directly to the central earth connection of the circuit, in this case 0 V of the power supply. The loudspeaker must also be connected to this point. National Semiconductor Application If the voltage at the differential input of an OTA, such as the XR/LM 1 3600, is strongly positive or negative, the output current will equal the maxi- mum value lABC- Furthermore, a Schmitt-trigger with trigger points to the value of ± lABC • RV is obtained when the output voltage (across load resistor R) is identical to the voltage at the positive input. Therefore the switching hysteresis depends on lABC : hysteresis = 2 • I ABC • R volt I The control current I ABC can be V influenced by changing the value of V Rq. Alternatively, a control voltage (U c ), can be connected across R c , so that a voltage controlled hysteresis is obtained. . 2 • R . (U c + 3.8) . hysteresis = volt Exar/National application H The TRS80 computer is a fairly good recognised as a clock pulse, and from completely unimportant. The rectified machine, but the cassette interface has this point onwards the whole thing signal is passed on to the filter and also already driven many an owner to the gets totally out of hand. The situation a peak detector D3/D4 and C2 When depths of despair. Why the tapes are deteriorates still further when playing the amplitude of the cassette deck out- read back so unreliably has never been back commercial tapes. These are put varies a little (when an older or a worked out, but despite this fact there very often recorded at high speed, and different type of tape is used! no are a number of suggestions on how to this has the effect that there is not so critical adjustment of the output level improve matters. The circuit given much a pulse on the tape as a damped is required . here also produces good results, but as sine wave. In all fairness, most home The filtered signal is compared in A3 with so many good suggestions we do recorded tapes may not appear very with part of the peak rectified signal, not really know why. elegant when viewed with an oscillo- In this way the comparator becomes The TRS80 records clock pulses and scope during playback. independent of the input amplitude data pulses on the tape at a constant The following circuit attemps to solve (within reasonable limits) . This means amplitude. The time interval between all these problems by integrating the that P2 must be used to set a suitable pulses is 2.4 mS. The logic is written signal coming back from the tape level so that the data arrives 'clean' at by inserting a pulse between two clock recorder. This has a few advantages. the output. The combination C6 and pulses after 1 .2 mS. If this pulse is not Short interference pulses are filtered RIO converts the data into short there this signifies a logic 0. The ironic out by the low pass filter (R5, R6, R7, pulses with a 5 V output amplitude thing now is that although the ampli- C4, C5), so they do not lead to ideally suitable for passing to the tudeof the pulses is constant during incorrect data. Drop outs also have less flipflop included in the TRS80, recording, when the tape is played effect on the circuit because, even if especially for this purpose, back the volume setting is extremely the pulse itself does not come out so LED D6 is included as a simple critical. One possible explanation is well, the transients which follow the indicator. Provided there is sufficient that one small interference pulse can main pulse will still be there, and after signal level present (in the order of a easily convert a logic 0 into a logic 1 . integration will provide sufficient few volts), the LED will light. The gain On the other hand, a drop out in the amplitude. To ensure that these pulses is set by PI . The current consumption tape can convert a logic 1 into a are not missed A1 and A2 are uski as is only a few mA which can easily be logic 0. Matters get even worse if a a two phase rectifier. This has the obtained from the supply of the clock pulse gets itself lost. In this added advantage that the phase of the TRS80. It should be noted that D6 case, a following data pulse may be signal coming from the cassette deck is can draw up to 50 mA if it is included. 2 x BC 547B Normally, the high impedance input of the 'front end' amplifier in a digital voltmeter is protected against excessive voltages by means of two diodes. One diode is connected between the input and the positive supply rail, while the other is connected between the input and the negative supply rail. In principle, this form of overvoltage protection is perfectly satisfactory. However, the diodes used would have to have a very low leakage current. The main problem here being that they are relatively difficult to obtain and also they tend to be rather expensive. Electronics enthusiasts prefer to utilise general purpose devices such as the 1 N4148 silicon diode. This does mean that with an input impedance of 1 Mfi the leakage current of the diode gives rise to an offset voltage of a few millivolts. As it is quite common nowadays to wish to measure voltages this low accu- rately, a solution had to be found. By replacing the diodes with FETs, the following result is obtained. With a reverse bias voltage of 1 5 V the diode has a leakage current of 5.2 nA, whereas the leakage current of the FET 'diode' is a mere 12 pA! This means that the input impedance of the meter can be increased to 10Mf2 with no difficulty. The circuit of the input section of a high impedance voltmeter based on the principle outlined above is shown in figure 1. Resistor R1 constitutes the 10 Mn input impedance. Transistors T1 and T2 are the protective FET 'diodes'. They can withstand a maximum current of 10 mA. The remainder of the circuit, IC1 and T3 etc., comprises a voltage follower which provides a relatively low output impedance. The operating voltage (Ug) may be anywhere between 5 V and 15 V, and the rating of the zener diode should be two volts less than the supply. Calibration of the unit is very straight- forward: preset potentiometer PI is adjusted until the voltage obtained at the output is the same as the voltage applied at the input. In principle, the input can be protected against voltages up to 1000 V, but to achieve this the input resistors will have to be suitably high- voltage types. stable amplitude low frequency oscillator Thermistors and even light bulbs have often been used in oscillator circuits to stabilise the output amplitude. The resistance of such components is dependent on temperature and there- fore on the effective voltage across the particular component. The curve of resistance versus temperature ensures that the sinewave signal generated by the oscillator is stabilised so that it is L virtually distortion-free. Due to the ^fairly slow response of thermistors and light bulbs to rapid changes in voltage, the non-linear temperature/ resistance characteristic means that there is virtually no distortion in the sinewave signal. Things are different when the thermal inertia diminishes with respect to the time period of the signal. As far as oscillators are concerned, this normally happens at frequencies below 10 Hz, or thereabouts (for instance, the vibrato signal in electronic organs). This means that in this application a different approach will have to be taken. In the circuit described here a zener diode is used to limit the voltage. A bridge circuit (comprising resistors R1 and R2 and capacitors Cl and C2) determines the frequency of the oscillator. For the circuit to oscillate, the active devices (T1/T2) must give a gain of almost exactly X3. When the amplitude of the output signal rises, elektor july/august 1982 — 7-33 the zener diode starts to conduct and reduces the gain of the amplifier stage, thereby damping the oscillation so that the sinewave tends to decay. In order to prevent the zener diode from limiting the output signal too abruptly, resistor R5 is connected in series with the zener diode. This combination is in turn connected in parallel to resistor R4. Once the voltage threshold of the zener diode is reached the impedance of the network will gradually diminish allowing the sinewave to be stabilised in a 'gentle', low-distortion manner. Even though only the positive half- cycle of the sinewave signal is in fact limited, the negative half -cycle does not last long enough to allow the amplitude to rise significantly. Potentiometer PI should be adjusted carefully to avoid severe clipping of the output signal. The negative half-cycle of the signal is extremely linear, but the positive half-cycle is slightly distorted due to the limiting. However, this will not be a problem where most applications (vibrato etc.) are concerned. The oscillator output voltage can be adjusted by means of potentiometer P2 between 0 V . . . 4 V pp . The frequency of the oscillator can be determined from the formula: 2irR1C1 ( R 1 «R2;C1 =C2) This gives a frequency of around 6 Hz with the values shown on the circuit diagram (0.01 Hz with the values shown in parentheses) . Resistors R1 and R2 should have a value of at least a few hundred kilohms. Lower values may overload the amplifier stage and with excess- ively high values the input impedance of the amplifier starts to play a role. At very low frequencies the negative half -cycle of the sinewave signal may start to clip, which will lead to considerable distortion. The DC component of the output signal may be filtered out by including a high value electrolytic capacitor in series with the output. ITT application note positive triangular waveform generator This circuit contains a small addition to the usual two opamp square/ triangular waveform generator. This is the diode included in the feedback loop of IC2 and is responsible for the rather strange behaviour of the [•oscillator. The triangular waveform output is entirely positive in contrast to a conventional circuit. Without the diode the output will be a waveform that is symmetrical about the zero axis. All this is necessary because some equipment, such as curve tracers, are unable to process a negative waveform. We start the operation of the circuit with IC2. When the output of this opamp goes negative the diode will conduct passing the negative potential to pin 3 (non-inverting input). Since the inverting input, pin 3, is grounded the output will remain negative. This output is also fed to the inverting input of 1C 1 via R 1 . The output of this opamp does not change suddenly however, but due to Cl charging, begins to rise at a linear rate. When this voltage reaches the point at which pin 3 of IC2 becomes positive, the output of this opamp will 'flip . over' and also become positive. The V inverting input of IC1 will follow ^^uit ending the charge cycle of Cl . ^ VSA-ov > S -K)| a LF^>S-i-0^ This capacitor will now discharge causing the output of IC1 to fall, again at a linear rate. The diode is now reverse biased so that when the non- inverting input (pin 3) of IC2 reaches the zero point, its output will again go negative, starting the whole procedure from the beginning. We end up with an oscillator having two outputs, one a square wave centred about the zero axis, and the other a triangular waveform above the zero point, which is exactly what we started out to get! The peak to peak voltage of the triangular waveform can be calculated from the following formula: The frequency can be found a follows: Using the formulae here, a frequency of 1 00 Hz is obtained at 5 V peak to peak (Ub = 15 V). 7-34 - elektor july/august 1982 Smoke detectors are part of any sophisticated alarm system. Most of the professionally made ones use some form of gas-sensor, ionisation- chamber, or radio-active element. The circuit described does not use any of these rather complex components but makes good use of two light detecting resistors (LDRs), and a LED. A special 1C LM1 801, allows the cir- cuit to be constructed using the mini- mum of components. It is an 1C designed specifically for use in smoke detectors, containing among other things, an internal supply zener, two reference voltage outputs, a voltage comparator, and a 500 mA output transistor with clamp diodes. The complete circuit is connected to the mains supply. Diode D1 rectifies the supply, R7 reducing the voltage to a workable level for the 1C. Capacitor C2 smoothes this, and the internal zener of the 1C stabilises it. The circuit uses a pair of balanced light detecting resistors (LDRs). By using these in a bridge arrangement, any changes in resistance due to temperature or aging effects are cancelled out. This bridge circuit is constructed by the network of R 1 , R4, the two LDRs (R12, R13), connected to one of the comparator's inputs from the junction of R4 and R13. The other inputs for the internal comparator are from the junctions of R 1 , R 1 2 and the voltage divider R2 and R3. This arrangement ensures that both LDRs are biased at the same voltage to ensure proper tracking. Physically, the LDRs should be situ- ated such that smoke particles will reflect light from the LED (D2), onto R13, causing its resistance to drop. As soon as the comparator detects this drop in voltage, the 1C triggers the thyristor Thl , causing a mains powered horn to 'sound'. PI adjusts the sensitivity of the circuit. The most difficult part of the con- struction is the placing of the LED and LDRs. Basically the LED should be positioned exactly in the middle of the two, ensuring that there is no air flow between the LED and R12. This can be easily achieved by placing a small perspex box around R12 and the LED. H Many readers may judge that the subheading is rather simple. Just take a calculator and choose a number, then press the 1/X key and the result will be displayed instantly. However, to 'treat' a d.c. voltage this way, in order to use its reciprocal value in a k measuring circuit, is something else entirely! The normal circuit design for a re- ciprocal amplifier uses four ICs. Two opamps, ICs 2 and 4, serve as input buffer and output driver respectively. Half of a dual timer IC3a forms a clock oscillator for a modulator, IC3b (the other timer). Gates N1 and N2 convert the output signal of IC3b into a 'pure' square wave signal. This cir- cuit is based on the PPM (pulse pause modulation) principle and the variable pulsewidth of the square wave signal is dependent on the DC voltage level fed to the modulator. Note, the frequency remains unchanged! For example, if the input to the circuit is a high voltage level, the pulse width of the square wave signal will be small. elektor july/august 1982 - 7-35 now but this does not imply that a voltage of 1 0 mV at the input becomes 100 V (1/10 mV) at the output. Firstly, the input of the amplifier is limited to an operating voltage of no higher than 1 0 V. Secondly, from a mathematical point of view, 1/10 mV = 100 V is not quite correct. Therefore a correction factor 'C' has to be introduced. This is about 20 • 10’V when PI is set to min- imum. Now the output voltage level will range from 2 V to 20 mV with an input voltage of 10 mV ... 1 V. The calibration procedure is very simple. Feed a voltage level of 20 mV to the input and set P2 so that exactly 20 mV can be measured between the emitter of T 1 and Ub- As already mentioned, PI determines the correc- tion factor 'C' and last but not least, P3 takes care of the offset (if necess- ary). One final point, the supply voltage must be fully stabilised. blinky ' a creature from the world of electronics Electronics can reach far beyond the frontiers of earth as a glance at the illustration shows. This electronic alien is a new arrival discovered by one of our extraterrestrial readers. In fact the entire population of the planet Kapa Sitor look like this when viewed from '•the other end of the soldering iron. Their internal construction is shown in figure 1 , which shows that, even with their 'way out' appearance, they are rather 'square'! A 555 timer-IC is used as square wave generator. The flashing (blinking) LED is not connected to the output (pin 3) as many readers might v/ have expected, but to the discharge output. The reason for this peculiarity is the fact that the normal output is used to drive the other 'Blinkies'. Since they are symbiotic it is possible to obtain a complete Blinky family, living in complete harmony. Figure 2 shows how the Blinky family group must be made up. All the components are mounted carefully together as shown in the illustration. The legs have to be connected across the 9 V battery connector. One 'hand' must be bent as a hook and the other as an eye. The same applies to the connections A and D (do not forget the insulation sleeving). Finally interconnect them as illustrated in figure 2 and the electrical connections will be made automati- cally. If their body is 'deformed' in any way, they will not be able to carry out their allotted task in life: that of blinking out goodwill to the nations of the universe with their heads! And there is a lot to be said for that . . . 7-36 - elektor july /august 1982 making life difficult for burglars Most alarm systems can be divided into two main categories. They are normally activated by closing, or inter- rupting a circuit loop. One of these basic principles is used irrespective of the electronic method adopted, (micro-wave, infra-red, photo-cells, contacts etc.). Today's burglar is not the simple- minded individual normally portrayed in comic strip cartoons. The professional certainly keeps up to date with the latest technological advances in alarm systems, and keeping him out is going to be difficult. Even part- timers unfortunately know something about electronics and alarm systems. Anyway the point is, that an average burglar can easily and quickly deter- . mine what principle the system uses \ and at least try to deactivate it. This is sometimes made easier for the thief, ^^^because the hiding of the connection The circuit described here should pose a more difficult problem. It is intended to protect a single door, window or item of equipment - a TV set, for instance. A resistor, R2, is mounted inside the item that is to be protected and two leads are brought out (via break contacts or even an audio plug) to the alarm circuit proper. Should the burglar locate the wiring and try to either cut of bridge it, the alarm will activate. Resistor R2 and the connections to capacitor Cl form a make or break loop. If the loop is interrupted or the two connection wires bridged (shorting out the hidden switch) the alarm will sound. The circuit uses a window discrimi- TCA965. The operation of the alarm is rather simple. When pin 8 receives a higher voltage than pin 6, or a lower voltage than pin 7, the 1C will drive T1.T1 conducts and activates the relay Re. A high frequency mains driven horn connected via the relay, should be enough to panic the thief. elektor july/august 1982 — 7-37 It is often said that two heads are better than one but this numerical advantage applied to hands could also be a great asset, especially when using probes to test a complex printed circuit board. It is an absolute certainty that the test probe that you have just painstakingly con- nected will flip itself off at the instant that the power is switched on. Further, it is a known fact that it will land with unerring accuracy on the most 'sensitive' part of the circuit - and discharge the smoothing capacitor across the input of the circuit! How well we know the problem! The title of this circuit could well have been 'Frayed temper adjuster' since it is capable of just that. It the use of both hands to on and hold the probes while power to the circuit is applied automatically, after a short delay. It even tells you (visually) when this is about to happen. An astable multivibrator with a frequency of about 2 Hz is formed by gate N1. Its output is buffered by two further gates, N2 and N3, in parallel in order to provide enough current drive for the input of the decade counter 1C 1 . The counter is reset on power up by the C2/R2 combination before providing an output to the second 1C, a binary-to-decimal decoder. The first of the ten LEDs connected to the output of this 1C will light two seconds after power is applied to the circuit. It will be followed at 2 second intervals by the other LEDs until DIO lights after a total of 20 seconds. As can be seen from the circuit diagram, the final output at pin 1 1 is buffered by the gates N4 . . . N6 in parallel. These provide sufficient base drive current to allow transistor T 1 to activate the relay at the same time that DIO lights. Power to the circuit under test is then provided via the relay contacts (not shown here) and will remain until the delay circuit is switched off. This 'latch' is provided by the link between the N4 . . . N6 outputs and pins 6 and 7 of 1C 1. The time periods can be varied by altering the value of resistor R 1 , a larger value will lengthen the time. A simple stabilised supply consisting of a 7805 regulator can be used to power the delay circuit. However, the 'delay on' switch should be placed between the regulator and the delay circuit to ensure that the initial reset works reliably. N The dynamic RAM card in the April 1982 issue of Elektor has found many friends among Junior Computer owners. However SC/MP owners will also be pleased to discover that the same RAM card can be used by them. As a reward to all the SC/MP owners (for staying with us for so long), here are the modifications required to • adapt the dynamic RAM card to their systems. 16K in 8 ICs on a single card is worth quite a lot, and SC/MP users should find whole new possibilities for their systems. Unfortunately the basic version was not suitable for the SC/MP system. This is due to the fact that the SC/MP system would interrupt the refresh instructions resulting in data being lost. The simple interface consists of a single 1C, two resistors and two capacitors. Furthermore a set of wire links and connections (as shown in the table) must also be made. The circuit consists of a retriggerable monoflop MMV1 , with a pulse length of approximately 1 0 ps. As long as the NADS pulses keep on coming, the output IQ is always at logic 1 . This readies the second monoflop via N2 and N43 on the dynamic RAM card. If within 10 ps no further NADS pulses occur, IQ becomes logic 0, and via N43 the second monoflop is triggered. 2Q provides a 300 ns pulse as a refresh instruction. The refresh signal also appears at the 20 output of MMV2 and retriggers MMV1 . Output IQ will then become logic 1 again for 10 ps. This means in effect that as long as NADS pulses are not occuring the dynamic RAM is refreshed every 10 ps x 1 28 = 1 .28 ms. The circuit can also be used for other systems with manual reset. Table Wire links on the RAM card 1 -1 ', 2-2', A-B. J2, J3, J5, J6, J9 IC22 is omitted Connections on RAM card 5' to +5 V. 3' to C Connections from interface to RAM card Pin 13/MMV1 to 4', Pin 9/MMV2 to Pin12/ N43, Pin 1 /MMV1 to J4-A-Pin 1 2/N1 , Pin 5/MMV2 to J4J3-J5, Pins 3, 1 1 , 10, 16, inn R1 and R2 to +5 V, Pin 8 M 1982 economical crystal time base a 50 Hz 'bench mark' This time base circuit is built using normal readily available CMOS ICs and a cheap crystal. The operation of this circuit is practically identical to that described in the 'Crystal stroboscope’ article in the April 1981 issue of Elektor. The difference between the two projects is that whereas the first one only produced an output of 50 Hz this new circuit gives the constructor the possibility of 50 Hz, 100 Hz or 200 Hz. The 50 Hz reference frequency is an ideal time base for the construction or calibration of electronic clocks, frequency meters and so on. Because of the flexible supply voltage require- ment, it is also a good basis from which to build a digital clock for the IC1 contains an oscillator and a 2 14 divider. Providing the oscillator Capacitors: Cl - 22 p C2- 2. . . 22 p tr C3 “ 10 p/16 V loop is correctly calibrated using C2, the output at pin 3 (Q14) will produce a 200 Hz square wave. With the help of the two flip-flops in IC2 FET field strength meter . . . this square wave voltage is then divided by two and then by four resulting in two further outputs of 100 Hz and 50 Hz, the latter from pin 1 . Readers who have a frequency meter can calibrate the circuit by simply connecting the meter to pin 7 of I Cl (Q4) and adjusting C2 until a reading of 204.800 Hz is indicated. As a matter of interest, anyone with- out a frequency meter should not despair since setting trimmer C2 to about midway will provide sufficient accuracy for most applications. The 100 Hz output is useful for the construction of digital counters. For this purpose we suggest that a 1 : 10 divider (like the 4518) is connected to the 100 Hz output pin. The power supply requirements are: from 5 ... 15 V and 0.5 ... 2.5 mA. . . . with RF amplification A field strength meter is necessary when checking the power output and aerial of transmitters. With this circuit it is possible to measure the energy V radiated by the aerial. This is useful not only for hams, but also CB w enthusiasts and radio control modellers. For various reasons this type of meter must be very sensitive. First of all, there should be a distance of as many wave lengths as possible between the measuring instrument and the trans- mitter. Secondly, other people will not be jumping in the air for joy when you are calibrating the aerial with a strong carrier signal. A weak signal will suffice when using a sensitive field strength meter. Thirdly, most trans- mitters only have a weak output power (for example, 500 mW). July /august 1982 - These are three of the main reasons why our field strength meter is equipped with an RF amplifier stage consisting of a Dual gate MOS-FET, T1 . The amplification factor is set with PI . Switch S2 enables one of the three ranges to be selected: 480 kHz... 2.4 MHz (LI); 2.4... 12 MHz (L2) and 12 ... 40 MHz (L3). A rod of approxi- mately 30 cm will be enough to serve as aerial. As with all RF circuits, care during construction is necessary! 4 People with a passion for hifi equipment and active speaker units are bound to have sought ways in which to switch on the output units via the pre amp. Funnily enough, many hifi manufacturers seem to regard automatic switch mechanisms as an unnecessary luxury. Automatic switches are, however, extremely useful and avoid having to lay yards and yards of leads throughout the house. Instead, a single or several 'remote' active units may be switched on by way of the original AF lead. As the switch mechanism is always 'listening in' anyway, it is also able to detect the prolonged absence of a signal, in which case it will simply switch off the output unit. Relatively few components are required for the circuit. Basically, it involves a double opamp, a timer 1C and a relay to switch the mains voltage. Opamp A1 is connected as a non-inverting AC amplifier. Note that its negative input is connected to the positive supply voltage by way of R3/C2. This prevents the relay from operating as soon as the supply voltage is switched on. The gain of the opamp is high enough to prevent even low voltages from de-energising the relay. The second opamp, A2, is a comparator. PI sets the switching tnreshold for AF signals at roughly , 2.5 mVrms- V Should the output voltage of A1 ^kexceed the threshold value of the' W. Wehl comparator due to the arrival of The supply voltage for the circuit an AF signal, the comparator output is derived from the mains by way will go high. As a result, capacitor of a 12 V or 1 5 V voltage regulator C3 is charged by way of diode D1 and a small transformer together and resistor R7. When the charge with a rectifier and smoothing level of the capacitor reaches about capacitor. 2/3 of the operational voltage, the timer 1C output will go low and the relay will be pulled up. The relay Warning! The relay contacts are contacts connect the active unit to connected to the mains, so take the mains. If no more AF signals care when constructing the are applied, C3 will discharge via circuit. R8/P2 within 1 ... 5 minute(s). The relay will then drop out. jgust 1982 Wn| mini high performance\ wlol voltage regulator . . with only 1 V drop One thing is common to virtually any voltage regulator; the input voltage level must be several volts higher than the expected output. Admittedly those fewer volts at the output are very nicely regulated. However, if for some reason there are very few volts at the input to start with, then there is a limitation in the output voltage range (far less volts to throw awayl). In this case it is not possible to use a normal 1C voltage regulator and we have to resort to a discrete design. The circuit shown here will operate with a 6 V input and provide a regulated 5 V output, which is ideal for battery powered equipment. With a little study the 'trick' in the circuit will be apparent. The load is connected to the collector of the series transistor. This means that this transistor can be switched hard on into saturation, so that the voltage between emitter and collector is only the very small saturation voltage. This voltage level depends of course on current and transistor type. In this case at a maxi- mum current of 0.5 A the voltage loss will be only 0.2 V. Add to this the voltage drop across R6, required for current limiting. At approximately 0.5 V across R6, T3 starts to conduct and limits the output current. LED D1 has two purposes in life; as an indicator and as a voltage reference diode which sets , a level of 1 .5 V to 1 .6 V at the emitter of T1 . The base drive current for this transistor is derived from the voltage divider consisting of R4, PI and R5. Depending on the difference between the reference and output voltage levels, T1 is more or less conducting. The same then applies to T2 which will supply more or less base drive to T4. Capacitor Cl is included to filter the output stage. Instead of the BD 438 other well- known types can be used like the BD 136, BD 138 and BD 140 for instance. However, these transistors do have a slightly higher saturation voltage. It must be noted that since D1 acts as a reference source, it must be a red LED. Other colours have different parameters. M The analogue brother of this 1C is our old friend, the 555. The digital version here, the LS 7210, is less well-known. It can be used to set delay times between approximately 1 1 ^is and 42 minutes. The 1C contains an oscillator of which the frequency determining elements are connected externally ( R 1 and Cl). This then provides the frequencies as shown in table 1 . The 1C is programmed for internal oscillator operation by con- necting pin 4 to 0 V. The delay time T is derived from the formula: where 'f' stands for frequency according to table 1, 'N' is the multi- plication factor as determined by pins 8 . . . 12. These pins have the following values: pin 12 = 1, pin 1 1 = 2, pin 10 = 4, pin 9 = 8 and pin 8=16. For example, if N is to be 25 then pins 8, 9 and 1 2 must be logic '0' (0 V). In this case, with the oscillator frequency set to 0.013 Hz, the total delay time will be 34 minutes. As shown in the circuit diagram, the 1C is used as a retriggerable monoflop. The output becomes logic '1' at the same time that a negative going edge T = (1 + 1.023 -N)/f 1982 — 7-4 elektor july/august arrives at the trigger input, pin 3. The output level reverts to logic '0' at the end of the preset delay time period providing no further trigger pulses arrive at the input. Should this happen, the preset delay period will be initiated again, but the output will remain high. A positive input edge has no effect on the timing. The result of this is that, in principle, any length of time period can be realised by cascading 2 or more ICs in series. The output of the 1C consists of a FET with open drain connection. There- fore, to obtain current switching between '0' and T, a pull-down resistor, R2, is necessary. However, if the output is to be used as a current source this resistor can be omitted. LSI application M The performance of varicaps is im- proved when the voltage across them is increased. Besides better inter- modulation rejection, a 30 V circuit has a considerably higher Q than a 9 V I version for the same capacitance variation. However, with battery- powered circuits, this high voltage will cause a problem, since deriving a tuning voltage of 30 V from a low 'supply voltage can only be realised with the aid of a converter. The circuit diagram shows the design for a converter especially constructed for this purpose. The LM 10C, from National Semiconductor, which contains two opamps and an internal reference source is ideal for this particular application. The oscillator is constructed around a dual-gate MOSFET (type BF 900) and functions at a supply voltage as low as I.5.V. The output voltage level of the converter is controlled via the supply voltage of the oscillator. Unlike most converters, this one does not have to be switched, so that there will be no distortion. The oscillator frequency is approximately 28 kHz. An AFC voltage can be connected to one of the opamp inputs via a series resistor; which of the two inputs depends on the polarity of the AFC voltage. With the values indicated in the circuit diagram, the output voltage can be varied between 1 and 30 V by means of the 220 k potentiometer. The supply voltage can range from 3 to 16 V. N st 1982 N1 ... N4 = IC1 = 4011 B = to top octave generator The limited five octave range of many electronic piano's and organs can be extended by one octave lower with the aid of the circuit here. It is con- nected between the main oscillator (input point A) and the highest octave generator (output point Bl. A monoflop is constructed with N1 , N2, Cl, PI and R4. Its time period is set by PI so that the monoflop divides the frequency of the main oscillator by two, switch SI provides the ability to switch between the original tone range and the extra lowered tone range. The diodes D1 and D2 protect the input against high level and negative input signals. The value of Cl depends on the frequency of the main oscillator, but can be found quite easily after some experimenting; the frequency of the piano or organ will suddenly be lowered by one octave when turning PI . If this does not occur, the value of Cl must be increased. When the correct value is found, the correct position for PI is that when the frequency is lowered, plus a little extra 'tweak' to retain stability. This completes the 'calibration procedure'. A final note (!), the input voltage at point A must be at least 60% of'the supply voltage. After last year's welcome drop in prices of good quality EPROMs, computer enthusiasts have a great incentive for taking on more ambitious programming projects. Although normal operation calls for a 5 V supply voltage, 25 V is needed to program a 2716. In some types, the 25 V programming voltage need not be switched off while the operator checks freshly stored data. On the other hand, there are types for which the voltage has to be switched from 5 V to 25 V continuously. It therefore follows that a suitable EPROM power supply has to meet certain requirements: it needs to be straightforward, fast (often the speed is specified by the manufacturer as being, say, between 0.5 and 2 /js), accurate (no danger of overshoot or undershoot) and short proof. The well-proven 723 voltage controller 1C fits the bill perfectly. As the circuit diagram shows, the 723 is at the heart of an ordinary 5 V power supply. Preset PI limits the reference voltage (pin 6) to 5 V and feeds the signal to the non-inverting input. When transistor T1 stopsconducting, the whole output voltage is fed to the inverting input (pin 4) and 5 V will therefore be available at the output. Resistor R7 limits the current. So far so good, but what about the 25 V we said we needed? This is obtained by changing the feedback loop to pin 4. The output voltage is increased by adding a voltage divider to this section in the circuit. T 1 activates the voltage divider. As soon as the base of the transistor is driven. elektor july/august 1982 — 7-43 the 723 produces the 25 V voltage. In order to obtain different voltage levels the values of R5, R6 and P2, will have to be changed. Calibrate the circuit as follows: use PI to set the output voltage to 5 V without driving T 1 . Then drive T 1 by applying 5 V to R3 and set the output voltage to 25 V with P2. That's all there is to it! The upper trace in the photograph represents the signal controlling T1 (between 0 and 5 V) and the lower trace shows the output signal. The 723 is especially fast because pin 13, the frequency compensation input, is not used here. Normally speaking, a grounded capacitor is included at this point to smooth the signal edges. Note that it takes the output signal another 2 jis to go low again, once the control signal has gone low. This is because it takes transistor T1 quite a while to stop conducting. In applications where the time factor is highly critical, this may be a problem, in which case it is best to replace T1 by a CMOS switch (such as the 4066) ora V-FET (such as the BS 170), omitting R3 and R4. Alternatively, a proper switching transistor, the BSX 20, also provides excellent results. M A remote control system having 20 channels with analogue functions can only be realised with the use of special ICs. Any other method would require an enormous quantity of components. However, it is all very easy, thanks to Plessey who produce a range of ICs designed specifically for this purpose. Our designers selected three of these for the remote control system here. It is capable of transmitting no less than 32 com- mands when used in conjuction with the receiver and associated circuits. The transmitter basically consists of a keyboard decoder 1C, an output and transducer stage and a small battery. In much the same manner as a pocket calculator, the commands ordered by the keyboard are fed into a matrix. This is arranged in 4 columns and 8 rows enabling 32 keys to be used (32 junctions or cross-points). It must be pointed out here that only one key can be operated at a time or the 1C will simply ignore the entry. The key command (one key pressed) is converted into a corresponding 5 bit binary code. No detailed descrip- tion of codes or their allocation to the keys or matrix will be given here but is available from the data source mentioned at the end of the article. The 5 bit code is transmitted by means of the infra-red transducer diodes D1 and D2. The code is in the form of a pulse sequence consisting of 6 equal pulses interspaced by 5 spaces or pauses. The binary data is contained in the pauses, a long pause for a logic '0' and a short pause for logic '1 '. This is termed 'pulse-pause' modulation (PPM). The length of the pulses and pauses can be calibrated with the aid of the preset potentiometer PI . The relation- ship between a logic '0' and a logic '1 ' ideally should be 1 .5 : 1 . The pulse width is approximately 3 ms while the interval between two command words will be about 54 ms. The trans- mitter will radiate an infra-red light signal when the output at pin 3 of IC1 is 'high'. This will be a 1 5 ns pulse which can produce a current of up to 8 amps through T2 and the diodes. The 1C also contains an electronic stand-by switch which will reduce the quiescient current consumption of the 1C down to a miniscule 6 pA when not in use, that is, between key oper- ations. Reference: Remote Control Data, Plessey Semiconductors. P-44 - elektor july /august In the Elektor December 79 issue the pros and cons of charging NiCads rapidly were discussed at length and two suitable circuits were put forward. The circuit here elaborates on the 'old' idea in order to produce something new . . . The graph in figure 1 shows what happens during a (fast) NiCad charge cycle. At first, the voltage rises very quickly from its initial 0% charge to attain as much as 1.42 V with a 25% charge level. After this point, the voltage will tend to rise more gradually. Just before the fully charged level is reached, the voltage surprisingly surges once more. In the first of the two fast charger circuits published in the December 79 issue, the rise in voltage was used as a parameter for monitoring the charge cycle. In the second circuit, however, a similar system was used to interrupt the charge cycle when the battery was 'overcharged' by about 20%. The manufacturer assures us that this cannot damage the battery. As figure 1 shows, the gas produced when the battery is about 75% charged, causes a dramatic increase in the pressure and temperature inside the battery. By using the temperature curve relative to the charge, a simple procedure involving two special temperature sensor ICs serves to switch off the supply current when the temperature of the battery has risen 2 by 5°C. As can be seen in the graph, this is fairly conservative: an almost 'dead' battery will be charged to 50%, and even an almost 'full' battery will remain within the 20% overcharge margin. Figure 2 shows the circuit diagram. The differential switch is similar to the one described in last year's Summer Circuits (no. 50). The output of the comparator opamp IC1 goes low whenever the voltage at its negative input is equal to that at its positive input. PI sets the voltage level at the positive input so that it is 50 mV above that of the negative input. When the operational voltage is switched on (don't connect up the battery yet!) sensors D1 and D2 must be given enough time to reach the same temperature. Depending on the temperature of D2, the voltage at the negative input will increase by 10 mV per degree °C. As D2 is mounted on top of the NiCad (preferably tightly strapped by a rubber band), the rise in battery temperature will automatically switch off the charge current. A different voltage may of course also be set at the positive input. As illustrated in figure 1, the battery has only reached 50% of its charge level by the time the temperature has risen by 5°C, if it was initially completely dis- charged. However, there is a reason for this. The graph shown here can not be taken as 'gospel' for every battery and for all possible charge currents - and it is better to err on the safe side! There is an alternative, of course: you can progressively increase the temperature difference that the circuit will tolerate before cutting off, until your particular type of NiCad cell proves to be fully charged. The advan- tage is obvious, but the risk should be equally clear . . . Fortunately, temperature rises quite steeply once the cell is fully charged, so the chance of getting too far off the mark is not so high. According to the graph, 12°C (120 mV) is still quite safe. The circuit works as follows. After closing SI and operating S2, the charger starts to pump about 1 A into the battery. The current is provided by the variable voltage regulator 1C, the LM 31 7T, which serves as a constant current source. If the comparator out- put is high, D3 and D4 will be cut off. As a result, the internal reference volt- age of IC2, 1 .25 V, will be across R8, enabling about 1 A current to flow Letters from readers together with the numerous comments expressed during the last Breadboard exhibition showed that there was a large demand for a low cost tape playback pre-amp. Readers either wanted to improve the quality of their existing low cost re- corder, or to build an auxiliary deck using one of the easily available drive mechanisms. In both cases the extra deck would be very useful especially for tape copying. The circuit is constructed using a new, low cost 1C from National Semiconductor, which was designed specifically for tape playback appli- cations. The 1C is very interesting due to its low noise, wide voltage supply range, and low power consumption properties. It also requires very few external components in order to con- struct a complete circuit. The distor- tion factor is less than 0.1 % at frequencies ranging from 20 Hz to 20 kHz, at an output of 1 Vrms. The printed circuit is quite small and can be mounted easily onto any cassette chassis. A power supply delivering approximately 10 mA, at a voltage of anything between 10 and [)0— T — r- 78L12 - T — @ 12v ,6v .^TTu-ITt 16 V is sufficient. The circuit is compatible with the noise reduction circuit (DNR), published in our March 1982 issue. The LM 1897 is a dual gain pre-amp for any application requiring optimum noise performance. It combines the qualities of low noise, high gain, with good power supply rejection (low Hum) and transient free power up! No 'power up' transients are achieved primarily because no input coupling capacitors are used. This eliminates the 'click' or 'pop' from being recorded onto the tape during power supply cycling in tape playback applications. The omission of these capacitors also allows a wide gain bandwidth with unlimited bass response. The external components in the feedback loops, determine the gain and form an equalisation circuit. Using the values shown in the diagram (figure 1), a gain factor of 200 is achieved at a frequency of 1 kHz, correspond- ing to an output level of 100 mV rms . Most available tape heads should give results of this kind. The equal- isation time constants are 3180 and 120 /is for ordinary low noise cassettes. For all other types of tape, such as ferro chrome and chromium dioxide, the defined constants are 3180 and 70 ps, in which case the two R4 resistors are replaced by 33 kO ones. Constructors not wishing to use the muting option, can leave out switch SI and the two R7 resistors. Screened two or four way cable should be used to connect the circuit to the tape heads. The choice is up to the constructor, but please keep in mind that if two way cable is used the screening sleeve is to be connected to the ground of the printed circuit board. A good ground connection between the printed circuit board and the drive chassis is also essential! An unstabilised, filtered D.C. voltage of between 10 and 16 V will be suf- ficient for the circuit because of the high power supply rejection (low elektor july/august 1982 - 7-47 hum), of the 1C. Batteries can also be used successfully. A voltage regulator such as the 78L1 2 is required only when the available supply is unfiltered or likely to be 'noisy'. The output of the pre-amp has not been decoupled since virtually every power amplifier contains some type of input coupling capacitor. Constructors who are in doubt about this fact can insert capacitors C5 and C5', as shown in the circuit diagram. The pre-amp has a low output impedance. This should not present any problems as the input impedance of most amplifiers and other 'HI FI' equipment is around 1 k£2. This voltage controlled oscillator (VCO) is capable of providing a triangular as well as a squarewave output signal. As with any other VCO, the frequency of the output signal depends on the level of the control voltage (U c ). Remarkably, this design features a wide control voltage range; between 0 V and the positive supply voltage. The power supply voltage can be anywhere in the region of +3 V to +25 V. However, care should be taken when using low voltage supplies that 1 the maximum output level is at least 1 .5 V below that of the supply. The circuit is based on the 'integrator - comparator' principle. Capacitor Cl is part of the integrator (constructed around opamp A1) and is charged by a constant current level determined by the instantaneous level of the control voltage. Consequently, the output of A1 will fall linearly. The output of the comparator (constructed around A2) will change state and transistor T1 will start to conduct when the lower switching threshold of the comparator is reached. Capacitor Cl is now discharged causing the output of A1 to rise (again, the voltage rise will be linear). This process will be repeated when the output of A1 reaches the upper switching threshold of the comparator and T 1 is turned off. The duty cycle of the output signal will be 50% when the values of R2 and R3 are the same and when the value of R1 is twice that of R4 (R2 = R3 and R 1 = 2 x R4). The relationship of the values of resistors R9 and RIO determines the DC level of the triangular output signal. With the values indicated in the circuit diagram, the DC level will be half the supply voltage. The peak-to-peak output level |Vp p )«e t |u.ito ps 55 ra xUb. The characteristics of the VCO with two (common) supply voltages are shown in figure 2. The maximum frequency (when U c = Ub) supplied by the circuit can be increased or decreased by selecting a lower or higher value, respectively, for capacitor Cl . Due to the slew rate of the opamp, the steepness of the squarewave signal will fall off at higher frequencies. 2 H graphic oscillator 'And we don't mean graphic equaliser'. An oscillator operating along similar lines to an equaliser. I n the case of the latter, a set of slide potentiometers adjust the frequency response and the level can be directly deduced from the position of the levers. Here slide potentiometers are used as well, only now for the purpose of setting the waveform on the screen. analogue monoflop , To understand the object of the graphic oscillator the circuit diagram should be looked at 'back to front'. P2 ... PI 7 sets the DC voltage in the 0 ... 5 V range. Electronic switches ESI ... ESI 6 feed the voltages to the output of the circuit. Normally speaking, the article should end here, were it not that the circuit has an additional interesting feature to offer . . . When an oscilloscope is connected to the output, a waveform appears on the screen that can be adjusted to contain up to 16 steps. Fortunately, this does not have to be done manually for the remaining components produce a constantly repeated switch cycle. The counter IC8 provides a 'bit' pattern at its outputs to the rhythm of the pulses generated by IC9. The bit pattern, decimal numbers 0 . . . 1 5 in binary, drives the multiplexer IC7, so that its output goes 'low' whenever the input data is addressed to the output concerned. For example, where A = high, B = low, C = high and D = low, output 5 = low. Since a logic one inhibits the electronic switches, 1 6 inverters are required to make sure the right DC level reaches the output. By adjusting PI and Cl, the clock frequency can be set to a very wide range. Where Cl = 1 n, theoretically: f = 123 ... 710 kHz and where Cl = 1 0 ft, f = 123. . . 710 Hz. . using an opamp as a comparator Monoflops are automatically associated with digital circuits, but there is no reason why they should not be used for analogue purposes. Obviously, the opamp involved will not be used as an amplifier, but as a . comparator. The 741 is implemented V in both of the circuits shown here, although, as a matter of fact. practically any type of amplifier will suit this application. Modern 1C technology makes life much easier for the designer in that four opamps can be incorporated in a single tiny package. More often than not, however, one of the opamps is not required, which is a bit of a waste, and what's more, an additional digital chip is needed to effect a specific time delay. But the latter can be omitted by combining an opamp with a monoflop. Operation is quite straightforward. The inverting input is set at a fixed voltage level, (slightly more than half the supply voltage). The non-inverting input is grounded by R5and PI. The elektor july/august 1982 - 7-49 output is therefore also at ground potential and diode D1 will not conduct. When a positive pulse is applied at the input, it is fed to the non-inverting input by capacitor Cl. For a short time this becomes higher than the inverting input. Asa result, the output of the opamp will be connected to the positive supply voltage. Diode D1 will now conduct and make sure that point A remains positive even when the input signal is no longer applied. This situation will not change until capacitor Cl is charged by way of R5 and PI and the voltage at pin 3 is lower again than that at pin 2. The opamp will then 'flip' over, its output being grounded In principle, the same procedure applies to the negative response circuit. As can be seen in the pulse diagrams, the input signal should be either longer or shorter than the required output signal. The resultant mono time is around 0.5 (R5 + PI ) - Cl. PI sets the exact value, which is determined, to a certain extent, by the saturation of the opamp output, and so can only be calculated approximately. Just make sure that the input signal is always slightly smaller than the variation in amplitude at pin 6, because the signals might affect each other, especially if the input and output pulses have the same duration. the simplest PDM amplifier 'pulse duration modulation The term PDM merely stands for pulse amplifier connected to an integrator duration modulation. A PDM amplifier which together convert the amplified consists of a pulse duration modulator, which converts an analogue audio signal into a digital PDM signal, and an PDM signal back into an analogue signal. This particular circuit is prob- ably the most straightforward PDM amplifier in the world. In the wake of digital audio technology 'break- throughs', PDM devices (or digital amplifiers) are rapidly gaining popu- larity. Some Japanese manufacturers are even including PDM technology ll-o* — poll in their current ranges of stereo amplifiers equipment. The circuit described here is based on the fact that the transmittance curve of a buffered (B version) 4066 CMOS switch is extremely steep. Asa result, the device can be used to reliably obtain a high gain factor. The circuit shown to the right of the figure represents the analogue equivalent of the PDM circuit. This corresponds to an inverting analogue amplifier, which unfortunately has a rather high distortion factor thereby making it totally unsuitable for hifi purposes. The gain of the circuit using the component values shown is 10. A gain of 1 00 can be achieved if the values of the components marked with an asterisk are altered to 1 Mf2 and 1 n. iuly /august 1982 Class A amplifiers are well-known in the audio world for their low distor- tion figures and big heat radiation. Manufacturers have always tried to design an amplifier having the advan- tages of class A without the drawback (heat). During the last few years they came up with several solutions. One of them was found by the Japanese manufacturer Matsushita, who developed an ingenious method that makes a 350 W class A amplifier possible without the 'heat problems'. The amplifier described here follows the same principle, but with one major modification: The output power is reduced considerably, in order to sim- plify the construction. After all this is a 'summer circuit' not an 'annual circuit'. The circuit diagram shows a normal power amplifier at the left-hand side with an output stage consisting of a TDA 1034. The final stage (T1 . . . T4) is set in class A mode. The dissi- pation remains low, because the final stage is fed by ± 5 V. However, this supply voltage is much too low for the amplifier to deliver enough power. For this reason, the zero of the sym- metrical 5 V supply is connected to the output of a second, straightfor- ward power amplifier consisting of IC2 and T5 . . . T8. This amplifier is in class B mode and is fed with the same input signal as the first amplifier. The main difference is the fact that it operates with a higher supply voltage: ± 18 V. The amplification factor of the second amplifier equals that of the first. The loudspeaker is connected between the output of the first amplifier and the zero of the 18 V supply. The zero of the 5 V supply is connected to the output of the second amplifier. Any input signal will now drive both amplifiers simultaneously. This means that a voltage is 'added' to the zero of the 5 V supply by the output of the second amplifier, which has the correct value and polarity for the first output stage to deliver the desired power to the loudspeaker. During the positive swing of the signal waveform, the collector of T3 is at the necessary output voltage plus 5 V. When it swings negative, the collector of T4 is at the required negative output voltage minus 5 V. In this way the amplifier operates in class A mode, but the dissipation remains nearly the same as that of a class B amplifier, as the supply voltage 'runs along' with the input signal. When using this method it is a must that the input amplifier (IC1) can be driven to the high supply voltage. Therefore IC1 is supplied with ± 18 V. Furthermore, the 5 V supply must deliver a current that at least equals the peak current flowing through the loudspeaker. The power supplied by this amplifier is approximately 1 5 W into 8 Q (this is class A). When constructing the circuit, make sure that the 5 V supply is completely separated from the 18 V supply. Use a mains transformer with two com- pletely separated secondary windings with a centre tap, or even better, use two transformers. Only the zero of the 18 V supply serves as ground for the circuit and the loudspeaker. omnivore ' not fussy about voltages ^ Ordinary LEDs have a rather monot- onous diet: they will only 'swallow' DC current with the right polarity, in which case a series resistor cuts down the current appetite to a moderate 10 ... 30 mA. This type of provision has a drawback in that the value of the series resistor must be calculated for each separate supply voltage, and that fluctuations in the supply signal can only be handled within a limited range. Substituting a FET for the series resistor affords a number of advantages. When the gate and source are linked, the transistor forms a current source without the need for any additional components. In the type used here, the BF 256C, the con- stant current is between roughly 1 1 and 1 5 mA, with a wide supply range of 5 ... 30 V. A universal silicon diode (DUS), such as the well known EX(N)OR opamp 1N4148, will provide polarity protec- tion when connected in series with / the LED. As a result, the 'Omnivore' LED can be driven with AC voltages in the 5... 20 V (=7... 30 Vlas well. At the normal 50 Hz mains frequency, the LED will barely flicker at all, except that its brightness will be a little dulled due to the half-wave rectification, compared to that at an equivalent DC voltage level. an anlogue digital gate Nowadays, digital techniques are finding their way into more and more analogue circuits. Fortunately, this does not always call for the use of special integrated circuits, as it is quite common to see opamps being used to provide the logic functions NOT, AND, NAND, OR and NOR. However, this does not (normally) apply to the logic functions EXOR and EXNOR. Nevertheless, the latter can be obtained by using LM 324 or LM 358 type opamps. These opamps have the advantage that their outputs can be driven to 0 volts without the need for a negative supply voltage. As can be seen from the circuit diagram, when both inputs A and B are grounded (= logic zero) point a will be low. As a result, resistor R5 will have no effect on the state of the inverting input of the opamp. Resistor R6, however, does affect the non- inverting input via diode D2. This causes the voltage at the non-inverting k input of the opamp to be lower than that at the inverting input, leading a low level at the output. If the two inputs A and B are taken high (= supply voltage), point b will also go high via diodes D5 and D6. Thus, resistor R5 now affects the state of the opamp instead of R6. This causes the voltage at the inverting input to be greater than that at the non- inverting input, therefore the output of the opamp is once again low. If one of the inputs is held high and the other low, point a will go low and point b will go high. This means that now the voltage level at the non- inverting input will be greater than that at the inverting input, resulting in a high voltage level at the output of the opamp. In other words, a genuine EXOR gate! The EXNOR function can be obtained very easily indeed. Simply swap around the inverting and the non- inverting input connections. Now the output of the opamp will go low whenever the two input levels are different and will go high when the input levels are the same. july/august 1982 Several relatively popular broadcasting stations can, in some areas, only be received on MW or LW. The repro- duced sound quality of these trans- missions is normally quite low. Nothing like HI-FI is normally possible because of the limited bandwidth of transmissions. However a greatly im- proved sound quality is possible, obtained quite easily by using just a few widely available components. The improvement is so remarkable that it can be noticed distinctly. The out- standing feature of this receiver is its unconventional concept. The tuning stage of the receiver also serves as an active aerial, which can be favourably placed in order to get the best possible reception. Furthermore it is com- pletely separated from the rest of the receiver, that is from the demodulator supplying the AF output. This part can be inserted into a separate housing, and placed next to an ampli- fier or the HI-FI equipment. The inter- connection between the two parts should be made using standard coaxial cable. This cable feeds the RF signals and the tuning voltage (which is the operational voltage of the aerial) to the modulator. The plastic aerial housing contains an aligned input circuit, consisting of a ferrite rod (L2), and double varicap. The aerial signal is coupled to the tuning stage by an emitter/follower transistor (T1), ensuring that a high impedance output signal is fed to the modulator. This improves the selectivity. T2 together with its surrounding components forms a current source for T1 . The received signal is not amplified whatsoever in the active aerial stage, but in part of the TBA120 1C which forms part of the modulator. L2 serves as an emitter decoupler for T1 . L3 decouples the supply and tuning voltage thereby short proofing the RF output of the active aerial. L4 effectively doing the same for the demodulator. PI can either be a trim- ming potentiometer allowing preset tuning of a particular station or a multi-turn (helical) type for normal variable tuning. TBA120 1C is the amplifier and quasi-synchronous- demodulator for the signal fed from the active aerial. Apart from the unusual method used for modulation, the receiver follows the standard 'straight-through' principles having a good $ignal-to-noise ratio. Unfortunately the main dis advantages of this design is that it suffers from bad selectivity and low sensitivity. Consequently the constructor should not expect the receiver to work miracles, especially during the evening hours or when trying to tune in to distant stations. However for most relatively local stations it will per- form well. Potentiometer P2 sets the gain of T3, thereby allowing the output level to be matched to the input requirements of any amplifier. Should the con- structor desire to improve the selec- tivity then we suggest inserting a positive feedback loop with its as- sociated components as shown by the dotted lines (see the circuit diagram). Except for LI , standard chokes can be used for the coils. LI consists of 250 windings of 0.2 mm enamelled wire for LW and 80 turns of 0.3 mm wire for MW, wound onto a ferrite aerial approximately 20 cm in length with a diameter of 10 mm. The extra positive loop should be connected by tapping into the coil approximately a quarter of the way up from the earthed end. Keep all interconnecting wires and links as short as possible. The length of the coaxial cable is not critical. I elektor iuly/a 1982-7-53 The novel use of components in this electronic temperature indicator make it very simple and economical to build. It uses only three ICs, an LM335 temperature sensor, a723 (old faithfull) voltage regulator and a TL489 five stage analogue level detector. The temperature sensor (IC1) is supplied with a constant current from the reference output of the 723 (IC2). This provides a stable zero point setting enabling accurate readings to be achieved. The circuit around the 723 is arranged to allow the output of the regulator to vary between zero volts and one volt. It also acts as an amplifier with an effective gain of 20. The output is fed to the input of the analogue level detector IC3. Depending on the voltage level at its input, this 1C will light one or more of the LEDs D1 . . .D5. Since the sensitivity of the sensor is 10 mV per degree centigrade ( 1 0 mV/°C), and the gain of the 723 is 20x, it follows that the TL489 requires an increase in voltage level of 200 mV at its input to light each successive LED. Therefore, one LED will light for every 1 degree rise in temperature registered. Calibration is very straightforward. The temperature measuring range (or temperature 'window') is set by PI . For example 18 ... 23 C (5°C). This range can be altered if desired by simply changing the values of resistors R6 and R7. For two degrees temperature change per LED, the resistor values must be 100 kfl. H J, The duty cycle of a square wave signal is normally measured by means of a pulse counter or an oscilloscope. Ub (5 ... 20 V) However, this can be simplified considerably by using two VMOS-FETs and a voltmeter. The FETs are switched in turn by the input pulses. The R2/C2 network combi- nation provides an average DC level corresponding to the input waveform: U av - T/r • Ub- The meter reading can be interpreted as follows: The indication of the duty cycle can be expressed as a percentage (link A). For link 8, a voltmeter with a centre zero is preferable. A DVM would also do the trick, but not quite as well. The voltage level at the input of the meter will be half the supply voltagb when the duty cycle is 50%. Since the other side of the meter is connected to half the supply voltage (via voltage divider R3/R4) there will be no current flow through the meter (hence a zero reading). The duty cycle can be read directly in % if the scale is divided into 1 ... 1 0 (Ub = 10 V) and the centre point (5) marked as 50%. An important note: It is imperative to ensure that the input waveform switches abruptly between a low level (less than 0.8 V) and 'high' (Ub -0.8 V or higher). Between these values both FETs would start to conduct, thus causing a short circuit across the supply voltage source. Moreover, the maximum supply voltage must not be exceeded. One final remark: The internal resistance of the meter must be at least 1 00 kS2. 1982 This circuit makes it possible to control the speed of single phase motors with squirrel cage. This is not to say that every motor can now be made to run at any desired speed, but, that a speed range to a factor of 2 should be readily obtainable with suitable motors. That is to say that the range is from half to full speed. This range may not seem to be much, but, for fans, pumps and other equipment of this nature, it is quite a useful range. It can conveniently reduce both the current consumption, and noise levels of this type of appliance. The circuit described here makes use of an SGS Ates 1C that was specifi- cally designed for phase control. An asynchronous (short circuit rotor) motor has two windings, their mag- netic fields being at 90° to each other. One winding is connected directly to the mains, the other via a capacitor to ensure that the current passing through one winding is out of phase to the other. This invariably results in a rotating magnetic field, enabling the motor to run. Only the winding which is connected directly to the mains supply needs to be controlled, by what could be termed as a standard type of triac speed control. There are two main points to note. Firstly, irrespective of the speed set- ting of the controller, the motor will initially run up to full power (for a brief period), immediately the mains supply is switched on. Secondly, the current flowing through the motor is determined by the value of R8. The voltage across R8 is relatively con- stant, and held within well defined limits. This means that the speed of the motor (once set), will remain reasonably stable. The circuit is not suitable and was certainly not intended for use with motors which have varying loads (such as a drill). The minimum number of revolutions can be set by means of P2. Between this minimum (say 1800 rpm) and the maximum (3000 rpm), the speed can be controlled by PI . The circuit was designed to handle up to 90 W. Higher powered motors are possible but then R8 will have to be changed accord- ingly. The 1C deserves a special mention. Looking closely at the circuit diagram, you will notice that box 1 of the 1C derives a negative and positive supply voltage of 1 1.5 V from the mains, via R1 . The smoothing is effected by Cl and C2 respectively. The stabilised positive supply voltage at pin 6 is approximately 9 V. At each zero crossing of the mains supply, the ramp generator (saw tooth oscillator) in box 4 starts. The com- parator in box 5, compares the ampli- tude of the saw tooth waveform with the amplitude of the signal (output voltage) of the opamp in box 6. The output voltage of the opamp depends on the setting of PI, and therefore in turn, to the voltage across R8. As we have already explained the voltage across R8 determines the current flow through the motor. -E°D- 1982-7-55 With the aid of this circuit, the duty cycle of a signal may be adjusted very accurately in 1% steps within the 1% . . . 99% range. At the same time, it is possible to keep the frequency of the output signal completely independent of the duty cycle setting. Accurate pulse generators are needed whenever a meter or a circuit that calculates the level of a signal on the basis of its duty cycle and evaluates and/or processes the signal to be calibrated. The type of circuits in mind are remote control (PPM) and phase cutoff angle meters. The pulse generator in figure 1 can be constructed quite easily using three CMOS ICs. The decimal counters IC1 and IC2 are connected as divide-by- tens. Flipflop N2/N3 is set via R1/C1 upon the falling edge of the Q 9 signal of IC2 (which corresponds to the rising edge of Q 0 1) and the Q output of the circuit goes high. The inter- mediate count reaches gate N1 via the select switches S2 and S3. As soon as the required count is attained, N1 sends a reset pulse to the flipflop and the Q output goes low. Figure 2 shows what happens in the form of a pulse diagram. The clock signal may well be transmitted by an external device. As it is divided by ten twice, the output frequency will be 10 kHz at a maximum input frequency of 1 MHz. Alternatively, the internal oscillator may be switched on via SI , in which case an output frequency of between 20 Hz and 200 Hz, approxi- mately, (variable with PI) will be CLOCK juuuuuuiniuif HITTF 4.M-1 ll _._n n n n._ j.ji jl i | j j-j- ■pi [-H — j i | — | 1 r~ 1 , , Y ! ; 1 f i — ) j i i J j4= obtained at an operational voltage of 1 2 V. The preset range may be adjusted by altering the operational voltage (within the 5 ... 1 5 V range). In addition, the frequency range may be varied by selecting a different value for C2. Back to the pulse diagram. By way of an example, a duty cycle of 1 2% has been set here (see figure 1 ). Initially, the set pulse makes Q go high. But as soon as Q 2 of IC1 and Qi of IC2 are high, Q will go low again, etc. Supposing we wish to set the dwell angle of a 4 cylinder engine, we will have to take the following into account: the dwell angle is defined as a certain period of time, during which the contact breaker connections are closed. This corresponds to the time interval during which the signal is low. Thus, the definition of the dwell angle is the exact opposite of that of the duty cycle! What all this boils down to in this particular application is that the maximum dwell angle is 90°. This may be adjusted to, say, 54°. As a result, the variable duty cycle will be: (90° — 5 4^) _ 40% , 90° • 100% display data point connector If analogue signals are converted into digital and then displayed on an oscilloscope it will be obvious that the legibility will be less than perfect. This is due to the fact that the display consists of a (sometimes) large collection of short horizontal lines which can appear to have little or no relation to each other. Inter- connecting these 'dashes' will make the displayed information far easier to read and this circuit was specifically designed for this purpose. It produces a fairly complex 'waveform' on the screen but nevertheless, the legibility is considerably improved. In keeping with most good ideas, the operation of the circuit is very straightforward. An initial require- ment is a clock signal that becomes a logic '1 ' whenever the displayed data jumps to a new value. This can be derived from the circuit under test with the aid of a monostable. Opamp A3 is designed as an integrator which serves as a memory. If the incoming voltage level does not correspond to the voltage level at the output of A3, the difference between the two levels will be present at the output of A1 . Obviously the difference will be greater the more the new level deviates ‘“jJULOJL from the previous value. Consequently the output of A3 will change in an at- tempt to correct the 'error'. The rate of change will depend on how big the difference is, the greater the 'error', the faster the change will be at the output. Providing the R5/C2 combi- nation has been chosen correctly, the difference between the input and out- put voltage levels will be zero at the end of each cycle. Opamp A2 is simply a high impedance voltage follower and is included to ensure that the voltage level across Cl remains stable between clock pulses. Strictly speaking, ESI is not really required but without this switch the output would be an exponential curve which would reduce legibility. As mentioned previously, the time constant of the integrator must be identical to the data change frequency and the formula f = can be used as a rule of thumb for determining these values. The circuit can be cali- brated with the aid of a preset connec- ted in parellel with R5 if desired. H elektor july/august 1982 — 7-57 mini EPROMmer ' the simple programming circuit Fortunately the prices for widely available EPROMs is falling consider- ably. It might therefore be worthwhile to construct complex logic functions with EPROMs instead of the normal digital ICs (gates, flipflops, and so on). This would make the construction of the circuit much more compact and straightforward. The EPROM 2716 contains 1 1 inputs (address lines A0 . . . A10) and 8 data lines (D0 . . . D7), which are connec- ted as inputs during programming and as outputs for other functions. There- fore it is possible to program complex logic functions. For example a programmed EPROM can be used as code converter. This leaves us with the problem of finding a suitable program- ming device. It is rather expensive to build or buy a programmer, if it is only to be used occasionally. In this case a straightforward circuit i will suffice, with which the associated data of the logic functions can be stored in the EPROM quite easily. The circuit described in this article offers this possibility. Any program can be programmed step by step with the aid of this circuit. There is one crucial point which has to be considered, when using EPROMs and that is the access time. The oper- ation speed of the complete circuit depends on it. The circuit must be • constructed in the conventional manner, using gates, flipflops and so on, if the EPROM is too slow, due to the access time, for a certain application. The next question is what is to be pro- grammed? First, switch S21 must be set to position 'b'. In this case, pin 21 of the EPROM will be connected to the programming voltage and the data connections D0 . . . D7 are connected as inputs. The corresponding data can now be set bit by bit by means of switches SI . . . S8. An open switch then stands for logic 1 . After that, the corresponding addresses can be set with the aid of switches S9 ... SI 9. Again an open switch denotes a logic 1 . Once the correct data and address bits have been selected, depressing S20 is sufficient to transfer them into EPROM. The LED D9 lights to indicate the programming time. Obviously some form of check is necessary, when the complete pro- V gram is stored in EPROM, because the readers who have programmed by hand, will agree that it is very easy to make an error. Switch S21 in pos- ition a, in order to check the program. The LEDs D1 . . . D9 will now indicate which data is stored in the address set with S9 ... SI 9. A stabilised voltage of 5 V and 400 mA will be enough to supply the circuit, and a 30 V at 30 mA supply is sufficient to produce the programming voltage. sv( E*f ib 3b r 1 2b 2& The subject of power supplies seems to be of little interest since the introduc- tion of the well-known 3 pin voltage regulator ICs. However, the usefulness to the average home constructor is usually restricted to the versions that can deliver up to a maximum output of 1 A. Anything above this requires some form of heavy duty regulator stage. Regulator ICs capable of 5 A and 10 A do exist, but it usually works out more economic for most people to go straight into some form of discrete regulator. The idea of adding a power output stage consisting of one or more tran- sistors in parallel is not bad at all! For this reason it is applied, with one or two modifications, to the circuit described here. Power supplies that are insensitive to interference and can deliver high current levels to large microprocessor systems would cer- tainly benefit from such an approach. The ideal 1C for this job still remains the good old 723. This 1C may well have been over- shadowed by the new 3 pin regulators, but its versatality cannot be ques- tioned and its technical specifications are in many respects superior. It is used here in a standard circuit, in- tended to deliver output voltages between 2 and 7 V. The necessary supply for the 1C is obtained after voltage doubling of the smoothed and rectified secondary voltage of the transformer, via a voltage regulator, which in this case is of the three pin variety. This TIP 142 BD 139 I? BCE ECB ■Tf? reason that the secondary voltage of the transformer must be kept as low as possible, in order to hold the power drop across the series transistors T1 . . . T3 to within reasonable limits. While on the subject of power dissi- pation the heat sinks for T2 . . . T3 must obviously be sufficiently large. For the same reasons the values shown for R4 . . . R6 are best obtained by connecting several resistors in parallel. For R4 and R5 in other words, twice 0.33 fi 5 W, for R6 and an output current of 6 A twice 0.22 H 5 W or three times 0.33 Zl 5 W for an 8 A out- put. Furthermore these resistors must be mounted with plenty of space between them and the printed circuit board. The output voltage can be increased up to about 14 V if the following components are modified accord- ingly. The transformers, resistors R1, R2 and capacitors C5 and C6. The voltage doubling components Cl, C2, D1 and D2 are also unnecessary. The anode of D3 must then be connected directly to the rectified and smoothed supply. It should be noted that although the TIP142's look like any other power transistor, they are in fact Darlingtons. In other words, they cannot be re- placed by any ordinary power tran- sistors. One more point to give some idea of the good performance of this supply. The output voltage of the prototype was set at 5.5 V when loaded by a to 5.32 VI This is a drop of 3.3% at 7.8 A. Furthermore, under the same conditions the ripple was less than 25 mV rms . Resistors: R1.R2 - 3k3 R3 - 100 fi/1 W R4.R5 ■ 0,1 5 Jl/5 W R6 = 0,1 n/10W* Capacitors: Cl ,C2 = 470 p/50 V C3 = 220 u/50 V C4 = 1 p/16 V C5.C6 = 1000 V p/25 V C7 * 10 p/1 6 V C8 = 470 p B = 10 A/40 V bridge rectifier (not p.c.b. mounting) D1 . , . D3 = 1N4001 T1 = BD139 T2.T3 = TIP 142 (Darlington) IC1 = 7812 IC2 = 723 Miscellaneous: Tr= 10 V/10 Ate SI = double pole i 0.68 £2 resistor (which corresponds to d a current of 8 A). The voltage dropped r i i H b D1 ... D3 = V C & 1N4001 ^ ^2 r [ V '--'BD139 vSp j CS 2 ■i i ■[if 723 " cs^ 40 V/ 10 A alektor july/august 1982 - A descriptive and constructional article for an SSB receiver was published in the June issue of Elektor. The intention was to encourage readers to construct this type of equipment. It was mentioned at the time that the basic design could be used as the basis for other amateur bands providing a converter was available. This means that the receiver frequency must be mixed with an oscillator signal in such a way that the output is tuneable in the range of 14 . . . 14.35 MHz. The oscillator frequency together with the special component values required for the specific amateur band required are given in the table. The circuit itself consists of three sections; the input stage (VLF), the oscillator T2 and the dual gate MOSFET mixer stage T 1 . Components Table Band VLF 40 m 30 m Frequency Crystal (MHz) (MHz) 10... 140 kHz 14.0 1.8 15.8 3.5 17.5 7 21.0 10 24.1 L1/L2 Cl C2.C4 C3 (l*H) 86 dB the constructional tricks to reduce the inherent noise of the amplifier stage, in order to obtain a high signal-to- noise ratio. The preamp does not contain a coupling capacitor at its input, as this slektor july/a would produce additional noise. Therefore the transmission range already starts at the DC voltage At first sight the constructor might be worried about the large number of transistors, but you will soon find that it is not difficult to mount all components on the printed circuit board. This design does not suffer from oscillation tendencies or other 'semi-professional hobby amateur' problems. The price for the components is quite reasonable. The voltage regulator ICs are only required once and the com- ponents Cl 1 . . . C14 and IC1, IC2 can be omitted when constructing a second (stereo) channel. The connections II©, II© and ll©on both boards must be connected together. A small 2 x 15 V ... 24 V/50 mA transformer will suffice for the power supply. The value of the smoothing capacitors must at least be 470 pF . The input impedance of the preamp can be adjusted to any cartridge by simply changing the values of R 1 and Cl . The amplification factor is deter- mined by R 14. When using a 100S2 resistor for R 1 and a 27 S2 resistor for R 14, the preamp will be suitable for moving coil cartridges. In contrast to other preamps, the output connects directly to the auxiliary socket of the amplifier. The time base shown here uses a crystal for series resonance. This method achieves a greater stability factor than parallel resonance circuits. The two main requirements of the active elements are: 1 . The phase-shift between input and output must be 0°. 2. Both the input and output must be low impedance, in order that the Q factor of the crystal is not affected. This improves the stability. It therefore follows that a CMOS crystal oscillator cannot cope with the above requirements. A TTL version, although having very little phase shift (up to a frequency of 1 0 MHz), comes no where near to complying with the second parameter. The circuit described in this article meets both requirements. The design allows frequencies of up to 30 MHz to be produced without any phase shift. Higher frequencies are possible but then T1 and T2 will have to be changed for another type (such as the BFR 91), and the values of R1 . . . R4 will have to be reduced. Point 2 is well taken care of by the fact that the crystal is positioned between two emitters of a push pull stage achieving a low impedance input and output. The MOSFET buffer in the output stage 'insulates' the oscillator from any circuit connected to it. H 7-74 - elektor july/august 1982 After the transmitter using the SL490, published elsewhere in this issue, we come to the receiver, once again using Plessey ICs,SL480and ML920. Pulse pause modulation (PPM) is used with or without carrier, and automatic error detection is also incorporated. Although initially designed for TV remote control, the ICs can also be used for controlling 'HI-FI' equipment, lighting, toys and models. Figure 1 shows the circuit diagram of the pulse amplifier. This mainly consists of three gain stages, each being decoupled by capacitors, so as to achieve low frequency roll-off, there- fore eliminating AF noise. The transistor capacitor network around T1 actively simulates induction, pre- venting the diode D1 from saturating. In other words, it gets over the problem of high ambient light, such as sunlight, from saturating the receiver diode. The photo diode D1 (which is buffered), sends negative going pulses to the input of the 1C. This input is then amplified by the three stages, finally being inverted to give positive going PPM, compatible with the MOS decoder inputs. Figure 2 shows the circuit diagram of the actual receiver, using the ML 920. The M L 920 demodulates the PPM signal, but not into simple on/off commands! 3 outputs are available which can be split up into three groups. 3 analogue (A1 . . . A3), 5 digital (D1 . . . D5) and 5 channels (Cl . . . C5), which although specifi- cally for TV control can still be thought of as digital outputs. These five outputs allow the switching of up to 20 TV channels. The information is present at the 5 outputs in binary coded form; EDCBA = 00000 . . . 10011. This information remains the same until a pulse re-addresses them. Whenever a switch-over is required (from one channel to another), this switching operation is simultaneously followed by a pulse released from D4. The receiver automatically ignores any attempt at switching to a channel above 20, and also ignores any instruc- tion transmitted when more than one key (on the transmitter) is depressed at the same time. Should the channel information be required as separate outputs (instead of in binary), then the CMOS 1C 451 4 can be used to decode the information from binary. In this case the constructor must bear in mind that the M L 920 operates with negative logic. A logic 0 is inter- preted as the operational voltage, a logic 1 asOV. The analogue outputs of IC2 are used to control colour, volume and bright- ness. From now on it is probably better to itemise the pin functions of the 1C but that would take up most of the issue, so it may be better to refer readers who are really interested in building a TV remote control to the 1 PLESSEY consumer application notes available on the ML 920. Apart from the analogue outputs just described, the 1C has outputs for: on/off, recall display, AFC, mute, colourkill oscillator monitor, standby, step, and so on. Quite an extensive array of control facilities! Remote control data PLESSEY SEMICONDUCTORS ML 920. elektor july/august 1982 - 7-75 The problem of not having the right 1C to hand is an well known stumbling- block for constructors: When a VCO is required urgently, the ideal 1C is invariably not available and those that are will probably not suit the purpose. It is therefore very handy to be able to have something 'home-made' for emergencies. This circuit will make sure that your hair will turn grey because of age and not because of this particular problem. Whenever an oscillator with an adjustable frequency is required, it is desirable to use one that is voltage controlled, because this is as versatile as it is possible to get. Whereas a potentiometer is fine for manual setting, a control voltage is far more useful for automatic frequency control purposes. The circuit must have a wide frequency and supply range in order for it to be suitable for the majority of applications. This particular circuit has a frequency range of more than 1 : 1 000 and can be used from AF up to 50 MHz. The basis of the circuit is the well- known TTL Schmitt-trigger oscillator. The emitter follower T1 , connected in front of N1 , increases the input resistance and allows high values for the feedback resistor R1. The following section around T2 is the frequency control stage, which is connected in parallel to R1 . Diode D1 ensures that the capacitor charges very quickly. However, its discharge via T2 is controlled by the input voltage Uj. Therefore the output of the gate consists of a train of 'needle' pulses with a variable frequency. Strictly speaking R1 is superfluous, but it guarantees that the oscillator will start to operate, even in the absence of an input voltage. The pulse duration mainly depends on the propagation delay of the Schmitt- trigger used (N1). Standard and LS TTL need about 30 ns and S TTL about 15 ns. A divide-by-two circuit (N2 and N3) follows the actual oscil- lator. This supplies a square wave output signal of half the oscillator frequency. The top end frequency limit is 1 5 and 30 MHz for the LS and S-type respectively. With the very small coupling capacitors in mind, care must be taken with wiring. Further, a ceramic capacitor of 10 . . . 1000 nF must be fitted between pins 7 and 14 of the TTL 1C. Resistors R2 and R3 must be used with standard and LS TTL, in order to prevent the divider from oscillating. Negative feedback via C3 and D2 is provided to linearise the non-linear control stage of T2. A frequency proportional, negative voltage level is N. Rohde provided across C2. Resistor R4 determines the level and was calculated in this circuit for a control voltage range of 0 ... 10 V. The higher the control voltage, the bigger R4 can be, the better the linearity. Figure 2 shows the control characteristic of the oscillator with standard LS TTL (curves St) and with Schottky TTL (curves S). The negative feedback can be switched off by means of SI . The curves indicated with 'b' are produced when using the negative feedback switch in position V. 1 make provision for another alternative: the use of optocouplers to drive standard coloured light bulbs. A design on these lines, the 'big VU meter', was published in the January 1981 issue. Whichever circuit you choose, it won't be a 'flash in the pan!' □ P Q ri H H n o n stable start stop oscillator 'for video character generators Start/ stop oscillators are indispensible in video interface circuits. Such oscillators have to be synchronised with differentiated character clock pulses and produce 7 ... 1 2 pulses between character clocks. There are two aspects which are important to note here: • The oscillator must start producing pulses after a delay of about 1 5 ns. This prevents the first pulse (the output signal) from coinciding with the positive-going edge of the trigger signal. • The oscillator must stop as soon as the control signal goes low again. The oscillator shown in the circuit diagram meets both of the above requirements. It starts after a slight delay whenever the input signal goes high and stops immediately the input signal reverts to logic zero. Most T.V. games systems commercially produced allow the user to actually hear what is happening on the screen. When you shoot down a space invader, then an explosion or whatever is heard. It certainly adds to the overall enjoyment of the game. With the following circuit the Elektor T.V. games computer can now give you the extra audio effects needed to add that further touch of realism to a game. The left-hand side of the circuit shows all the connections to be made to the main printed circuit board of the games computer. After the flip-flops contained in IC1 come the data-lines D2 . . . D7. Data is switched from the input to the output on every negative going edge of the clock pulse. IC1 is enable when input B is addressed by line 1 E80. The effects produced really depend on the rest of the programmed data in the computer. The basis of the sound generation is transistor T4, which is connected as a noise source. A1 and A2 amplify this signal up to a usable level, making it available at the output of A2. A3 creates the explosion effect. With a logic 1 on the data line D4, A3 releases the noise signal suddenly! With a logic 0 on D4 the signal decays gradually with the speed of decay being determined by the rate C6 dis- charges across R17. A simple low-pass filter (R21, C7), feeds the signal to the programmable amplifier A4. The gain of A4 depends on the data present on lines D6 . . . D7. The amplification changes in steps of, 1 x, 1 %x, 3x and 4x, the highest occuring when data 00 is present. The audio output volume is controlled by PI . Finally an output power amplifier (IC3) completes the circuit. Points X and Y are connected to the outputs of the two programmable sound generators (PSGs) of the 0900 7620 0902 0C1E89 0905 9A7B 0907 04 FF 0909 CC1FC7 090C 0410 090E CC1E80 extended games computer. The PSGs together with this circuit should give you all the sound combinations ever needed. With a games computer which has not been extended and therefore does not have the two PSGs, either X or Y must be connected to pin 22 of the programmable video interface i . 1 ^ .1 ”|sl S r H® ^ BC 547 , |_?P\ fi © 1 ' ,f N “0 r& J L nr 8 or JC (PVI). Transistor T3, on the main board of the games computer is then not required. The sound generator requires a voltage of 12 V. The computer itself cannot supply this. However, if the main computer power supply transformer has a 1 2 V tap, then a simple supply can be constructed using a diode and a 7812 regulator, as shown in figure 2. The unit consumes approximately 1 5 mA from the +5 V supply, whereas the +12 V supply must be capable of delivering about 1 50 mA, with the volume control fully up. A change-over switch can be incorpor- ated, to allow the effects to be bypassed if required. In this case each PSG output is connected to a 10k resistor. The two resistors are inter- connected and fed to one side of the switch via a 100 n capacitor. The details are shown in figure 2a. Figure 2b shows the function of the different 'bits'. The table illustrates a demonstration program. Depressing 'WCAS' will produce the explosion effect. When the sound generator is switched off, depressing the same code will result in a loud hum being heard! liscellaneous: jgust 1982 This application of the 'miracle chip' LM/XR 13600 deals with a voltage controlled triangular oscillator. The OTA is fed back from the output to the input via the voltage divider consisting of R1 and R2. This feed- back from the output to the input is performed via capacitor C, which has a linear charge and discharge rate. The current through C also flows through one of the two diodes; there- fore the trigger points are at ± 0.6 V. The frequency can be calculated as follows: The output voltage is: R1 + R2 \. It is assumed that the OTA input differential voltage is always so high that the current through C equals the maximum lABC. which in its turn is identical to: U c + 15 Rc National /Exar Application M 1982-7-81 fcCG = electrocardiogram, EMG = electromyogram and EEG = electroencephalogram. All these 'grams' deal with measurement and display of electric voltages being produced by the heart beat (ECG), the muscular activity (EMG) and brain activity (EEG). The heart 'supplies' the strongest signals and the brain the weakest (didn't we all know that?!) Many microprocessor enthusiasts may have had some thoughts of performing physical tests by means of their computer. Unfortunately no suitable interface has been avail- able . . . until now; this circuit solves that particular problem. Three copper plates are used as electrodes. They are connected via screened cable to the differential amplifier which forms the input of the circuit. The circuit concsisting of A1 ... A3 can also be described as an Instrumentation amplifier'; a differential amplifier with opamps and two high impedance inputs. The output signal of this input stage is filtered by the active low-pass filter A4 before being fed to the 'transmit- ter diode' in the optocoupler. One essential remark: It is advisable to derive the operational voltage for IC1 from two 4.5 V batteries. This is the only sure method of guaran- teeing complete isolation of the measuring circuit from the power sup- ply of the microcomputer system. For obvious safety reasons we strongly recommend that a mains derived power supply is not used for the circuit! dissipation limiter The 'receiver transistor' in the optocoupler conducts the signal to IC2, where it is converted into a pulse-width modulated signal. The duty cycle of the output signal (at the 'shorted' input of the differential amplifier) is set to 50 % by means of P2. The frequency of the output signal can be selected with the aid of P3. Last, but not least, the ampli- fication factor of the input signal can be set with PI . Developing the software is up to the constructor. Those who are interested in bioelectronics and want to know something more about it can read the book mentioned at the end of this article. Literature: HolzAreysch, bioelectronics, Frankch, 1982. ' energy saving circuit Variable power supplies have to meet a lot of requirements which are very hard to realise from a technical point of view. The maximum output voltage must be as high as possible while the current capability needs to be at least one or two amps to be of some use. Constructors who have already tried to build their own power supply will know that the dissipation of the power transistors can become extremely high. One of our readers found a way to get around this problem for the majority of cases - and quite economically! Maximum dissipation occurs with high currents at low output voltage levels. For this reason switched primary windings on the transformer are used in many cases, as an effective way to limit the losses. However, the circuit shown here might present a solution to many readers who do not want the added expense of a transformer of this type. It is possible to realise double the voltage and half the current with the aid of a single switch contact, which can be operated manually or automatically. The two electrolytic capacitors are the most expensive components in the circuit. The existing power supply is inside the dotted lines shown of the circuit dia- gram. Either the normal full wave recti- fication or voltage doubling can be . selected by means of switch contact V SI . In the first case SI will be open. The transformer voltages shown in the circuit diagram are intended as an example. The circuit will function just as well with other voltages of course, on the condition that the electrolytic capacitors and transistors are able to cope with these values. Automatic switching can be achieved by the circuitry constructed around T1, T2 and a relay. As soon as the output voltage of the stabilisation circuit exceeds 30 V (this value can be set by varying R3) T2 will conduct and the relay will drop out. SI , which is a normally open contact of the relay will now close, so that voltage doubling is achieved. The auxiliary circuitry with T1 and T2 can be fed from a separate supply, preferably with a voltage that has the same value as that of the relay coil. However, it is also possible to derive this supply from the voltage across both smoothing capacitors. In this case particular attention has to be paid to the fact that T1 and the relay must be able to cope with the maximum voltage and T2 should be able to deal with at least half of this value. 7-82 - elektor july/ai 1982 stereo power amplifier / a complete stereo power 'amplifier on one chip National Semiconductor's, LM 2896 contains not one, but, two high performance power amplifiers able to handle supply voltages up to 15 V. With a 12 V supply the 1C can deliver 2,5 W per channel into 8 fl. With the same supply and load, it is capable of delivering 9 W in 'bridge mode'. These are certainly good performance figures, especially when you consider the low number of external components needed. Figure 1 shows the circuit diagram of the complete amplifier. As you will note, the components for each channel are identical. Resistors R1 and R2 together with capacitor C2 form the negative feed- back loop. The band-width of the amplifier is determined by R2 and C3. R3 and C4 ensure maximum gain, frequency response (-3 dB) 30 Hz ... 30 kf with R4 and C6 stabilising the output. Capacitor C8 smoothes the supply, eliminating any possible 'spikes'. When operating in stereo mode, coupling capacitors (C5) are required at the output. Figure 2 shows the track pattern and component overlay for a stereo version using a single 1C. A 10 k log poten- tiometer at the input is sufficient for controlling the output volume. When using the amplifier in 'bridge mode', certain changes have to be made. These are denoted by dotted lines on the track pattern and circuit diagram. Obviously in order to achieve high power in stereo, two complete circuits are required. Figure 3 illustrates the output power to supply voltage characteristics of the amplifier, for different modes and loads. When operating in 'bridge mode', RB and CB must be added, and the coupling capacitors C5 removed. power failure ^protection Nothing can be worse than having even a brief collapse of the mains supply voltage when working with a system using volatile memory, like RAM. After the interruption, no matter how small, it will be apparent that the data in the RAM has well and truly evaporated. For that reason a lot of circuits are designed to side step the problem of either long, or short term, mains supply failure. The circuit described here can be placed into the same general category. An additional bridge rectifier is added to the existing power supply, together with a relay Rel in series with resistor R1 . The contact for the standby power supply of 1 0 - 1 5 V is made by Rel . The circuit must detect a mains voltage collapse as early as possible. As soon as Rel is no longer activated, the batteries take over. Obviously, no matter how quickly this changeover takes place it will take a finite period of time, therefore capacitor Cl must be able to supply the necessary current during this period. Any slight drop in voltage across this capacitor is catered for by the regulator IC1 . An AC relay can also be used and, in this case, the bridge recti- fier B2 can be dispensed with. When using a DC type, the hold voltage of the relay should be about 1 .2 V below that of the secondary voltage of the transformer. The following formula should be used to establish the correct type. • UrmS-\/ 2~ Uh - 1-2 ‘V T T. 0-9 • URMS ~ 1-2- Uh R! = the series resistor in SI, Rr 6i being the resistance of the coil of the relay. Uh is the hold voltage, lh the hold current, and 1 .2 V is the tolerated voltage drop across the bridge rectifier. The relay should be sufficiently slow to bridge the gap when the voltage drops below the 'hold' level, but, not too slow, otherwise Cl will get into difficulties and cause the relay to 'buzz'. The tighter the operating tolerances, the faster the switch-over to the standby power supply. Remember that the standby supply does not necessarily have to power the complete system, but only the RAMs. In this way the accumulator will last that much longer. It is possible to trickle charge the accumulator by connecting it via a series resistor from the voltage across Cl (in parallel with the relay contacts). The value of the resistor will depend on the specific accumulator (NiCad) in use. st 1982 Since their introduction in the early 70's OTAs have become a classical component for voltage controlled filters. This is especially true of the dual OTA XR 13600 because it already contains the necessary buffer stages. The dual version has an excellent synchronous operation and is ideal for second order filters. The circuit diagram here shows a low-pass filter of this type. A modulation range over several decades can be obtained, with a good linearity. The 3 dB cutoff frequency of the filter depends on the transconductance (g m ) of the OTAs, and on the values of the resistors R and Ra and the capacitors C and 2 • C. The value of fg can be calculated from the following: , Ra 9m 9 "(R + Ra)2ttC RC must be multiplied by two, as the The next question is: How do we know the g m value? This is really quite simple; at room temperature the g m = 1 9-2 • I b. where I b is the current that flows into pins 1 and 1 6 of the 1C (across Rc). The voltage at these pins is approximately 1.2 V more positive than the negative supply voltage (or —13.8 V with a ± 15 V supply voltage). We can now extend the first formula as follows: 9m = 19-2 • ■ 2 • R C current across Rc is divided between both OTAs. The data with the values indicated in the circuit diagram are: control characteristic; approximately 2 kHz per volt fg at U c = 0 V, 28 kHz f g at U c = -13 V, 1.5 kHz fg at U c = +6 V, 40 kHz Another value for the control charac- teristic and modulation range can be obtained quite easily by changing C and Rc. National application The circuit diagram shown here is a National Semiconductor application of the LM/XR 13600, in this case used as a kind of state-variable filter. The circuit contains a selective filter output (ul) and a low-pass filter (u2). The centre frequency of the selective filter and the cutoff frequency of the low-pass filter can be influenced by the control voltage level u c . Both integration capacitors C determine the range in which these frequencies can be varied. The corresponding formulas are: p = jw; r = |; S=19.2 -IabC; lABC^J^ R c =15kfl ui 42 pr uT "* 462 p 2 t 2 +21 pr+ 1 selective (bandpass) filter “a = 2 u i 462 p 2 r 2 +21 pr + 1 low-pass filter Cutoff frequency and central ___ Literature: frequency respectively are: «o ' When is an OTA not an OTA?, and The 13600, a new OTA' , ~21r 15 V 9 both published in the April 1982 issi of Elektor. simple frequency converter a TBA 120 application During the last few years the TBA 1 20 has become one of the most frequently used ICs in RF techniques. Although originally meant as IF amplifier/FM demodulator, the TBA 1 20 can be used for a wide range of applications. This converter circuit •s just one example. The initial requirements for a converter are a mixing stage and an oscillator. The multiplier in the 1C suits the needs of a mixing stage perfectly well. The oscillator can be realised by a selective (positive coupling) feedback of the amplifier section of the TBA 1 20 by means of the resonance circuit L1/C1. The oscillator will operate at a frequency of 46 MHz with the values indicated in the circuit diagram. Consequently, we are dealing with a circuit that converts an input signal of 35.3 MHz into 10.7 MHz 146 - 35.3 = 10.7 MHz). This can be used to convert the IF signal of a TV tuner into the intermediate frequency of an FM receiver. Obviously the circuit can also be applied for other frequencies, by modifying the oscillator circuit 'Ll/Cl) and the output filter (L2/C2) accordingly. When the oscillator frequency is considerably lower than 46 MHz, Tl IC1 3 A 120 ; HI r hj£ •1 , j ¥ Tl H>X Ox i II T- T- 1 s 3^ r v. the values of R 1 and C3 have to be increased slightly. However, their value is not very critical and can be determined quite easily after some experimenting. The construction of the converter is very straightforward, due to the fact that only a few components are required. However, some attention has to be paid to the common basic rules for RF circuits, such as: • Try to retain as much 'ground plane' as possible, when etching the printed circuit board. • Keep the tracking and wiring as short as possible. • Use the shortest distance from the point to be decoupled to the ground for the decoupling capacitors C4 . . . C8. high performance video mixer j find the right combination! Terminals, (the interfaces between computers and video screens), have to output two synchronisation signals in addition to the actual video signal. The Elekterminal also contains a video mixer which combines the two signals into the single video display control signal. The H and V sync signal control the horizontal and the vertical deflections of the electron beam, respectively, while the video signal incorporates the picture infor- t mation. All three signals are com- V bined in the mixing stage around ^.Tl and T2. 0_n_n_ T2 mixes the sync signal; the transistor forms a NOR gate together with R2 and R 3. Transistor T1 operates as an emitter follower. PI sets the amplitude of the output signal, enabling the circuit to be adapted to any type of monitor and/or TV set. A monitor will have to be used, should your TV not have a video input socket. The video combiner is suitable for band- widths up to 25 MHz. rear light monitor ' an effective dashboard monitor Even though car dashboards are beginning to resemble the control panels inside a cockpit, it is surprising how many LEDs are in fact totally superfluous. What is the point of an LED that indicates whether a switch is on or off, but fails to monitor the actual function of the equipment connected to it? Take the rear fog warning light LED, for instance; it will continue to burn irrespective of whether the light is working properly As it only requires five components, it can be fitted behind the existing switch. This is what's required. Break the ground connection (if included) of the switch LED and the connection between the switch and fog light (or any other that is required to be monitored). Now install the circuit as shown in figure 1. There should be plenty of room for the unit in the vicinity of the switch in question. Operation is straightforward: If everything is O.K., the load current will flow to ground via R 1 and Lai , the fog light. The voltage across resistor R1 will then be sufficient for transistor T1 to conduct and the switch LED to light. Should the bulb Lai fail for any reason, T1 will not receive igh base current and will stop conducting. In that case, T2 will also stop conducting and the LED will go out. The value of resistor R 1 may be calculated according to the following formula: ' elektor july /august 1982 - 7-87 in the April issue of Elektor, we published a circuit for a contact tester with an acoustic indication. As a result of this publication we received a number of requests from readers for a contact tester with an optical indication. The circuit described here fits that particular bill rather nicely. Like the original design this circuit has [ its own printed circuit, the only difference being that this one uses a LED, rather than a buzzer to denote a good contact. The theoretical aspects of this circuit were discussed in detail in the April issue so for now we will restrict ourselves to recapping the calibration procedure. Place a 1 12 resistor be- tween the probes and adjust PI until the LED is just about to light. Remove the resistor and create a short circuit between the probes. The LED should i fiow light. T o make sure that calibration is correct, place a resistor of only a few ohms between the probes. If the LED lights up now, the calibration procedure will have to be repeated. After correct adjustment, only resistances of up to 1 12 will be tolerated. A value lower than this will either indicate a good contact or a short circuit. Keep in mind that the supply voltage of the circuit under test should be switched off, otherwise the tester could be damaged. As long as the LED is only allowed to remain lit for short periods, the consumption of the tester will not exceed 8 mA. The battery should last at least a year. Resistors: R1.R3- 22 k R2- ion R4,R5 ■ 1 k R6 = 470 k R7- 1k2 Capacitors: Cl - 10p/10 V Semiconductors: IC1 = 741 IC2 = 4093 D1 = 3 mm LED red Miscellaneous: PI = 10 k preset SI = single pole switch P.C.M. Verhoosel 1 7mA iuly/august 1982 This AC/DC converter 'translates' the value of an AC voltage into a corresponding DC voltage. It allows AC voltages to be measured with the aid of a high impedance (DC voltage) voltmeter. The circuit diagram shows an active rectifier which is designed around a CA3130. It contains a few little tricks that make it possible to ap- proach the effective value measure- ment as closely as possible. The signal to be measured is fed to the non-inverting input of IC1 via input capacitor Cl . Diodes D3 and D4 protect the input against excessive voltages. The capacitors C4/C6 and C2 make sure that the output and negative feedback are only AC coupled, so that any offset of IC1 will not effect the measurement result. Resistors R1 and R2 look after the DC setting of the 1C, while R3 takes care of the DC amplifi- cation factor (lx). Bootstrapping is achieved by C2, which consider- ably increases the input impedance of the circuit. D2 will conduct on a positive edge of the input signal, at which the ampli- fication factor of the opamp is determined by the relationship of the resistors R4, R5 and the setting of potentiometer PI . Capacitor C5 will then be charged via resistor R6. During the negative edge of the input = (^E3- BS 170 BC 549C signal D1 will conduct, causing C5 to discharge again, but only partly, because (a) the gain of the opamp is only lx when D1 is conducting and (b) because the resistance value across which C5 must d ischarge is larger than that when it discharges. This relationship has been calculated so that the DC voltage across the capacytor equals the effective value of the input signal. Actually this is an average-value-measurement that is corrected before giving the effective value. Obviously this only holds good for sine wave signals. The circuit requires a symmetrical supply having a value between ±2.5 V and ±8 V. The current consumption is slightly more than 1 mA. Figure 2 shows how the converter can be used with a voltmeter, in fact, the LCD meter published in October 1981 issue. In this case, R1 , R2 is a wire link, R8, D1 and D2 are omitted; connect link A. The voltage divider is used for AC as well as DC voltages. The decimal point of the display can be switched by adding an extra con- tact to switch SI . Since the voltmeter itself produces an artificial 'zero', a 9 V battery will suffice as power supply for the converter. Of cource, it is possible to use any voltmeter, as long as its input impedance is 10 Mft or more. The LCD meter must be calibrated on the 200 mV range with switch S2a in the DC position before the AC/DC elektor july/august The use of the small circuit described here together with the output routine in table 1 , enables the high-speed printing of information from the SC/MP. With the aid of the Elektor terminal, data can be displayed on the screen at a rate of 19200 baud. In effect this means that a 4K 'string' which would normally take 38 seconds to print (at 1200 baud) can now be dis- played in approximately 2.5 seconds, display on a VDU at approximately this speed. In practice the 74LS373 is used in this circuit as a latch and three stage output buffer. The data on the SC/MP bus is latched when the decoding address (in this case 0800 . . 09FF) and the NWDS are at logic 0. Simul- taneously to this, because the software controlled Flag 2 is at logic 1 , a pulse (between 0 ... 5 V) is latched. As a .jesult of all this, the UART of the elektor terminal, is brought into tri- state operation, and the data in the latch is transferred into the output buffer of the 74LS373. The outputs of IC1 and Flag 2 are connected to the Elekterminal at the pins shown on the circuit diagram. Pins 4 and 1 6 of the UARTs have to be disconnected. N1.N2 = N3 - % Table F. de Bruijn OUTPUT ROUTINE. JUMP WITH 3F IXPPC 3) CAN BE SHIFTED. FFE3 01 XAE SAVE BYTE. FFE4 06 CSA SET FLAG 2. FFE5 DC04 ORI X'04 FFE7 07 CAS FFE8 C408 LDI X'08 LOAD OUTPUT ADDRESS. FFEA 37 XPAH 3 FFEB CAE6 ST X'E6 (21 SAVE P3 HIGH. FFED 40 LDE GET BYTE. FFEE CBOO ST X'OO (3) STORE OP OUTPUT ADDRESS. FFFO D460 ANI X'60 INSTRUCTION OR CHARACTER? FFF2 9C02 JNZ X'02 TOFFF6 NOT 0, SO CHARACTER. FFF4 8F08 DLY X’08 WAIT. FFF6 C2E6 LD X'E6 (2) GET OLD P3 HIGH. FFF8 37 XPAH 3 FFF9 06 CSA CLEAR FLAG 2. FFFA E404 XRI X'04 FFFC 07 CAS FFFD 3F XPPC3 BACK TO MAIN PROGRAM. FFFE 90E3 JMP X'E3 TO FFE3 When making connections to a three- ahase mains supply, it is often essential to get the three phases in the correct sequence. Otherwise motors, for instance, have a tendency to rotate the opposite direction — which can have surprising results. Pumps become 1 suckers, and suckers become . . . forget it. In this well-regulated nation of ours, all connections of this type must be made by qualified electricians, so nothing can go wrong. End of article. 7-90 - elektor july/august 1982 for those readers who are not qualified electricians. For those readers who are still with us, the device described here can prove quite useful. In a nutshell, when its three inputs are connected to the three phases (the neutral connection isn't needed for this test), one of two LEDs will light to indicate a clockwise or anticlockwise phase sequence. In this connection (I), 'clockwise' is defined as U, V, W (or V, W, U or W, U, V) and corresponds to the green LED. Anticlockwise, not surprisingly, is the other way 'round'; the red LED will light. The basic idea can be derived from figure 1 . This is a plot of the three phases; as can be seen, at the zero- crossing of one phase the following phase is positive and the third is nega- tive. This is quite easy to detect! To simplify the connections, an artificial 'neutral' is created at the R1/R2/R3 junction. Only two of the phases are then used in the actual measurement; their value with respect o the artificial 'neutral' is detected, and used as .follows. V At each negative-going zero-crossing of ^^he voltage at the U input, the flipflop (FF1) clocks in the value at the W input as data. If the phase sequence is 'correct' (clockwise), the W input should be negative at this point — as can be seen in figure 1 . This means that T1 is blocked, so that a logic 1 is applied to the D input of the flipflop. The actual clocking of the flipflop is done in a similar way. by means of T2. When the logic is clocked through to the output, T4 will conduct. This causes the green LED to light. If the phases are inverted (anticlockwise), T2 will be conducting at the negative- going zero-crossing of U. This means that a logic 0 is clocked into the flip- flop. T3 will then conduct, and the red LED will light. Obviously, swapping any two phase connections will convert one phase sequence into the other. The two zener diodes (D1 and D2) protect the transistors - both against excessive base drive and against negative base voltages. Two final notes. For safety reasons, the complete unit must obviously be mounted in an insulating (plastic) case; the switch must also be a 'safe' type! Furthermore, battery supply is a 'must': try to imagine what might happen with a mains supply! junior paperware ' good news for Junior Computer fans Volume four is the final book in the Junior Computer series. Together with the additional system software, published in the April (Basic on the J.C.) and the May issue (Software cruncher and puncher) of Elektor, the books form a very useful library. Obviously there is a lot more new hardware and software for the Junior Computer that could be published! The problem is not what should be published, but how ? Hex dumps and source listings take up a lot of space and we would also like to keep Elektor interesting for readers who do not possess a Junior Computer. A book is another possibility, but it would take too long and for technical reasons would be too expensive. The ideal solution is to combine the best of the two possibilities and in this way come to a compromise. Therefore we intend to publish a certain number of articles under the title of 'Junior V Paperware', a kind of international copy service, consisting of several pages of A4 size. Production can be reasonably quick and cheap, which is a good thing for us as well as the readers. The first volume. Junior Paperware 1, is already available. It contains ad- ditional information concerning the 'software cruncher and puncher', including the hex dump and source listings. We have enough material for further 'Paperware' publications. For instance, additional details about the Junior Basic, a text editor/assembler and many other short subjects. Last, but not least, it will contain a lot of programs that have been sent in by industrious readers. Many thanks and keep them coming! In short. Junior Computer owners will not be able to complain of getting bored. ' july/august It is quite easy to connect the Elekterminal, or any other terminal which is equipped with a UART, for that matter to a low cost printer. Most if not all low cost printers incorporate what is known as a ‘centronics inter- face'. Basically the reason for this is that Centronics were one of the leaders in the field of budget printers, and as a result their original interface design has been used by a large number of manufacturers as an industry standard. The unversally available Epson MX 80 is a prime example. The advantage of using such a printer is that the I/O routines do not have to be altered. The Elekterminal already has a UART, converting the serial bit output of the computer into a 8 bit parallel code for video RAMs. So, it is just a simple matter of using the same code to drive the printer in parallel. Link data lines DO ... D7 on the printer interface circuit (which forms part of the printer) to connections BO ... B7 on the Elekterminal printed circuit board. It is obvious that there is no B7 terminal available on the board, so a new terminal has to be made up. This can be derived by making a connection to pin 5 of the UART. The next stage is to link up the strobe input of the interface to point T on the Elekterminal. In some cases the printer may go hay- wire, while the computer will still continue to apply data. This is simply because minor differences may occur as far as the interface specifications are concerned. Should this happen then the following procedure has to be adopted. • Connect the 'printer' busy line directly to the 'clear and send' line on the serial output port (such as ACIA) of the computer system, and not via the Elekterminal. As a result, the output data will then be kept back, allowing the printer to work without interruption. • Connect a 4k7 resistor between the CTS line and ground. This ensures that the line is disabled whenever the printer is not in use, allowing the operator to continue work with the computer even when the printer is switched off. A very important point to keep in mind is that the UART must receive the correct bit pattern from the computer. This should be: 8 bits, no parity and 2 stop bits. Any dis- crepancy or deviation from this pattern may prevent the printer from acknowledging the most significant bit containing the logic 1 for a character, making the printer virtually useless! N 7-92 - elektor july/augutt 1982 lit and keysoft for polyformant The actual sound generators of the poly- phonic synthesiser are still going to be analogue. All ten synthesiser channels consist of voltage controlled circuits (VCO, VCF, VCA). Therefore they require analogue control voltages to determine the pitch, and gate pulses which effectively start and stop the envelope generators. However, the microprocessor in the digital keyboard (on the CPU card), only supplies binary coded data (bits). Furthermore it does not address all ten channels simul- taneously; instead, it deals with them in turn. First channel 1, then 2 and so on. One cycle is completed when channel 1 0 is updated, after which new data is applied to channel 1 . Therefore the out- oiilpuf unit and keysoft fin* polyfbrmant the final stage of the polyformant together with the software and useful hints After the CPU described in the May issue and the 'Polybus' published in last month's magazine it's time to add the finishing touch to the project. The output unit ensures that each channel receives its respective correct information in the right order, such as control voltage, gate pulse and so on. This is the last unit needed to complete a basic version of a polyphonic synthesiser. U. Gotz and R. Mester put unit forms an essential interface, converting digital data into analogue control voltages and gate pulses. It distributes them to each synthesiser channel in the correct sequence and at the right time. Three completely different principles can be applied to analogue/digital conversion and distri- bution. Before describing the circuit of the out- put unit in detail, a summary of all the possible solutions is interesting. Static procedure and multiplexing The block diagram in figure la shows that a digital memory preceeds each D/A converter; the inputs of all these memories are all connected to one data bus which is fed by the CPU. The allo- cations of data to the VCO is performed by the 'enable' inputs of the RAMs (used as latches): For exampl e, latch 1 only receives the order WRITE from the CPU when the correct data for VCO 1 is on the bus. A multiplex procedure with software refreshing will also work, and it uses less components. The multi- plexer, controlled by the CPU, ensures that the voltages supplied by a single D/A converter are fed to the corre- sponding sample-and-hold stages of the VCOs (figure 1b). However, the CPU has to drive the multiplexer almost continuously; the capacitors of the sample-and-hold stage have to be re- charged again and again, at very short intervals. Since every byte is going to be needed when the polyphonic keyboard is extended, (presets keyboard-splitting) it seems a good idea to add a hardware counter that takes care of the 'read' from memory. The principle of multiplex operation with hardware refreshing, is the third method and the one used for the poly- formant. The hardware refresh cycle Every time a new key is depressed its value has to be stored in RAM. The counter transfers this key value to the RAM via the data bus. The bit pattern on the address-bus of the computer determines in which memory location the key value is stored. The CPU ad- dresses the RAM via a data selector MUX (see figure 1c). This data selector has two input busses and one output. The input busses are connected to the address-bus of the CPU and the output of the hardware-re fresh co unter. The logic level on the WRITE line deter- mines whether the computer address bus or the hardware-refresh counter is connected to the RAM; the CPU addresses the RAM when it writes a key value into memory. The RAM reverts to the 'read' mode once the key value has been stored. The memory addresses are scanned consecutively by the external hardware-refresh counter. Each VCO is allocated a specific mem- ory location. This means that the multi- plexer, (which distributes the D/A con- verted output), must always drive the same channel, when the corresponding location is read. This permanent allo- cation is obtained by interconnecting the address inputs of the RAM and the multiplexer. As before, only one D/A converter is required: in this case the Ferranti ZN426, an inexpensive 1C that fits the bill extremely well. Figure 2 shows the circuit of the output unit and the connections to the bus board on which the D/A converter is mounted. All the necessary connections to the bus should be made by using a multiway plug and socket, in the same manner as the CPU and input unit. IC3 is a BCD which is addressed by inputs A 0 . . . A3. It releases the single id keysoft for polyfc elektor july/august 1982 - 7-93 latches IC5 1 . . . IC5 10, consecutively. Each latch is in fact released via its 'enable' input every time the respective data for a particular channel is on the bus. The data actually reaches the bus via the driver IC4. The AND gate N1 . . . N6 take care that the WRITE pulse at pin 11 stores the data at the right time in the latch. The information at the outputs of the latches is permanently available to the D/A converter, therefore eliminating the need for any interruption to allow it to The D/A converter As already mentioned before, multi- plexing with hardware-refreshing, only requires one D/A converter. Unfortu- nately at the time of going to press, the prototype output unit has not been completed. Therefore, despite the high cost, anyone wishing to build a complete synthesiser will, for the time being, have to build it using the static principle, constructing as many converters as there are VCOs. But do not get alarmed! During the following months a new book on the Polyformant synthesiser, should be published, incorporating all the circuits and information needed for the multiplexing hardware-refresh ^ystem using only one D/A converter. Realising the circuit as shown in fig- ure 1 b is not as simple as it may seem! To keep the costs as low as possible the Ferranti ZN426E-8 was used. It is a very accurate and reliable 1C mainly due to its own internal reference voltage source. Each D/A converter circuit will require two of these chips. Even though we are dealing with an 8 bit converter with only four inputs connected, two are required for the following reasons: The computer determines the keyboard output voltage (KOV) level by com- paring two different sets of data. Firstly, which octave, and secondly the number of semitones being called for within that octave. For example code 3.7 could represent the seventh note (F sharp) of the third octave. The word 'could' is in the sentence simply because it is not the real software digital coding used, but only an expression to try to explain the basic principle. The D/A has to decode each octave 1 V at a time, as the VCOs produce 1 octave per 1 V. For the notes within any given octave the voltage sup- plied to the VCOs changes in one twelfth of 1 V per semitone. To interface the converter both outputs must be fed to a non-inverting adder, by using two opamps. The other two op- amps operate as impedance converters. Mechanical construction of the output unit Figure 4 illustrates the way in which each converter board is mounted onto the output unit main board. The con- struction is basically in the same format as the bus boards. The beauty of this method is that further extensions to the synthesiser can be made easily. Keep in mind that a D/A converter is required for every 'voice' or channel used! The converter printed circuit boards are quite small, therefore the wire link connections to the main board are sufficient to give the overall construc- tion ample structural stability. Each converter board has a KOV and gate pulse output. The method used to con- nect these to the analogue sections of the synthesiser was described in great detail in the Polybus article published in the May issue. The printed circuit board pattern and component overlay of the D/A converter is clearly shown in figures 5 and 6. Calibration of the D/A converter In order to calibrate the converter easily and correctly the tune shift printed circuit board has to be used. This circuit ensures that the correct digital data from the keys is fed to the D/A boards. Needless to say only one D/A converter at a time can be calibrated. The first stage in the procedure is to connect a digital volt meter (DVM) or any accurate instrument to the KOV output and the ground connections of the converter. We suggest the use of a DVM, as the readings have to be accurate and a digital display is much easier to read than a normal moving coil instrument. Next depress any key of the keyboard, and measure the voltage. Keeping the keyboard key depressed, push down and therefore switch on the first DIL switch of the tuneshift circuit. By the first DIL switch we mean the lower octave switch. key soft for polyformant elektor july/ai 1982 - 7-95 not advisable to set PI before P2 as this will lead to incorrect overall tuning. in other words the actual first switch ooking at the circuit from left to right. Readers who have not yet built the tuneshift unit should refer to figure 4 of the polyphonic synthesiser article published in the May issue. That dia- gram shows the DIL switch as being S4. Once again keeping the same keyboard