■1 Breadboard 1980 "^1

. 26th to 30tb November “
Open 1 0 a m. till 6 p.m. land till
8 pm Thursday. 4 pm. Sunday)
Royal Horticultural Halls,
Elverton Street. London

iNr Si wmeiiPan underground!

Visit our huge stand and
, see our new 'single-chip' organ,
a new sequencer /composer. '
T and lots more

mga Don t miss ltd |J

A massive new
catalogue from Maplin
that’s bigger and better
than ever before. If you
ever buy electronic
components this is the
one catalogue you must
not be without. Over
300 pages, it’s a
comprehensive guide to
electronic components
with thousands of
photographs and
illustrations and page
after page of invaluable data.

We stockjust about
every useful component
you can think of. In fact,
well over 5000 different
lines, many of them hard
to get from anywhere
else. Hundreds and hundreds
of fascinating new lines,
more data, more pictures and
a new layout to help you
find things more quickly.

'i^ii^piin

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noise reduction 2-04

Noise reduction systems have been in the news for about ten years and this
article takes a look at the various systems currently available. Particular
emphasis is paid to the High-Corn system from Telefunken as this will
form the basis of a constructional project in the near future.

process timer u. Meyer) 2-10

This timer will indicate the different time periods for the development,
stop bath, fixing and rinsing phases and in the correct order by means of
calibrated cards.

high voltage from 723 2-14

An economical and effective way of achieving a stabilised 60 V power
supply is by 'floating' the well known 723.

juniors growing up 2-16

A survey of the proposed extensions for the Junior Computer.

the voiced/unvoiced detector if. vi*»r) 2-17

The finishing touch to the Elektor vocoder is presented in this article on
the voiced/unvoiced detector combined with a noise generator.

coming soon 2-26

emergency brake for the power supply 2-26

Faster than a fuse in those situations where it matters.

1 50 W DC to DC converter for the car 2-27

This article describes a converter which will provide up to 60 V from the
car power supply.

low noise 2 metre pre-amp 2-32

2'A digit DVM 2-33

Three displays, six ICs and a handfull of components are the only ingredi-
ents needed to cook up this digital voltmeter.

wagnephon (Dr. Wagner) 2-36

A simple answer for those readers who would like to play the piano but
are unable to.

bookshop 2-39

market 2-40

missing link 2-44

advertisers index UK 22

TECHNICAL EDITORIAL STAFF

Holmes

elektor

1981

Why the Sinclair ZX80
is Britain's best-selling

Including VAT, post and packing, free course in computing, free mains adaptor.

Including VAT, post and packing, free course in computing.

This is the ZX80. A really powerful, full-facility
computer, matching or surpassing other
personal computers at several times the
price. 'Personal Computer World' gave it
5 stars for 'excellent value'. Benchmark tests
say it's faster than all previous personal
computers.

Programmed in BASIC - the world's most
popular language -the ZX80 is suitable for
beginners and experts alike. And response
from enthusiasts has been tremendous -
over 20,000 ZX80s have been sold so far!
Powerful ROM and BASIC interpreter /
The 4K BASIC ROM offers / A

remarkable programming /

advantages: /

* Unique one-touch' key word /

entry the ZX80 eliminates / ’

a great deal of tiresome /

typing. Keywords

(RUN. PRINT. LIST.

etc.) have their own

single-key entry. ^ —

* Unique synta* check. " S55!

A cursor identifies errors immediately.

* Excellent string-handling capability-
takes up to 26 string variables of any
length. All strings can undergo all
relational tests (e g. comparison).

* Up to 26 single dimension arrays.

* FOR/NEXT loops nested up to 26.

* Variable names of any length

* BASIC language also handles full Boolean
arithmetic, condition expressions, etc.

* Randomise function, useful for games
and secret codes, as well as more serious
applications.

* Timer under program control

* PEEK and POKE enable entry of machine
code instructions.

* High-resolution graphics.

* Lines of unlimited length.

Unique RAM

The ZX80'S 1K-BYTE RAM is the
equivalent of up to 4K BYTES in a
conventional computer-typically storing
100 lines of BASIC.

No other personal computer offers this
unique combination of high capability and

The Sinclair teach-yourself
BASIC manual

If the specif ications of the Sinclair ~

ZX80 mean little to you -don't worry
They're all explained in the specially-written
128-page book (free with every ZX80) The
book makes learning easy, exciting and
enjoyable, and represents a complete
course in BASIC programming -from first
principles to complex programs.

Kit or built - it's up to you

In kit form, the ZX80 is pleasantly easy to
assemble, using a fine-tipped soldering iron
And you may already have a suitable mains
adaptor- 600 mAat 9V DC nominal
unregulated. If not, see the coupon.

Both kit and built versions come complete
with ail necessary leads to connect to your
TV (colour or black and white) and cassette
recorder Plug in and you're ready to go (Built
versions come with mains adaptor.)

TheZXBOast

principles ol computing-
personal computer

elefctor february 1981

personal computer.

Now available for the ZX80...

before you can buy cassette-based software
using the full 1 6K-BYTE RAM . So keep an eye
on the personal computer magazines-and
brush up yourchess perhaps!

The RAM pack simply plugs into the
existing expansion port on the rear of the
ZX80 No wires, no soldering. It’s a matter of
seconds and you don't need another power
supply. You can only add one RAM pack to
yourZX80-but with 16K-BYTES who could

Massive add-on memory. Only £49.95.

The new 16K-BYTE RAM pack is a complete
module designed to provide you -and your
SinclairZX80- with massive add-on memory.
\bu can use it for those really long and
complex programs -or as a personal
database. (Yet it can cost as little as half the
price of competitive add-on memory for
other computers.)

For example, you could write an
interactive or ’conversational’ program to
show people what your ZX80 can do. With
16K-BYTES of RAM. they could be talking to
your computer for hours!

Or you can store a mass of data -perhaps
in a fairly simple program -such as a name
and address list, or a telephone directory.

And by linking a number of separate
programs together into one giant, but
modular, program, you can achieve the same
effect as loading several programs at once.

We’re also confident that it won’t be long

How to order

Demand for the ZX80 exceeds all other
personal computers put together! So use the
coupon to order today for the earliest
possible delivery. All orders will be
despatched in strict rotation. We’ll
acknowledge each order by return, and tell
you exactly when your ZX80 will be delivered.
If you choose not to wait, you can cancel
your order immediately, and your money will
be refunded at once. Again, of course, you
may return your ZX80 as received within
14 days for a full refund. We want you to be
satisfied beyond all doubt -and we have no
doubt that you will be.

To: Science ot Cambridge, FREEPOST 7. Cambridge CB2 1 YY.

Remember all prices shown include VAT. postage and packing

Please send r

rZX80 Personal Con

Ready-assembled SmclairZXBOPersonal Compuler(s)
Price includes ZX80 BASIC manual and mams adaplor
Mains Adaptoris) (600 mA al 9V DC nominal unregulated)

ZXSOkit

TOTAL:

Name: Mr/Mrs/Miss

Science of Cambridge Ltd.

6 Kings Parade, Cambridge. Cambs., CB2 1SN.
Tel: 0223311488.

jary 1981 - 2-01

Digital video

TV technology

Digital technology has been part and
parcel of television equipment, especially
in remote control devices, for quite
some time. Nowadays, however, even
video signals are being processed by
digital means. At the Valvo semiconduc-
tor plant in Hamburg, research led to
the production of an integrated ana-
logue to digital converter which includes
a digital filter and PAL decoder for
colour television. This system can also
be used to eliminate 'ghosting' caused
by multipath reflections.

Semiconductor manufacturers endeav-
our to integrate as many circuit func-
tions as possible of a particular device
onto a single silicon 'chip'. Conventional
signal processing technology in tele-
vision sets is principally analogue, even
though bipolar integrated circuits are
now being incorporated to a large
extent. Further integration as far as
analogue technology is concerned will
be somewhat limited. For instance, the
single chip colour decoder already con-
tains just about everything that could
possibly be integrated. The remaining
external components, such as piezo-
ceramic delay lines and filters, are as yet
'unintegratable' and in any case, they
require extensive alignment. The logical
solution lies therefore in digital tech-
niques. It is hoped that digitisation will
achieve the following objectives:

1. Further integration

2. Systems which will not require
(manual) adjustment and which can

be controlled by microprocessors, en-
abling them to be adapted for various
designs and signal sources by means of
software.

3. Improved picture quality without
long term drift and new technical

possibilities such as the elimination of
ghosting.

The road to digitisation is by no means
easy. Because of the high (6 MHz) signal
frequencies involved, total integration
of a digital video system like the colour
decoder can only be achieved with the
highest technology available.

Concepts

The colour decoder can be considered
to be the key towards digital video
systems. The components that are re-
quired are:

— fast analogue to digital converters

— digital filters

— microcomputers

Fortunately, all of these components
can be made up from 'standard' (bi-
polar) parts. However, by using VLSI
(Very Large Scale Integration) MOS
techniques it has now become possible
to develop a fast analogue to digital
converter having the following charac-
teristics:

— active 1C surface approximately
2 mm 1

— clock rate for A/D conversion
>20 MHz

— device dissipation < 300 mW

The internal view of the VLSI A/D con-
verter is shown in figure 1 . Until very
recently similar high speed analogue to

digital converters could only be manu-
factured using bipolar technology.
Rapid developments in VLSI tech-
nology, however, enabled MOS circuits
to operate so fast that even video signals
can now be processed by a single chip.
Following the development of the fast

A/D converter came research into using H

the same techniques for a digital PAL

decoder. Initially tests were carried out H

with a multitude of standard Schottky "MiitlfP RR

TTL devices. Figure 2 shows the block

diagram of how five double sized euro H

cards were arranged to produce the H

required result. One of these eurocards H

is shown in figure 3. It contains a digital H

luminance filter operating at a fre jflB 1

quency of four times that of the sub-
carrier (17.73 MHz). The next step in

the process is illustrated in figure 4. * - -

Here the entire luminance filter has flH|

been compressed into a single chip with H|_

a surface area of a few square milli- ^

metres.

The experimental set-up included both Figure 3. One of the double
programmable digital luminance and
chrominance filters and a digital delay
line for the PAL decoder. The system
was connected to a colour television
which had been especially modified for
the purpose and the excellent properties
of the digital decoder were demon-
strated to the technical press. The pro-
grammable characteristics of the lumi-
nance filter (see figure 5) enabled the
picture to be brought into much better
focus (due to aperture correction). It
also proved that an 8-bit definition for
the analogue to digital conversion of the
composite video signal was more than
adequate. Even if this were reduced to
five bits it would hardly affect the pic-
ture quality.

It was also found that 'ghosting' due to
multipath reflections could be impress-
ively reduced by means of phase equal-
isation. The ghosts are literally driven
from the screen! This solves a problem
that was previously thought to be in-
surmountable.

Now that video is 'going digital' the
trend should catch on in other fields as
well. Digital records and tape recorders
(audio) have been promised for the near
future; next in line are digital stereo
decoders, preamplifiers, duplexers and
equalisers. The digital amplifier already

The same luminance filter after integration. The surface area i

Figure 5. The amplitude curve of the digital luminar
programmable characteristics of the luminance filter
provide much better focusing of the picture (apertui

tt suitt i tunui i ntinn i :

1981 - 2-03

audio

Although this power meter was initially
designed as an accessory for the 200 W
power amplifier described in the
January 1981 edition of Elektor, it can
also be used with virtually any other
amplifier with one proviso ... as the
specification of the above amplifier is
200 W into 4 £2, this power meter has
been designed to be compatible with
4 £2 loudspeakers. In addition, the
unit has two ranges: 0 ... 50 W and
0 . . . 200 W. The easiest way to under-
stand how the power meter works is
to examine the circuit diagram.

ing coil meter, a good quality power
meter with a linear scale can be
realised.

As mentioned previously, the power
meter has been equipped with two
ranges: one for high power — 200 W,
and one for low(er) power — 50 W. This
is taken care of by the two potential
divider networks, R1/R2/R4 and
R3/P1/R4, and the range switch SI.
Neither network has to be a precision
type as the circuit is calibrated by
means of the two preset potentio-
meters PI and P2. On the other hand.

power meter

There now appears to be a great
demand for a visual indication
of the amount of output power
available from a particular
amplifier. This is especially true if
a great deal of time and money
has been spent on building such an
amplifier. The circuit described
here is intended primarily for use
with the 200 watt power amplifier
design published in last month's
issue of Elektor, however, there is
no reason why it could not be
used with other amplifiers. The
unit utilises a moving coil meter
to give a linear indication of the
power level being fed to the
loudspeakers.

Circuit

As can be seen from figure 1 , the circuit
diagram of the audio power meter could
hardly be called complex. Very few
components are needed to construct the
complete unit. The circuit would even
be simpler if the requirement was for
a logarithmic rather than a linear

How does it work? The majority of
readers will know that the output of an
amplifier is proportional to the square
of the output voltage:

p= u eff 2

Rl

Thus, by merely measuring the output
voltage, a power meter with a logarith-
mic scale can be obtained. In this par-
ticular instance use is made of the l/V
characteristics of a germanium diode by
feeding the amplifier output signal to
a bridge rectifier via a potential divider.
Provided the voltage across the diode
bridge remains below about 1.4 V,
the l/V curve will be exponential. This
means that the current flowing through
the diodes will be proportional to the
square of the output voltage (U e ff 2 ).
By monitoring this current with a mov-

the internal resistance of the meter is
fairly critical and should be somewhere
in the region of 100 . . . 1 80 £2.

Calibration

The output of the power amplifier is
loaded with a 4 £2 resistor — not a loud-
speaker. The power meter is then con-
nected in parallel with this resistor. A
1 kHz signal is then fed to the input of
the amplifier and the output voltage
monitored with a multimeter (30 V AC
or greater).

With the range switch in the '200 W'
position the volume of the amplifier
can be turned up slowly. It is important
to keep an eye on the meter while doing
this so that if the needle looks like bend-
ing itself around the end stop, P2 can
be adjusted to compensate. When the
multimeter indicates an output voltage
of 28.3 V, the power output will be
exactly 200 W. Potentiometer P2 is
then adjusted to give a full scale read-
ing on the power meter. The amplifier
volume is then turned down until a
reading of 14.1 V is indicated on the
multimeter. Switch SI is then placed in
the '50 W' position and PI adjusted to
give a full scale reading once more. H

D1 ... D4 B AA1 19

Figure 1. The potential divider! R1/R2/R4 and R3/P1/R4 ensure that the voltage across the
bridge rectifier never exceeds about 1 V. The diodes will then operate in the non-linear area of
their l/V curves so that the current through them will be exponential in relation to the voltage
across the bridge.

2-04 - elektor «e

sry 1981

noise reduction

silence is golden

Noise reduction systems, the electronic kind, have been in the news off
and on for about ten years, since the compact cassette became popular
in fact. Technology moves on however and recent developments in this
field have resulted in new and improved systems. One of the better
ones is the High Com from the giant Telefunken company. This system,
together with printed circuit boards, will be featured by Elektor as a
constructional project in the very near future. This article proposes to
take a look at the various aspects involved in noise suppression and to
compare the foremost systems currently available.

The market for noise reduction systems
has seen numerous changes in recent
years. It all began to happen in 1966
with the arrival of the professional
Dolby A system which soon became
part and parcel of every self-respecting
studio. The arrival of the compact
cassette and the ensuing argument of
tape versus cassette quality, made a
ready market for a simplified Dolby
noise reduction circuit. This was of
course the Dolby B system which fitted
the bill very nicely. Philips, the inven-
tors of the cassette itself, followed up
shortly after with their DNL (Dynamic
Noise Limiter). It was not too difficult
to see which of the two in existence
would be suitable for home construc-
tion. On the one hand, the Dolby B
system was fairly complicated, requiring
a great deal of skill and calibration on
the part of the builder (even if the ICs
were available). On the other hand, the
Philips DNL was not as effective as the
Dolby but was very much simpler and
was offered free to the world ... no
licences required. This of course pre-
sented a far better prospect to the home
constructor.

In retrospect the DNL could not survive
commercially. The world-wide appli-
cation of the Dolby B system in top
quality cassette decks made it the indus-
try standard and this, together with the
availability of 'Dolby-ised' cassettes,
allowed Dolby to monopolise the field
of noise suppression for a very long

Recent advances in noise
suppression

Japanese audio manufacturers in par-
ticular have been very busy trying to
develope their own noise reduction
systems. This is not so surprising since
the use of the Dolby incurs very high
licensing fees. With the production
quantities involved in the Japanese
export market, it would be very much
cheaper for them to have their own
system. Of course its specifications
would have to be an improvement to
make any significant headway.

The first manufacturer to follow in the
wake of Dolby was JVC with the excel-
lent ANRS and later the SUPER ANRS
systems. Other Makers appeared on the
scene, such as Toshiba with the ADRES,
the DBX-1 1-1 24 from DBX, the
PLUS-N-55 from Sanyo and the Phase
Linear 1000, an auto-correlator system.
Understandably, Dolby did not remain
idle either, they produced the Dolby
HX.

One name is missing from the above col-
lection and that is the High Com by
Telefunken. This is likely to become
Dolby's number one rival, as more and
more manufacturers are selecting it for
their cassette decks.

There are so many systems available
nowadays, that it is difficult to see the
field for grass, not to mention the less

>r february 1981 -2-05

known makes like Burwen and numer-
ous professional filters. The hobbyist is
of course Only interested in one aspect:
which system is the best to build? Un-
nfortunately, most manufacturers either
refuse to disclose their technical recipes
or ask exorbitant amounts for them.
Elektor decided on Telefunken's High
Com after intensive and extensive
research into the basic principles in
which the reader is now invited to
partipate.

Historic facts

Noise suppression systems all have one
thing in common: they are designed to
electronically eliminate noise as far as
possible. This became paramount once
the compact cassette was invented. This
cheap form of sound recording so small
in size and easy to use, soon became
immensly popular, but the standard low
tape speed of 4,75 cm per second
caused quite a few problems, not the
least of which was NOISE!

Figure 1. What the compander consists of and what it does to the dynamic range.

Figure 2. The block diagram of the DNL system. This will only work during playback.

Is there any escape?

The noise we're talking about is peculiar
to magnetic tape and so very difficult to
get rid of. The tape consists of a carrier
covered in a thin layer of magnetised
particles (Fe02, Cr0 2 or Fe). In the
process of recording an audio signal on
the tape the particles are magnetised by
the recording head. Since the particles
are not evenly distributed on the tape
and therefore not equally magnetised,
soft passages feature a high noise level
that is especially audible at high fre-
quencies.

There are two ways in which to reduce
noise: by speeding up the tape or the
use of higher modulation. The first
method is out of the question, because
the cassette is run at a standard speed.
(Nowadays there are a few manufac-
turers who provide tape decks with a
second tape speed of 9.5 cm per sec-
ond.) This leaves the second option as
the only possibility: make sure there are
t no soft passages on the tape.

Compressor + expander =
compander

The first serious attempt to suppress
noise by means of electronic circuits is
attributed to Dolby. Basically, the
system works as any other of its kind.
The block diagram in figure 1 gives a
general idea of what happens within a
complete recording and playback chan-
nel. During recording the dynamic range
of the signal is compressed and expanded
again when the tape is played back. The
term dynamic range is very important
in this respect, extending from the
loudest to the softest signal to be re-
corded. Peak modulation is usually
indicated as OdB. Thus, the lowest

signal to go on tape (cassette) will be
about 56 dB below that level. Con-
sidering the dynamic range of a high
quality record is 65 dB, cassette
recording means a loss of almost 10 dB
to start with. When a record is taped on
a cassette deck the difference between
them is very noticeable. The wide
dynamic range so characteristic of
records is largely lost.

The drawing in figure 1 gives a sim-
plified version of a noise reduction
system. Below the block diagram the
drawing shows what happens to the
dynamic range during various stages.
During recording the level of the input
signal is lowered to a value the tape can
accept (including a certain safety thres-
hold). During playback the dynamic
range is 'retranslated' into the original
value by the expander. This maintains
the noise level beneath the lowest signal
level recorded, so that (theoretically)
it should no longer be audible. The
entire noise suppression system is called

a compander (a combination of COM-
pressor and exPANDER).

Controlled filters

At this point a small digression is called
for. There is another form of noise
suppression which only works during
playback and only eliminates noise
when it is a real nuisance: at high
frequencies.

This type does not fall into the com-
pander category, as it is a kind of con-
trolled low-pass filter (where the slope
of the curve can be determined). Pro-
fessionals regularly use 'normal' low-
pass filters. Unfortunately, however,
they all have the same drawback in that
they affect the original signal. In other
words, what is needed is a system that
removes the noise without influencing
the original sound.

A typical example of a controlled filter
is the well-known DNL unit (see the

— ©

block diagram in figure 2). The input
signal is first split up into two com-
ponents, Ui and U : . Signal Ui proceeds
directly to an adder circuit at the out-
put. U 2 , on the other hand, passes
through a high-pass filter and is then
amplified. After this, the signal is
reduced again by a dynamic attenuator,
where the attenuation achieved depends
on the level of the higher frequencies
in the input signal. The entire circuit *
is preset so that the signals (Ui and U 2 )
are equally large for signals above 4 kHz
with a power 38 dB or more below the ,
reference level, but at the same time
they will be in phase opposition. In the
adder circuit they cancel each other out.
Thus, the noise or rather, high frequency
suppression will only be successful with-
in that range. Low and high frequencies
with an amplitude greater than -38 dB
will remain intact. The signal/noise ratio
can be improved by about 3 dB with the
aid of DNL.

The Phase Linear 1000 is a very elabor-
ate system and certainly worth men-
tioning. This uses digital means to cut
the noise level down. Figure 3 contains
the block digram representing this sys-
tem. It involves an auto correlator which
proces amazing results especially when

1981 - 2-07

combined with Dolby-B and the 'down-
ward expander' of the device. The input
signal is divided into nine frequency
bands and each one is examined for
interference. The auto correlator 'looks
at' the input signal, finds out which fre-
quencies it contains and then connects
the signal through to the output by way
of the corresponding band-pass filters
according to the signal's frequency
distribution and level. A wonderful
system but far from cheap. Its main
advantage is that it can be used with
every type of program material.

Companders

Most noise suppression systems are in
fact companders. This is largely due to

the fact that a noise reduction system
is essential in a cassette recorder and a
compander leads to good results with-
out being complicated as a circuit.

Let's take a closer look at the main
prototypes: DBX, Dolby and Telcom
(the professional High Com).

Dolby was the first to come up with the
idea of splitting the signal up into differ-
ent frequency bands and to feed each
one to its own control circuit so that
each band can be compressed separ-
ately. During playback the signal is
again divided into bands and each is
attenuated according ot its level.

Figure 4 shows the Dolby A system.
The audio spectrum is split up into four
bands, each with its own control
system. A low-pass filter at the input
makes sure that HF signals cannot ad-

elektor february

versely affect the control system. Next
the signal passes through an adder and
subtractor circuit and is then split into
four bands, one of over 9 kHz, over
3 kHz, between 80 Hz and 3 kHz and
below 80 Hz. Every band has a voltage
controlled amplifier (VCA) where
amplification will depend on the aver-
age signal level in the band concerned.
The four outputs of the VCAs are added
together and during the recording they
are added to the original signal and sub-
tracted during playback. This profes-
sional Dolby system can suppress up to
10 . . . 12 dB of noise, which is fine.

The second brand on the list is the DBX
version. Its block diagram (see figure 5)
looks very straight forward compared to
the Dolby. During the recording the
signal first passes through a band-pass

2-08 - elektor february 1981

qcH ss

s

g

Mi

ms

Hi

Si

g

m

ms

Hi

SI

si

g

jBj

n

Hi

B

jBj

HI

mi

s

Figure 6. The block digr

af Telefunken's Telcom systems.

filter (bandwidth 22 Hz ... 32 kHz)
which again prevents undesirable signals
from affecting the compression system.
The next stage amplifies the high fre-
quencies by 12 dB (pre-emphasis). This,
in combination with the de-emphasis
during playback, reduces modulation
noise at high frequencies. The VCA to
follow this section compresses the signal
by a factor of two. The control signal
for the VCA is derived from the output
signal that will first have to be filtered
once more (band-pass filter 11Hz..
, . 22 kHz) to remove any interference
from the tape. After this, there is a de-
emphasis stage to compensate the pre-
emphasis that occurred earlier and an
effective value detector then derives the
control signal from this for the VCA.
During playback the same circuits are
used as during the recording with the
exception of the input filter. Only the
configuration of the various blocks will
be different. The input signal passes
through the band-pass filter and after
de-emphasis returns to the effective
value detector which the VCA controls
in such a way that it expands the signal
by a factor of two. All that is needed
then is a de-emphasis to return the high
frequencies to their original level. As a
result, the signal/noise ratio is improved
by as much as 30 dB!

Finally, it is time to deal with High
Corn's big brother. This is called the
Telcom and on the face of it it would
appear to be a mixture of the DBX and
Dolby systems, as on the one hand it
uses several bands like the Dolby and
on the other it has a fixed compression/

expansion ratio like the DBX. The block
diagram in figure 6 is very similar to the
Dolby system. The input signal proceeds
to the output via a band-pass filter and
an adder and subtractor circuit (for
recording and playback, respectively).
After the input filter the signal is split
up into four bands. These are, however,
very differently distributed from those
in the Dolby. Each filter is followed by
both a VCA and another band-pass
filter. All the filters have a descending
slope of 6 dB per octave and the peaks
have been chosen to partly overlap each
other. The second series of filters are
again followed by VCAs, behind which
peak level detectors detect the control
signals for the VCAs. The control
system thus obtained is fairly complex
due to the filter combination and has
the advantage that the system does not
produce so much 'pumping' — which
causes other systems a fair amount of
trouble. The output signals of the first
series of VCAs are added together and
the result is sent to the adder and sub-
tractor circuit. The circuit is preset at a
fixed 1.5:1 compression/expansion
ratio. This remains linear within an
extensive dynamic range so that there
is no need for calibration. About 25 dB
is gained with respect to the signal/noise

In practice, the Telcom proves to be a
successful combination of the advan-
tages of both the DBX and the Dolby
systems. Although the DBX suppresses
noise very well, the system tends to be
rather audible when readjusting. In
comparison, the Dolby's noise sup-

pression is but mediocre, even though
its other results are excellent. Thus it
can be concluded that the Telcom with
its first class noise suppression and
quality performance is the best choice.
All the above professional systems have
since been developed for domestic
purposes with equally good perform-
ances. The only system suitable for
homeconstruction, however, is the High
Com. Thus, before going into detail on
how to build the Elektor noise sup-
pressor, let's find out what the High
Com consists of.

The High Com

Readers who think that the High Com is
merely a simplified version of the
Telcom, have got it all wrong. Sur-
prisingly, it has certain advantages when
compared to the latter. Obviously, the
circuit had to be made less complex. In
fact practically the entire compander
fits into a single 1C and this allows the
construction to be so much easier.

The High Com system is a 'broad band
compander', meaning that it works
throughout the entire audio band
instead of starting at 500 Hz like the
Dolby B. This has the advantage that a
'broad band compander' is insensitive
to the frequency characteristic and level
setting of the recording chain. In other
words, since the whole frequency range
is dealt with in the same way, an in-
correct level setting does not affect the
frequency response (within the permiss-
ible range of levels).

Figure 7 shows the block diagram of the

elektor febr

1981 - 2-09

High Com system. The blocks indicated
as A are identical and consist of a stage
to boost the high frequency. Behind this
there is a voltage controlled amplifier.
Block B is the expander and has the
opposite transfer function as that of the
A blocks. In addition, a kind of de-
emphasis (block C) and pre-emphasis
(blocks D) takes place. Finally, there are
two rectifiers (E) which produce the
control voltages for the various VCAs.
The signal passes through the circuit as
follows. First the high frequencies of
the input signal are amplified. Then the
output signal derived from the VCA
behind this is used to generate the con-
trol voltage. This requires an op-amp,
a VCA, a pre-emphasis and a rectifier.
Before the compressed signal reaches
the recorder, it will initially passthrough
a de-emphasis phase. When the cassette
is played, the opposite happens. First
there is pre-emphasis, then the control
voltage is derived with the aid of a cir-
cuit similar to the one in the com-
pressor section and finally the signal is
expanded in block B to its original

De-emphasis is applied to the recording
to prevent the tape from being over-
modulated at high frequencies. The cir-
cuit is designed so that a 10 kHz signal
will be amplified when it is more than
12dB below the top modulation level,
but will be attenuated when it is be-
tween -12 and 0. Pre-emphasis has the
exact opposite effect.

Figure 8 shows a graph of the com-
pression and expansion curves of the
High Com system. It can be seen to
what extent a signal having a certain
frequency and power (in terms of dB) is
compressed and expanded. You would
think a broad band compander's curves
would be the same for all frequencies,
nevertheless this is not the case since the
high frequencies are boosted during
compression.

There's no doubt about it, the High
Com gives excellent results. Using a
good quality cassette the signal/noise
ratio is improved by 20 dB - you have
to hear it to believe it!!

A more detailed description of the High
Com will be given at a later date in an
article devoted to the Elektor com-
pander, a high quality noise suppression
system. Until then keep the volume
down ... M

Miiia i'

Figure 7. This is the block diagram of the compressor and expander section of the High Com

8

Figure 8. A graph showing the compression and expansion curves of the High Com system
at different frequencies. 0 dB corresponds to peak modulation.

2-10 — elektor february 1981

J. Meyer

process-timer

The time factor is of vital importance in the development and printing
in photography. Each process, development, fixing and rinsing,
require different time periods depending on the chemicals and the type
of paper used. In short, the whole process can become something of a
hit and miss affair, especially if there are a pile of prints to produce.
Programmable exposure timers are available of course but these generally
give just one time period when several are really needed. The process
timer to be described here, however, is provided with a scale division
allowing each time phase of the complete process to be monitored.

The timer will indicate the different time periods for the development,
stop bath, fixing and rinsing phases and in the correct order. The time
intervals are determined by the use of 'process’ cards which are
calibrated according to the film, paper and/or chemicals used. Thus for
each combination of materials, a specific card will be needed.

The application of the process timer does not have to end at
photography. Any process that is divided into a series of timed events
will find a use for this type of timer.

Electronically, the process timer is not
at all exceptional and uses only a hand-
ful of readily available CMOS chips.
However, the method in which the parts
have been put to use is novel. The pro-
cess timer 'communicates' its activities
by means of a row of LEDs in conjunc-
tion with a small process card (see
photo 1). Pressing the start button will
cause the first LED to light. After
30 seconds this will 'jump' to the
second LED, a further interval of
30 seconds will cause the third LED to
light and so on down the entire row of
LEDs. A 'process' card, on which the
various process time periods are cali-
brated in 30 second steps, is placed
along the LED row. The lighted LED
will now indicate on the card how far
the process has progressed. If the pro-
cess is to be temporarily halted, the
interval switch can be operated and
the timer will stop and await further
instructions.

Take a practical example. It is required
to develop a photograph and the process
is indicated in four phases on the card.
Other important factors are also in-
cluded such as the temperature and type
of chemicals and paper being used. The
card is now placed alongside the row of
LEDs. After exposure, the photographic
paper is placed in the developing tray
and the start button is pressed. The
moment a LED indicates the end of the
developing time period, the paper is
taken out of the developer and placed
in the stop bath. The timer will now
continue to indicate 30 second intervals
through this phase. It is easy to see that
it is merely necessary to watch the
LEDs to get a fairly accurate indication
of the position of any phase in the
complete development cycle.

If a different brand of paper and/or
chemicals is to be used, a suitable
process card can be designed (with
'experience' built in). There is now no
longer any need for guess work since a
practical 'experiment' can be made
repeatable by marking the results
directly onto the process card.

A further card indicating the cost of
telephone calls could be used to allevi-
ate that sinking feeling commonly felt „
in the wallet when the telephone bill

The circuit diagram
The timer has been designed with a
battery supply in mind and for this
reason CMOS ICs have been used. It
will be obvious by this time that a shift
register forms the heart of the elec-
tronics involved, in fact the 4015
CMOS 1C. The LEDs in the display are
connected directly to the outputs of the
four registers, ICsl ... 4. To reduce
power consumption to a minimum the
current through the LEDs is switched,
by transistor T1, at a 2 Hz frequency
and with a 50% duty cycle.

The clock generator is formed with
gates N2 and N3 and the clock fre-

quency is divided by IC6. Pressing the
start button (S3) will clear the counter
and set the flipflop IC5 causing the LED
D1 to flash. About 15 seconds after the
start button is released the Q12 output
of IC6 will go high. This will be the first
clock pulse to the register and during its
positive transition a '1' will be entered
into IC1. Every 30 seconds thereafter,
a clock pulse will appear at the Q1 2 out-
put of IC6 and the ' 1 together with the
flashing LED, will move along the LED
display at the snail's pace of one 'jump'
per 30 seconds.

When designing the process cards it will
have to be taken into account that the
first LED only flashes for 15 seconds,
whereas the others light for 30 seconds.

The initial 15 seconds can also be used
to enable both hands to be free after
* pressing the start button, so that any
last minute jobs that need to be dealt
with before the process starts can be
, completed.

It was mentioned previously that
switching the LED current on and off
saves a considerable amount of energy.

There is, however, another reason for
this. When the outputs of the shift
register sink the LED current, their
voltage level will drop. In that case an
input connected to one of the outputs
will no longer recognise the level as
being logic 1. To remedy this, the phase
shift network R35/C1 ensures that
transistor T1 stops conducting when a
1 level is being shifted. The outputs of
the shift register will not be loaded at
that moment so that everything will photi
proceed as planned. incor

Whenever the process is to be inter-

rupted this is merely a question of
switching S2. The clock generator then
stops and transistor T1 continues to

is glued on top of the lid. The shortest
two edges can be turned over to allow
the process card to be slipped in.

It is best to use LEDs in two colours,

say, red and green, and mount them

alternately on the board, as this makes made from pieces of white cardboard, that the first LED will flash for pre-
it easier to see the light 'jump'. The Once the scale division has been in- cisely 15 seconds. The next LEDs will

it easier to see the light 'jump'. The Once the scale division he
board is designed for rectangular LEDs eluded, they can be covered
but other types are equally suitable, five layer of sellotape.
provided their width does not exceed The process timer can be cali
2.54 mm. The process cards can be an ordinary watch. Pot PI is preset

is been in- cisely 15 seconds. The next LEDs will
in a protec- then each light for 30 seconds. The
clock generator may also be preset at a
ibrated with different frequency to divide the pro-
is preset so cess into larger or smaller steps. M

high voltage from 723

up to 60 V with an 1C

Figure 1 gives a look inside the 723 1C.
It contains a temperature compensated
and relatively noise-free reference vol-
tage source U re f. From this a current
of up to 15 mA can be derived. A
correction amplifier controls a series
transistor which provides the output
voltage. In addition, there is a current
limiting transistor enabling a highly
stable and 'short' proof power supply
to be constructed with only a few exter-
nal components.

To see how it works, let's see what
happens at a stabilised 5 V (see
figure 2). A voltage of 5 V divided by
R1 and R2 is at the non-inverting
input.

This situation is called a 'floating
regulator' since the auxiliary voltage
literally 'floats' above the actual stabi-
lised output voltage. Figure 3 illustrates
this particular method. The auxiliary
supply U2 serves to drive the 1C, its
negative pole is connected to the
positive stabilised output voltage. The
723 1C regulates the drive current for
the external series transistor. By regu-
lating the drive current in parallel,
output voltages can be preset at 0 V.
By way of P the correction amplifier
measures the output voltage so that
this can be preset by P.

Figure 4 shows the finished product, a
power supply with an output voltage

lii^h voltage from 723

Hobbyists faced with having to
build a power supply that can
produce output voltages of 40 V
and higher know that the only
way to do this is to use discrete
semiconductors. After all, the
maximum input voltages of most
integrated voltage regulators are
nearly always too low. Even the
best known voltage regulator 1C,
the 723, has a peak input of only
40 V and produces an output of
not more than 37 V (see table 1 ).
As it happens, this particular 1C
can make up for its own handicap.
How? Read on . . .

The attenuation of this nominal value
is measured at the inverting input via
R3 and then calibrated by the correc-
tion amplifier automatically.

In order to stabilise voltages of more
than 40 V, the 1C will require a separate
auxiliary voltage to provide the supply.

that can be preset anywhere within
the 0 V ... 60 V range and having a
current capacity of 1 A. At the non-
inverting input (pin 5) there is the refe-
rence voltage now divided by R2
and R3. The slider of PI is connected
to the inverting input (pin 4) of IC1.

Figure 1 . The contents of the 723. It conteins the ective components required to form the basis
of a highly stable power supply.

2

5 V/100 mA power supply. This simple circuit illustrates clearly how the 1C

high voltage from 723

elektor febr

ary 1981 - 2-15

The correction amplifier will now com-
pare the voltage at the slider of PI to
that at pin 5 and with the internal
transistor connected as a parallel regu-
lator it will control the base drive
current of T1 across R5 and D5 in
such a way that the two voltages
become equal. If the one at pin 4 is
too high so that the output voltage
of the power supply is too low, the

base drive current of T1 will rise
bringing the output voltage back to its
correct value. With the given component
values it is possible to preset the out-
put from OV to 60 V with PI. As the
resistance of PI has a 10% tolerance
P2 in included to pinpoint the peak
output at exactly 60 V. The internal
current limiter transistor can however
not be used in this circuit, since it

would effect the exact opposite result.
In other words, the output voltage
would rise instead of drop!

For this reason T2 takes care of the
current limitation. Table 2 gives an indi-
cation of the figures which can be
obtained with this circuit. The advan-
tage of a floating regulator is that the
maximum output voltage is now only
dependent on the U ceo of the external
series transistor and so the formulae for
the values of R1 and R9are as follows:

R1 «P2
PI = 10 kfl
Umax > 40 V
R 1 = 5.9 PI
R9 = PI x 1.2

R8 determines the maximum output
current, therefore

These formulae enable the circuit to
cope with voltages of several kV de-
pending on components used such as
T1 , D9, D10, etc.

When currents rise above 1 A, an eye
will have to be kept on the dissipation
of T1. For currents below 3 A the cir-
cuit in figure 5 can substitute T1. In this
case, however, R8 will have to be re-
duced by 0.22 Q per 4 W. M

Figure 3. The block diagram of a floating regulator used with the 723 1C.

junior's droning up !

a survey of possible extensions

Since Elektor published the article
on the Junior Computer
(May 1980) the editorial office
has been inundated with queries
about expansion possibilities.
Basically, it all boils down to: how
can the Junior Computer be
expanded and to what extent?

As would be expected, there are a num-
ber of different possibilities for 'devel-
oping' the Junior Computer, however to
say 'the sky's the limit' would hardly be
wise. When thinking about possible
expansions for the Junior Computer it is
best to be selective and not concern our-
selves with equipment which has no real
purpose. For this reason a 'listing' of the
future hardware and software extensions
has been drawn up and is given below.
Obviously, this can be no more than a
brief summary at this stage, but full
details will be provided in the forth-
coming publication of Books 2 and 3.

1. Interface card

A cassette interface seems to be the
number one requirement as far as hard-
ware is concerned. This has been in-
cluded on the interface card and has
provision for two separate cassette
recorders. It is also compatible with the
KIM microcomputer. The cassette inter-
face can be controlled by means of
either the hexadecimal keyboard or an
ASCI I keyboard (in the latter case there
are a number of operational possi-
bilities).

The interface card also contains 1 k of
RAM (2x2114), user input/output
(6522) and a standard RS232 interface.
In addition, there is provision for two
1C sockets on the board which can be
used for further memory expansion.
One of the following memory devices
can be inserted into each of the two
sockets: 2708 (1 k EPROM), 2716 (2 k
EPROM) or 8114 (1 k RAM). This adds
up to a possible 3 ... 5 k of extra
memory.

2. Memory extension

An article describing the RAM/EPROM
card was published in Elektor number
65 (September 1980) and in the follow-
ing issue (number 66, October 1980) an
explanation of how to connect it to the
Junior Computer was given. We realise
that the price of 2732 EPROMs may
well exceed the budget of some of
our readers and so we are currently
examining ideas for developing a less
expensive version — no promises, mind!

3. Hardware

Various peripheral devices can be con-
nected to the computer such as a video
interface and ASCII keyboard (the
Elekterminal) and a printer. As men-
tioned previously, the forthcoming
books will explain exactly how these
peripherals can be added.

4. EPROM programmer

It's all very well developing programs
and storing them on cassette, but cer-
tain routines are better stored perma-
nently in system memory. For this
reason, an EPROM programmer is
currently being developed which will be
suitable for 2708, 2716 and 2732
devices, including their derivatives such
as those with JEDEC pinning. The pro-
grammer will consist of a basic unit with
'plug in' modules for the different
device types.

5. Firmware

Bearing point 4 in mind, comprehensive
editor, assembler and disassembler rou-
tines have been developed for use with
an ASCII keyboard (Elekterminal) and a
printer. These routines will enable you
to develop, debug and list programs
quickly and efficiently.

6. Suggestions

A host of items are still the subject of
discussion. Any useful suggestions that
readers may have will be more than
welcome. For example, would you like
to be able to program your Junior
Computer in a high level language? If so,
which one? BASIC? Extended or Tiny?
Broad Scots or Scouse? Or would you
prefer to jump in at the deep end with

Pascal? How about a floppy disc and
graphics? Send your answers on a post-
card please to ... . No seriously, if you
have any ideas please let us know (we
are not yet capable of reading minds!!).

7. Software: user programs

Along the same lines, what sort of
programs do you want to run on your
Junior Computer? Games? Business?
Accounts? There are far more
interesting possibilities than digital
clocks and reaction timers! Again, and
even more important, have you any
programs? If perchance you have
written any interesting programs, don't
be shy, send them to our editorial staff.
They may well prove useful in helping
out fellow Junior Computer operators
(or even ourselves!) and at the same
time you can bring your 'output' into
the limelight by having it published in
Elektor.

For those who can't wait . . .

We accept the fact that certain members
of our readership are somewhat anxious
for the various extension possibilities to
be published as soon as possible. How-
ever, it may well be an idea to bear
in mind the following points: By
publishing the details of the Junior
Computer project Elektor hoped to
interest a large number of potential
computer enthusiasts who merely re-
quired a bit of encouragement (to-
gether with equipment they could
afford!). The Junior Computer books
are therefore necessarily tailored to suit
their tastes and requirements. Those of
you who were already working with
computers are bound to grow a little
impatient at the step-by-step methods
employed.

Another aspect worth considering is
that Elektor does have a magazine to
publish which contains various topics
and projects that all require technical
research. The neat double-sided main
computer board and the interface card
both required a large amount of time
and effort to develop.

Think of it this way: when you go out
to have a meal and a good time, you
don't just pop around to the local
'chippy', you go to a proper restaurant.
Bear with us, it will all be well worth
waiting for! M

1

Blektor february 1981 - 2-17

the finishing touch to
the Elektor vocoder

On the face of it, the detector may seem
superfluous. However, when the block
diagram of the complete vocoder in
figure 1 is considered and the proposed
additions are momentarily forgotten,
their necessity will be readily apparent.
In the upper section the speech signal is
divided and split into control voltages to
feed the VCA's in the synthesis section.
The VCA's are thus provided with an
input signal consisting of the carrier
signal chopped into identical bits and
pieces. Fair enough. In practice how-
ever, the synthesised result proves to be
less satisfactory than expected. The
fault lies with the carrier signal which is
far from ideal.

remedy for this was the inclusion of the
'high frequency blend' provided by PI 7
shown in the dotted area in figure 1.
Part of the 'high frequency’ in the
speech signal is taken from the high pass
filter in the analysis section and is
blended directly with the synthesised
result. This is precisely what Harald
Bode applies in his synthesiser.

In practice this solves quite a few
problems. For unvoiced signals to be
properly synthesised, however, a circuit
is required which can distinguish be-
tween the voiced and unvoiced sounds
during analysis. Professionals call such a
circuit a vo iced /unvoiced detector and it
is found in relatively few vocoders to

die voiced unvoiced
detector

This article presents the long awaited voiced/unvoiced detector and
brings the series on the Elektor vocoder to a conclusion. The detector,
combined with the noise generator, enables the unvoiced sounds
(s, f, etc.) to be synthesised with ease. It succesfully eliminates a minor
shortcoming in the vocoder that was intended as a short term
compromise.

Most synthesised signals happen to be
incomplete as far as their spectrum is
concerned. This means that unvoiced
sounds such as s, t, k and p do not come
through very well, in fact they are often
F. Visser inaudible. The simple and effective

date. The reason for this is largely due
to the fact that the components re-
quired are fairly complex and therefore
increase the price of the vocoder con-
siderably. Technically speaking, it is by
no means easy to design and this of
course also deters many manufacturers.
When it is combined with a noise gener-
ator a decent voiced/unvoiced detector
is a great improvement on the blending
trick mentioned earlier. The latter
would not work, for instance, whenever
speech is to be synthesised without an
original speech signal. In other words, a
microprocessor and a DA converter are

2-18 - etektor f

roiced/unvoiced detf

unable to generate a complete, artificial
speech spectrum. The detection system
described here can however do this. It
enables noise to be fed to all the syn-
thesis filters in the vocoder whenever
there are unvoiced sounds in the speech
signal. With the aid of control voltages
derived from the analysis section the
required 'colour' noise can be produced.
In addition, the detector is fast enough
to provide a very true-to-life synthesis
of the s, t, k and p sounds.

How does it work?

Whereas the practical construction is
rather complicated, the block diagram
of a voiced/unvoiced detector is fairly
straight forward. Figure 1 shows the
general principle. The speech signal is
fed to a suitable detection system that
can distinguish between the unvoiced
and voiced sounds. This detector oper-
ates a switching circuit which interrupts
the carrier signal in the event of un-
voiced sounds and then substitutes it
temporarily for the output signal of a
noise generator.

Clearly the detection system is at the
heart of the matter, but the little block
in the diagram hardly gives an indication
of its function. What does it do exactly?
Figure 2 illustrates the frequency ranges
which the detector 'examines' before
deciding whether the signal is voiced or
unvoiced. The mere fact that there are
many high frequencies in the speech
signal does not mean that the speech

signal is unvoiced at that moment. This
assumption is totally incorrect, as the
high frequencies measured may well be
part of a complex signal with a funda-
mental frequency that is so low that it
is a voiced signal after all. That is why
the detector also checks the low fre-
quency range (down to 600 Hz). If at
that moment the range does not include
a signal, or if the signal is much smaller
that its high frequency counterpart,
chances are the sound is indeed un-
voiced. Thus, two elements must be
incorporated in the detection system:

a high pass filter with a cut-off fre-
quency of about 2500 Hz and a low
pass filter with a turnover point at
about 600 Hz.

The voiced/unvoiced detector

The complete circuit diagram of the
detector is given in figure 3. Points A,
B and C of figures 3a and 3b are linked.
Roughly speaking (there is a little more
involved) the diagram in figure 3a con-
stitutes the detection system and that in
figure 3b the section drawn as a switch
in the block diagram. Both circuits are
mounted on a separate board. The noise
generator is incorporated on a third
board, but this will be dealt with later.
First let us look at figure 3 in further
detail. It can be seen that the speech
signal derived from the vocoder initially
reaches the buffer/amplifier A1 and is
then split into two signals, each passing
the filters mentioned above. The high
pass is constructed around A2 and A3
and the low pass around A4 and A5.
Their peak values are at 2500 Hz and
600 Hz, respectively. The two filter
sections have a slope of 24 dB per
octave to obtain the best possible separ-
ation. They are each followed by a
rectifier (A6 and A8) and by a 1 2 dB/
octave smoothing filter (A7 and A9).
The latter's turnover frequencies are
around 300 Hz for the high pass system
and 30 Hz for the low.

The rectified and calibrated output
signals are now fed to three amplifiers
or comparators (A10, All, A12) fol-

lowed by a number of logic gates. All that from the high pass filter. This if this is considered superfluous, the sec-

that need be said about these is that means, the output of All will remain tion around T4 and T5 can always be

they take care of the trigger signals that low causing that of gate N2 to be high omitted.

are required later on to feed the carrier and N1 to be low. The final verdict will The switch indicated in the diagram

or noise signal to the synthesis filters at then be: unvoiced. actually consists of two VCAs, A16 and

the right moment. If, on the other hand, the low pass filter A1 7. These ensure that in the end either

The 'voiced or unvoiced?' decision produces a signal that is greater than the carrier or the noise signal is fed to

mentioned with regard to figure 2 is that from the high pass, N1 will no the synthesis filters.

taken by comparators A10 . . . A12. longer be low and the outputs of All

Supposing an unvoiced signal arrives at and A12 will both be high. The detector

the input, the output of A10 will be- then decides: voiced. Further particulars

come high and that of A1 1 will be low. The other tri-state gates (N10 . . . N13) Preset pots PI and P2 preset the switch

In other words, the output of gate N1 in figure 3b serve to switch off the to voiced or unvoiced, as required. This

will be low, that of N4 will be high and detector if in the future it is to be con- can be done by alternately uttering 'A'

that of Nil will be low as well. In the trolled by means of a computer or and 'S' sounds in the microphone.

case, where the signal is unvoiced, the microprocessor. The two LED indicators Depending on the results, the sensitivity

output from the low pass filter will D15 (unvoiced) and D17 (voiced) dis- can be readjusted if necessary. P3 and

either be zero or at least smaller than play the state of the detector. Naturally, P4 preset the trigger point of the com-

1981

2-22 - elektor (ebruary

parators A10 and A12. This must be
done simultaneously with PI and P2.
Switch S1 a b acts as a select switch for
the voiced state. It has been added to
enable musical instruments to be used
as modulators as well. Whenever music
is entered at the speech input, closing
SI will prevent a sudden noise from
being fed to the filters at every high
tone. Whatever the signal, the detector
will always decide it is voiced.

The inhibit input (Z) may be used to
'block' all the detector's decisions. Then
of course the control inputs (V, X) must
be provided with information. Again,
this will come into effect once the unit
can be controlled by a (micro)computer.
OTAs A16 and A17 in the carrier/noise
circuit need to be very carefully cali-
brated with the aid of P7 and P8. This
must be achieved by a rectified signal at
the control input (R66, R77). This
method is spelled out in last year's
March issue, vocoder constructors will
no doubt remember the details. If the

unit is not properly calibrated irritating
click sounds will be produced when the
detector is switched, which happens
regularly in speech and singing.

Figures 4 and 5 represent the track lay-
out and component overlays of the
voiced/unvoiced detector printed circuit
boards. The detection circuit in figure 3a
is incorporated on the board shown in
figure 4, the remainder (figure 3b) being
installed on the board in figure 5.

The noise generator

Figures 6 and 7 show the circuit dia-
gram and the printed circuit board
respectively of the noise generator. The
noise generator is not only suitable for
the vocoder but also for various other
audio and acoustic measurements that
demand a quality noise signal. The out-
put can be switched from pink to white
noise and vice versa. The unit consists
of 7 commonly used ICs and a few
passive components.

the voiced/unvoiced detector

There is no need to describe its oper-
ation in full detail here, as various noise
generators have been published in
Elektor recently. All of them have their
pros and cons and this particular design
may be considered a combination of
them with the addition of a zero inhibit.
This concerns pseudo random noise
which is generated with the aid of a
31 bit shift register (IC3 . . . IC6). How
this works was described in the January
1981 'Swinging Poster' article, where
incidentally the same ICs were used.

N1 and N2 together form a clock gener-
ator at a frequency of about 500 Hz.
About 70 minutes are needed to run
through a 31 shift register in all its
states at this clock frequency. This will
make the noise sufficiently 'random'.
Diodes D1 . . . D31 combined with N3
provide the zero inhibit. As soon as the
'000 ... 0' state occurs, a '1' is entered
in the shift register by way of N5. Gate
N6 makes sure outputs 28 and 31 of the
shift register are EXOR back coupled.

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Buffer N4 is followed by a filter which
can be switched to pink or white noise,
whichever is required. The white noise
filter is a low pass filter at 23 kHz with
an edge of 6 dB per octave. IC7 acts to
amplifier the signal. The pink noise has
to be slightly more amplified than the
white, because its high frequencies have
already been filtered out and so cannot
contribute any further to it. PI is used

to equalise the output voltages for pink
and white noise.

The value indicated for the supply is
based on that of the vocoder (± 15V).
However, the noise generator will work
equally well at ± 12 V.

The connections

We are left with three new boards that
have to be connected to the existing

vocoder. From the block diagram in
figure 1 it can be seen what the pro-
cedure basically involves. There are two
possibilities:

1. Take an additional 'half bus board'
(EPS 80068-2). The three new boards
are exactly the same size as the other
vocoder boards and can all be provided
with a similar connector. If a connector
is mounted on all three, they can be

inserted into the bus board straight
away and this will then take care of the
individual connections. That's all there
is to it. The supply voltage(s) and points
1 i, j and g obviously have to be derived
from the vocoder bus board. How this is
done is shown in figure 8. At the same
time the additional half bus board pro-
vides a simple connection for the
existing supply board belonging to the
vocoder. This is an advantage, as there
was no room for this on the original bus
board. Now the supply board may be
inserted into the additional half bus
board and the connections remain as
indicated in figure 8.

Two more remarks: As illustrated in
figure 1, the existing connection be-
tween points i and j in the vocoder will
have to be interrupted when the voiced/
unvoiced detector is connected. The
i-j connections will therefore have to be
broken both on the 'old' and on the
'new' bus board.

Finally, to avoid any misunderstanding:
for the drawing of the connections in
figure 8 the circuit board drawings of
the old bus board were used. Be careful
not to mount any components on the
new half bus board, in spite of the indi-
cations in figure 8.

2. Don't use an additional half bus
board — make the connections your-
self. This will be necessary if the case is
not wide enough for another three con-
necting boards so that the expansion
boards will have to be mounted else-
where in the case. The wiring required is
shown in the diagram in figure 9. Again,
of course, the i-j connection on the bus
board will have to be broken.

Final notes

The 'computer' connections indicated in
the diagram as: unvoiced in (V), un-
voiced out (W), voiced in (X), voiced
out (Y) and inhibit (Z) are all situated
on the front of the 'switch boards' given
in figure 5. If required, these points can
be led out quite simply with a connec-
tor. This will enable experimenters to
control the unit by means of a com-
puter without having to cope with
complicated wiring problems.

As the connection diagrams of figures 8
and 9 show, both the voiced/unvoiced
detector and the noise generator can
derive their supply voltage from the
existing vocoder power supply. The
current consumption of the three

expansion boards adds up to about
100mA for the +15V voltage and to
about 50 mA for the —15 V. Since the
vocoder was issued with a 400 mA
transformer, the extra consumption will
by no means overload the circuit.

People have told us that the —15 V
section of the original vocoder supply
may encounter stability difficulties.
This can be remedied by substituting
C83 for a 2ji2/25 V tantalum electro-
lytic capacitor and C85 for a 1 /t/25 V
type. H

2-26 - elektor february 1981

emergency

i for the voltage supply

coming

soon.Tr

Those readers who suffer from a critical
ear for sound quality may cast a jaun-
diced eye at an article devoted to noise
reduction systems such as that pre-
sented on page 2,04 of this issue. They
may be forgiven for considering the
subject to be rather academic since it is
usually the case that a great deal of
information and data are discussed at
length but to no avail. The same prob-
lem always occurs, the special ICs are
only available to licenced manufac-
turers. And there it ends!

But not this time because Elektor are
going to publish a constructional article
on a very high quality noise reduction
system. This will not be the usual run
of the mill opamp circuit but an Elektor
designed and tested system featuring
the well known High-Corn 1C from
Telefunken. What is more, we will also
supply the 1C together with the printed
circuit boards, to provide the sort of
high quality project that readers have
come to expect from Elektor. To top it
all, the whole system has been tested
and approved by Telefunken.

. . . and it will not cost the earth to
build either.

emer^ney brake
for the Wta^e supply

This very simple circuit consisting
of a fuse and a zener diode
provides a useful method to avoid
damage to components sensitive
to excess voltages, such as
MOSFET IC's.

Overvoltage protection can only be
effective when the supply voltage is
carefully limited, in other words, by
including a voltage control system.
Even so, this still does not prevent the
output voltage from being able to rise
above the preset value. Even electronic
voltage controllers, whether they be
discrete or integrated, can fail. Although
the power supply usually suffers very
little damage, the circuit itself is often
considerably damaged. Brief voltage
peaks in the supply as well as complete
power failure can also write off ex-
pensive IC's. Prevention being better
than cure, or in this case, repair it is
worthwhile to include a fuse and a
zener diode at the output of the power
supply, as shown in figure 1.

1

This 'emergency brake' works simply
yet effectively. The zener voltage of
the diode is chosen at a level 2 V
higher than the output voltage of the
power supply, although this must of
course be below the upper threshold
of the supply voltage (absolute limit)
which the components in the circuit
can endure. If, for example, a CMOS

circuit is supplied with +15V, the
absolute limit for the IC's will be 18 V.
Thus, a 16 V zener diode will be in-
cluded with a break down voltage in
the 1 5.3 ... 1 7.1 V range.

Normally the zener diode will not
conduct, as nothing is wrong. As soon as
the output power supply voltage rises
excessively, however, the zener diode
will start to conduct and will prevent
the voltage from rising any further. A
high current will then flow through the
zener diode, causing the fuse to blow
after a very short period of time. Thus,
the minimum operating current rating
of the fuse must be above the normal
current consumption of the circuit.
The zener diode must be able to resist
this short burst of high current. The
zener diode should be cooled for
improved thermic resistance. At the
same time, the fuse will also protect
the circuit from 'shorts'. The zener
diode will then add to the protection
by limiting the voltage to about 0.7 V
when the supply is incorrectly polarized.

2

If the fuse operates at a fairly high
minimum current rating, a correspond-
ing power zener diode will cost quite
a bit of money. A more economical
solution is to control the fuse via a
thyristor, as shown in figure 2. As soon
as the supply voltage reaches the critical
level, the zener diode in the gate of the
thyristor will conduct, the thyristor
will switch on and blow the fuse.
Resistor R at the thyristor gate limits
the gate current and also the zener
current through the diode. H

DC converter

1981

elektor february

Specification

maximum output power 150W

output voltage range 30 ... 60 V
efficiency approximately 60%

quiescent current 800 mA maximum

KOW DC to DC
converter
for die car

Run-of-the-mill car radios and cassette
players rarely boast an audio power
output of more than 3 ... 6 watts. For
this reason many people look for im-
provement, such as combining two 4W
output stages to form a 1 5 W bridge am-
plifier. Unfortunately, that is just about
the limit, unless you resort to using an
output transformer, due to the low
(12 V) supply voltage available. An
output transformer reduces the load
impedance requirements for the loud-
speaker, thereby allowing a higher AF
output than could be expected from
the power supply alone. The main pro-
blem is, of course, that transformers
cause distortion and reduce the fre-
quency response. Apart from that,
suitable transformers are not really
readily available for the amateur con-
structor.

What is the alternative? Low cost ICE
(In Car Entertainment) amplifiers or
boosters seldom exceed the 1 5 W
threshold, and they can hardly be clas-
sified as Hi-fi equipment. Obviously,
quality high power amplifiers do exist,
but the price is something else! This
article describes a unit which will
increase the available voltage from 12 V
to the level required by powerful Hi-fi
output stages.

Converter

Basically, there are three types of con-

1. Conversion by 'chopping' the DC
input voltage and then multiplying

the AC waveform produced via a net-
work of capacitors and diodes (see fig-
ure 1). Technically, this is a very elegant
solution and one which requires very
little space. However, the principle,
especially where high power is involved,
requires the use of high quality (expens-
ive) components.

2. The second type of converter also
uses the chopping method, but this

time the AC waveform is fed to a trans-
former before being rectified and
smoothed (see figure 2). By using a
transformer the voltage can be
stepped up (or down) to vir-
I tually any desired level, how-
ever, the higher the voltage, the
less current available.

3. The third method consti-
tutes a compromise between
the two with respect to the
value of inductance required.
Here, the DC input voltage is
again chopped before being
rectified and smoothed, but
this time the output level is
monitored and compared with
a reference level. In turn, the
itor controls the switching speed
of the chopper so that a steady output
voltage is maintained. The principle of
the above is shown in figure 3. By using
this method the circuit requires a rela-
tively small coil with few turns thereby
making the device more compact than
if a full sized transformer were used.

Readers who consider that the power output and performance of their
car stereo equipment is rather on the low side, in spite of the 15 W that
the normal booster will provide, are now able to "drive" a high power
Hi-fi output stage by incorporating the DC to DC converter described
here.

high power Hi-fi

2-28 - olektor february 1981

150 W DC to DC

for the

Circuit

The complete diagram of the DC to DC 1
converter is given in figure 4. The con-
trol circuit consists of a reference
voltage source (R4, D1 and C3),a com-
parator (opamp IC1) and a 555 timer
(IC2) connected as an astable multi-
vibrator (or, more accurately in this
instance, as a pulse generator). Transis-
tors T3 . . . T5 (connected in parallel)
form the electronic switch shown in
the block diagram (figure 3).

The output signal from the pulse gene-
rator is (current) amplified by transistor
T2 to drive the three switching tran-
sistors. Inductor LI acts as a voltage
conversion coil. Components D3,

C8 . . . C11 and L3 have been included
to rectify and smooth the output
voltage.

The comparator (IC1) compares the
output voltage of the converter via PI ,

R2 and R3, to that of the reference
voltage across D1. If the voltage at the
inverting input of the comparator is
lower than at the non-inverting input,
the comparator output voltage will
be high. This then enables the pulse
generator which produces a series of
negative-going pulses until the output
voltage of the converter corresponds to
the value preset with potentiometer PI.

As soon as this is the case, the output
of the comparator will go low and the
pulse generator will be inhibited. The
output level of IC2 will then remain
high, so that transistors T2 . . . T5
will no longer conduct. This ensures
that the conversion efficiency and
quiescent current figures take on accep-
table values.

Transistor T2 amplifies the output
pulse current to about 1.5 A. This
means that as transistors T3 . . . T5 are
overdriven by this base drive current,
superfluous charge carriers arise in
their bases which would prevent the
transistors from turning off fast enough.
Normally, when inductive loads are
switched at high frequencies a large
amount of energy would be dissipated.

For these reasons, components R12,

L2 and D2 are included in this section
of the circuit to speed up the transfer
of charge carriers. The voltage rise at
the collectors of the switching transis-
tors is slowed down to a certain degree
by D4 and C7. During the final phase,
capacitor C7 discharges through R22.

When the three transistors are turned
on, the current flowing through the coil
(LI) produces a magnetic field. When
the transistors are turned off, this mag-
netic energy is re-converted to electrical
energy by means of the back emf gene-
rated in the coil. The voltage at the
anode of D3 will therefore rise rapidly
until it become slightly greater than that
of the reservoir capacitors C8, C9 and
CIO. The D3 will conduct and the capa-
citors will be "topped up". Three reser-
voir capacitors are used, instead of the
usual one, so that a large capacitance
can be obtained from relatively small
(dimension-wise) capacitors. This ena-

bles the reservoir capacitors to be
mounted on the printed circuit board.
The low pass filter L3/C1 1 ensures that
any high frequency noise is eliminated
from the DC output voltage. Which is
just as well, as this could influence the
AF amplifier connected to the
converter.

Selecting the components

Since the DC to DC converter needs

to be capable of delivering up to 1 50 W,
either large inductors or high switching
frequencies will have to be employed.
As the output voltage is to be greater
than the input voltage, very short
switching times will be required. In this
particular instance, a compromise be-
tween the size of the inductor and the
availability of HF components was
sought.

The switching frequency is no higher

150 W DC to DC converter for the car

than 40 kHz and the duration of each
pulse is approximately lOps. These two
parameters are determined by the values
of R5, R6 and C4. As the pulse length
is relatively short and the repetition
rate relatively fast, the switching tran-
sistors will have to be quality devices -
the ubiquitous and universal 2N3055
just doesn't come up to spec! slightly
more expensive, but not too difficult
to obtain, are the BD240B and BD245B
types. Other transistors with a similar
specification to that shown below can,
of course, be used instead:

UCEO = 80 V, fj = 3 MHz,
toff = 300 ns.

Three parallel connected output transis-
tors are incorporated to cope with the
high peak current produced. Diode D3
must be a fast recovery type such as
the BYX371/600 (or equivalent).

Inductors

Coil LI is made up from a ferrite pot
core with the following dimensions:
diameter = 50 ... 70 mm;
height = 30 ... 40 mm.

The core must be capable of operating
to frequencies of 100 kHz. The number
of turns around the core depends on
the value of inductance required and the
so-called A[_ value of the core (expres-
se d in nano-henries). The required
inductance for LI was calculated to
be 144 pH. Therefore the number of
turns required will be

The cores indicated in the parts list
have A|_ values of either 1000 or

630 nH. In other words, the number of
turns required will be

V I 44 x 1 0' 6

1 000 x 1 0~ 9 ” ’ 2 "

V 144 x 10' 5
6307W* =15

Since high currents are involved, and to
ensure that the circuit operates effi-
ciency, the diameter of the wire used
needs to be fairly large. However, when
high frequencies are being employed,
the 'skin effect' becomes most distur-
bing, therefore, it is far better to wind
several thinner wires around the core.
Thus:

— core with A|_ of 1000 nH:

12 turns of five 1 mm diameter
enamelled copper wires in parallel
- core with Al of 630 nH :

15 turns of five 1 mm diameter
enamelled copper wires in parallel
If wires of different diameters are used,
the number of turns will have to be
calculated accordingly.

Pot cores without an air gap have much
greater A|_ values, therefore the results
from the above calculations will no
longer be applicable. However, an air
gap can be introduced by inserting a
cardboard disc between the two halves
of the pot core. Unfortunately, this
does mean that the pot core can not be
mounted on the printed circuit board
with a single (central) screw. The size
of the air gap determines the efficiency
of the circuit. To obtain the optimum
efficiency level, various thicknesses of
cardboard should be tried until the
quiescent current is down to a minimum

elektor february 1981 - 2-29

(no load).

Inductors L2 and L3 are ring core
interference suppression inductors as
used in thyristor and triac control
circuits. The former is a 2 A type and
the latter a 6 A type.

Construction

The entire circuit can be mounted on
the printed circuit board shown in
figure 5. The two halves of the pot core
should be glued together to prevent the
coil from 'whistling'. The output tran-
sistors, T3 . . . T5, and the fast recovery
diode, D3, can all be mounted on the
same heatsink (2° CAW or greater). Mica
washers and a layer of heat conducting
paste should be used when installing
the power transistors and the diode. In
the prototype unit the heatsink was
mounted vertically on the board
(200 mm wide by 75 mm high) and
therefore doubled as the rear panel of
the case. A metal case is an absolute
must in this instance as without one the
converter would radiate interference
signals throughout the entire long and
medium wavebands. Not only will the
Post Office object to this, but also it
will be virtually impossible to receive
any broadcasts on a radio powered by
the converter. Transistor T2 also requires
a small heatsink.

Heavy gauge wire (at least 2.5 mm 2
diameter) must be used to connect the
DC to DC converter to the car battery
and to the load. The leads can be solde-
red directly to the printed circuit board,
but some provision for retention must
be made to avoid damage to the actual

Figure 4. The complete circuit of the DC to DC converter. Inductor LI is made from a pot core which has an air gap (see text for winding details).

Inductors L2 and L3 are 2 A and 6 A interference suppressors as used in thyristor and triac control circuits.

board. This can be achieved by utilising
'P' clips or rubber grommets (or both).
If desired, standard %" 'blade' connec-
tors can be used to facilitate construc-
tion and installation. In addition, the
unit requires an on/off switch and a
30 A fuse must be included in the 'live'
battery lead.

Operation

Once construction has been completed,
the circuit can be connected to a 12 V
DC power supply having an output
current capacity of around 2 ... 3 A.
A multimeter switched to the 10 A
range is connected in series with the
positive supply lead. With no load con-
nected, a large amount of current will

flow initially until the capacitors in
the unit become fully charged. After a
short period of time, the quiescent
current should drop to about 50 . .
. . 800 mA, depending on the preset
output voltage. The required output
voltage can then be selected, between
30 ... 60 V, by adjusting preset poten-
tiometer PI . When a load is connected
to the converter it should be possible to
hear the change in pulse repetition fre-
quency from one (or more) of the
coils.

Uses

This multi-purpose circuit is ideal for
powering two 40 W Edwin amplifiers

driven from the car stereo system. It
is also possible to install a low voltage
temperature controlled soldering iron
into the car (provided, of course, that
it is a DC model). Linear amplifiers and
other ham radio equipment (and shortly
CB?) requiring more than 12 V can also
be connected to the converter. M

Note: The output voltage can be reduced
by decreasing the value of R2. If the
converter only needs to supply 50 W,
the following components can be
omitted: R11, R16...R21, CIO. T4
and T5; the value of R7 must then be
increased to Ik 5.

knv noise
2 metre
preamplifier

This preamplifier is intended for use in receivers of the 2 metre amateur
band wave (144 MHz). By changing a single resistor, it can be made for
either very low noise or a low intermodulation distortion.

Internal noise

This low noise VHF preamplifier
operates with a particular type of
extremely low noise, high frequency
transistor, the BFT 66. This transistor
ensures that the noise contribution of
the amplifier stage thus obtained is
small, for usually the lion share of
noise produced is caused by the transis-

The noise contribution of an amplifier
is rather an abstract concept and it
is not the purpose of this article to
define it. However, it will be clear that
even this can be expressed as a factor:
the noise factor. Basically, this in-
dicates the relationship between the
quantity of noise present in the output
signal of an amplifier and the quantity
of noise which it would contain if the
amplifier would merely amplify without
adding to the noise itself. Usually this
ratio is expressed in dB.

An amplifier which produces no internal
noise at all has a noise factor of 0 dB.
The output signal will then contain
exactly (relatively speaking) the same
amount of noise as the input signal.
Such amplifiers unfortunately do not
exist, although there are a few which
come near to meeting this figure.

The amplifier described here has a
noise factor of less than 1 dB, which
means the signal to noise ratio only
deteriorates by 1 dB. For a VHF
preamplifier this is an excellent per-
formance.

The circuit diagram

The circuit diagram of the 2 metre
preamplifier is not nearly as com-
plicated as most circuits of this nature.
It is possible to connect a normal 50
ohm aerial to the input. However, since
the impedance of the aerial often
deviates from that required for an
optimum noise factor at the base of the
transistor, the aerial cannot be directly
connected to the base. For this reason
a pi network is placed between the base
of T1 and the aerial input. This consists
of trimmers Cl and C2 and the coil
LI. The pi network literally matches
the impedances.

In the collector lead of T1 there is a
resonance network consisting of L and

C4. The ferrite band FB is included
to prevent oscillation. In many cases
it may not be necessary. An alternative
is to replace it with a 1 5 S2 resistor.

The collector current of the transistor
will be the main determining factor in
the noise contribution of the amplifier.
The preset PI is used to adjust this.
With the component values given in
the figure, the collector current may
be preset to 3 mA since at this figure
the BFT 66 gives its best noise perform-
ance. The collector current can very
easily be determined by measuring the
total current consumed by the circuit,
or the voltage across R3.

With a collector current of 3 mA,
the internal noise contribution of the
amplifier will be less than 1 dB. To
give some idea this means that in a
receiver bandwidth of 3 kHz, an input
signal of only 25.6 nV (0.025 /iV)
will already produce an output signal
that can be detected. The 3 dB
bandwidth is 5 MHz.

Intermodulation distortion

It is clear that the collector current has
a great influence on internal noise.
However, something else is also highly
dependent on the collector current,
namely, intermodulation distortion.

low noise 2 metre preamplifier

This is the creation of all kinds of
by-products in the output signal which
were not in the input signal. The reason
for this is that the transistor is not
linear.

The intermodulation distortion can also
be expressed in dB, as the ratio between
the desired signal and the intermodu-
lation products. For obvious reasons,
this ratio should be as high as possible.
In other words, an ideal amplifier would
have virtually no intermodulation dis-
tortion coupled with an extremely low
noise factor. It would be ideal if the
collector current producing the lowest
possible noise could at the same time
ensure the lowest possible intermodu-
lation distortion. This however is
wishful thinking.

From the noise point of view the ideal
collector current would be 3 mA. How-
ever, this would produce intermodu-
lation products of only 10 dB (for
800 MHz admittedly). By increasing
the collector current to 10 mA, the
intermodulation can be reduced to
—60 dB - a considerable improvement.
The price that has to be paid for this
is an increase in the noise factor by
approximately 0.5 dB.

Depending on what you want, the
amplifier can be tailored quite simply.
A collector current of 3 mA will give
a low noise amplifier. Taking the
current up to about 10 mA and chang-
ing the value of R3 to 330 fi will
produce an amplifier with very little
intermodulation distortion.

Constructional details

It is advisable to use low noise metal
foil resistors for R1 and R2. Both the
coils LI and L2 are 'air cored', that is,
wound on an 8 mm diameter former
which is then removed. The 1 mm
copper wire used for the coils should be
silver plated. The winding details are
x =, for LI 6 turns, and for L2 4 turns
with taps at the first and second turns as
shown in the diagram. M

2'A digit DVM

>r febri

1981 - 2-33

Building a digital voltmeter is so simple
nowadays that there is really nothing
to it. All it takes are a couple of ICs
which incorporate the entire circuit:
A/D converter, counter and display
control, so that a display will usually
be enough to complete the job. The
advantages speak for themselves: the
circuit is easy to build, requires little
calibrating and is fairly accurate. Unfor-
tunately, however, all this is outweighed
by one major disadvantage — try obtain-
ing the prescribed ICs at your local
dealer's and you'll find he won't have
them in stock. In other words, the
hobbyist ends up not constructing a
voltmeter at all.

To make it more worthwhile, our
designers racked their brains to come up
with a solution using 'ordinary' parts.
The result? A 2V4 digit voltmeter with a
very reasonable accuracy of ± 0.5%. It
is sufficiently accurate for all the nor-
mal chores, especially considering that
the accuracy of a decent analogue multi-
meter amounts to several percents.

21 digit DVM

Three displays, six ICs and a handfull of components are the only
ingredients needed to cook up this digital voltmeter. No attempt has
been made this time to produce an exotic recipe using rare ICs and
extreme accuracy, but a plain, simple voltmeter including readily
available components.

An interesting circuit
Figure 1 shows the diagram of the DVM.
Most of the work is done by IC1, a
kind of jack-of-all-trades where
controlling displays is concerned. This
CMOS 1C contains a number of items,
a 4 digit counter, a latch, a seven seg-
ment display control and a multiplex
circuit. In this particular circuit only
three of the four displays that could be
connected to it are used. The multiplex
outputs A, B and C switch the display
common cathodes by way of transistors
T1, T2 and T3.

A falling edge at the latch input shifts
the contents of the counter to a slave
flip-flop. A logic one at the reset input
resets the counter. The contents of
the slave flip-flop can be seen on the
displays.

The latch and reset signals are provided
by IC5, N1 and N2 and their corre-
sponding components. IC5 is connected
as an astable multivibrator with a fairly
large pulse to interval ratio having a
frequency of about 2 Hz. With the aid
of C6, R17, C7 and R18 gates N1 and
N2 derive two pulses for the reset and
latch control from the multivibrator.
Since the reset pulse arrives a little

later than its latch counterpart, first the
contents of the counter are shifted to
the slave flip-flop and then the counter
is reset. The number of pulses entering
via the clock input of IC1 during the
interval between the reset and latch
signals is therefore shown on the dis-
play.

Use has been made of a voltage to cur-
rent converter to convert the voltage
measured into a frequency. At the same
time this determines the time constant
of the multivibrator and consists of a
voltage controlled current source.

The voltage to be measured is now con-
nected between the supply of the cur-
rent source (6.8 V) and the non-
inverting input of IC3. IC3 then regu-
lates its output voltage in such a way,
that T6 conducts until the voltage of
the inverting input is practically the
same as that at the non-inverting input.
This means the voltage across R12 and
P2 is equal to the voltage that is to be
measured. Thus, the current passing
through P2 and R12 is equal to the
test voltage. This current is derived
from the collector of T6. Its level
determines the charge time of capacitor
C9. The multivibrator constructed with
IC4 is arranged so that C9 will be dis-
charged whenever its voltage is equal to
half the supply voltage (in this case the
stabilised 5 V). In other words, when
the input voltage is high, capacitor C9
will charge and discharge very quickly,
as a result of which IC4 will generate
a high frequency to the clock input of
IC1. The final outcome is a large figure
on the display.

The charge current of the capacitor is
Uj n

equal to pg + Ri 2 ' S ° t * lat meter

can be calibrated with potentiometer
P2. PI takes care of the zero setting.
Diode D1 serves to protect the input
against voltages with the wrong polarity.
There is also a circuit that protects
against excess input voltages in the dia-
gram, even though this is difficult to
see. The DC at the cathode of D2 is
maintained at 3.9 V by R11 and D3.
The supply for the current source is
also at the ® input and is 6.8 V. If the
input voltage is higher than the differ-
ence between two zener voltages plus
the threshold voltage of diode D2
(6.8 -3.9 + 0.6 = 3.5 V) D2 will con-
duct and the remaining voltage will
be dropped across RIO. This helps pro-
tect the circuit against input voltages
of up to 100 V.

The supply of IC3 has been delibera-
tely chosen at a higher level than that
of the current source, because when
the input voltage is 0 V the output vol-
tage of theopamp should be 6.8 - UbE-
This would not be possible if the supply
of the opamp were also 6.8 V.

In addition, high voltages can be indi-
cated on the display. IC1 has a carry
output which generates a pulse when-
ever the maximum level of the counter
is exceeded (read-out 199). By way of

2-34 - 1

1981

; digit DVM

Figure 1 . The DVM circuit diagram. The dotted area is incorporated on the universal display board and the rest on the DVM board.

a peak detector (R19, D6, Cl 1 and R20)
this pulse is detected causing N3 to flash
the point of Dpi via T5 at the fre-
quency of IC5.

Finally, it should be noted that the Dpi
readout is suppressed whenever the
input voltage is 0.99 V or smaller.

The supply for the circuit (apart from
the current source) is provided by an
integrated voltage regulator 7805.

Construcion

All the parts involved in building the
DVM are mounted on the two printed
circuit boards shown in figure 2. The
dotted area in the diagram given in
figure 1 is mounted on the display
board. This section happens to be
universal and therefore suitable for
various circuits. Although four displays
and four control transistors are shown,
the DVM only requires the first three
displays and transistors. The decimal
points all have external connections so
that they can be converted if the meter
is to be used in various measurement
ranges. For the standard range
(10 mV... 2 V) the point of Dpi is
connected to resistor R8.

2

Figure 2. This resistor divider enables the
number of measurement ranges to be
extended. In this case, resistor R9 may be

The input circuit and the oscillators
are incorporated on the second board.
The unit allows the two boards to be
placed one behind the other after which
the corresponding points are connected
by wire links. The only component left
to be added is the transformer as this
is not included on the board.

Using the resistor divider illustrated in

figure 2 the meter can be provided with
several ranges. R9 can can then be omit-
ted. Make sure neither input is connec-
ted to the ground of the supply.

Calibration

As mentioned earlier, the DVM ranges
from 10 mV to 2 V. Its accuracy will
them be at ± 0.5%. To start with, the
input is 'shorted'. Then the wiper of PI
is turned towards pin 5 of IC3 (anti-
clockwise) slowly until .00 appears on
the display.

Now the input may be re-connected
and the meter can be calibrated. A refe-
rence voltage is fed to the input and the
meter is calibrated with P2. Usually,
however, there will be no accurate,
known voltage available. The simplest
thing to do is to compare the result
with that of another meter, an accurate
one, at an input voltage of about 1 V.
Whether or not the meter has been
correctly calibrated will then depend
on the quality of the other one used.

If the meter has the resistor divider
shown in figure 3 added to it, the accu-
racy in the other ranges will of course
depend on the resistors used.

2-36 - etoktor february 1981

Dr. Wagner

The first thing that catches the eye is of
course the instrument's shape. Hardly
that of a piano or organ! The version
shown in the photograph certainly
looks sophisticated but the electronic
contents are not nearly as elaborate as
the appearance of the prototype might
suggest. Nevertheless, no short cuts
were taken in the lab and all the vital
components are there, alive and singing,
including a build-in AF amplifier, a
vibrato circuit, a spring line reverb unit
and even a connection for a microphone,
turning the player into a regular one-
man band.

Inquisitive readers will have sneaked a
look at figure 5 by now, but let us
return to the circuit diagram presently
and now consider the musical aspect.
For one thing, the 'octave shift mechan-

wnjgnephoiie

A mini keyboard with an 'octave shift mechanism'.

This ingenious little instrument will save worn-out fingers and relegate
years of tedious practice to the dim past. As can be seen from the
photo, it includes a microphone for vocal accompaniment and a unique
'octave shift mechanism' which moves the notes up or down an octave
without stretching the player.

ism' alone is worth a closer inspection,
as this is what makes the 'Wagnephone'
so much easier to play than an ordinary
keyboard instrument. In fact, it cuts
the average learning time required by
more than half.

Scales that tip the scales
There's no doubt about it, the world
would be a boring place to live in
without any music to liven it up.
Thousands would love to be able to
play the piano or the organ, but are
deterred by the great difficulties in-
volved in learning how. If only it were

Looking at a 'normal' keyboard, any
short cuts around the problem seem to
be out of the question. To start with,
so much fingering is entailed! The
fingers are worked to the bone, as they
not only have to strike the keys in a
vertical movement of the hands, but
they also have to move horizontally,
up and down the keyboard, in order
to find the right notes . . . and often at
a cracking pace!

People who have tried sitting in front
of a keyboard and striking a note while
keeping one eye on the manuscript
know that this takes years of practice.
After all, what it scales down to, is
teaching the fingers to build up a com-
plicated system of reflexes, so that.

vagnephone

elektor february 1981 - 2-37

eventually, they can pick out a tiny
white or black target from the total
length of the piano and 'hit' it without
exceeding the 1% tolerance limit. Every
wrong note jars the ear instantly,
so there are no second chances here.
Playing slowly requires a certain amount
of effort as it is, playing at high speed
with the feet is a feat indeed . . Jokes
aside, the technique and patience in-
volved in playing a simple piece are
such, that it's hardly surprising most
people will think twice before even
trying.

Fingering on a small scale
Now that we've scared the daylights
out of any would-be pianist — yes, there
is an easier way to learn thanks to the
Wagnephone. The instrument in the
photograph closely resembles a recorder,
even though it has a keyboard. Why?
Because a recorder happens to be rela-
tively easy to play. The time to master
the art takes somewhere between Vs
and Vio of that on the piano or organ.
The big difference between the two
types of instruments is that the recorder
involves fewer movements (of the
hands, not the music!). The fingers
cover a row of holes and remain fairly
stationary, except that they have to
be lifted every now and then. To be
fair, the breathing technique is a differ-
ent story, especially when an octave
has to be jumped. However, this is
irrelevant here, as in spite of its shape
the Wagnephone is not a wind instru-

Avoiding the key-search problem
altogether, the Wagnephone manages to
jump an octave higher or lower without
the player moving an inch. All he has to
do is lift one finger. Again, before dis-
covering the magic principle, there are a
few interesting facts to know about
music in general.

Western music is based on major and
minor keys which in turn are divided
into tetrachords. C major, for instance,
consists of the two tetrachords c-f and
g-c. These have nothing to do with
chords and are sequences of four notes.
The first three notes are always separ-
ated by a whole tone, whereas the last
two are a semitone away from each
other. Thus, in the C major example, e
and f on the one hand and b and c on
the other are semitones. Together the
two tetrachords constitute an octave.
Playing the C major scale up and down
the piano requires a certain amount of
practice until the right fingers instinctly
hit the right note. Since a hand only
has five fingers and a piano has many
more keys, it is obvious the pianist soon
runs out, which means tucking the
thumb under in certain places (for
which there are special rules): The in-
ventor of the Wagnephon wished to
reduce the number of physical move-
ments involved in playing on an ordi-
nary keyboard. As can be seen from the

Figure 2. This is the seated position for
playing the Wagnephone.

photograph, the Wagnephone keyboard
is designed so that the four fingers (ex-
cluding the thumb) of the left hand
cover the first tetrachord (c . . . f) and
those of the right hand cover the
second (g . . . c). Each finger thus has
its own note and each hand its own tetra-
chord, making it so much easier to
learn that within no time the player
will be a musician 'to his/her finger-

The Wagnephone and octave shift

Now that the principle behind the
instrument is clear, what about the
instrument itself? Does it accomplish
what it sets out to do? The 'octave
shift mechanism' that it incorporates
is a very common organ stop and is
simply a switch enabling the music to
go up or down an octave without having
to extend the keyboard range. The

3 *

Figure 3. If you wish to sing along, you can
also stand like this.

Wagnephone's one and only octave
is divided into two tetrachords. When
these are convered by the fingers of
both hands, the thumbs are free to
depress the two pushbuttons con-
veniently positioned between the two
rows of keys shown on the right-hand
side of the photo in figure 2. In the
photo all the fingers (except one) are
held a little above the notes for clarity's
sake. Usually, however, they will rest
gently on the keys. As always, the black
notes provide the semitones in keys
other than C major.

Whenever the melody exceeds the
single octave range, one of the thumbs
will have to be depressed. If the music
is to go up an octave the right-hand
thumb will depress the pushbutton, if,
on the other hand, the melody is to
go down an octave, the left-hand
thumb will depress the other switch.
As soon as either switch is released,
the original octave will return.

It will be clear from the above how
little learning effort is entailed before
the fingers race along the keys smoothly.
Soon, for instance, the little finger on
the right hand will become automati-
cally associated with the upper C . . .
and all without looking. You can't
miss!

The person in figure 2 is showed to be
seated as she would be to play the piano.
For beginners this has the disadvantage
that they will have to look up from the
keys to read the notes. Once they have
mastered this, they can easily change
over to the piano or organ.

Reading music has always been con-
sidered a problem and there are plenty
of skilled amateurs who manage
without. Sometimes it is remedied by
drawing the bars vertically rather than
horizontally. In the case of the Wagne-
phone the difficulty just does notarise,
as there is no need to keep an eye on
the fingers (see figure 4).

The circuit

At the heart of the circuit diagram in
figure 5 there are two function gener-
ators of the 8038 type. This 1C was
chosen for two reasons. First, it features
excellent stability and is linear within
a wide range. Second, it requires only
very few external components, which
means the circuit can be highly com-
pact, easy to control, simple to build
and requires very little calibration.

To start with, let's take a look at IC3.
As usual the frequency of the 8038
is determined by an external RC net-
work. In this case by C11 and the
resistor network R c . . . Rh (either 1%
or 5% types). According to which of
the keys is depressed (C...C1) the
resistor chain is connected to +Ub at
a certain point. The function generator
then provides the corresponding fre-
quency. The keyboard can be tuned
with preset pot P\/- The LED, D1,
shown in the diagram has a purely

the wagr

bookshop

1981 - 2-39

decorative purpose, lighting as soon as
the supply voltage is switched on.

The frequency can be obtained from the
output of the 8038 (pin 9) in the form
of a square wave voltage. At outputs 2
and 3 additional sine wave and triangu-
lar signals are available at the same
frequency for those who enjoy
experimenting.

Using the information given so far, all
the notes of an octave can be played
(including all the semitones, bringing
the total up to 12). With the aid of
input 8, which in principle is meant for
frequency modulation purposes, the
range can be moved either up or down
an octave. Thus, the octave shift
mechanism is very simple. If the voltage
at pin 8 is decreased by way of the
octave switch S3, the frequency will
move up an octave; if the voltage is
raised via S4, the frequency will drop
an octave. The octave shift is tuned by
means of P4, P5 and P6.

The filter which is connected to Pin 9
acts to 'shape' the sound. It mellows it
somewhat — this effect can be partly
counteracted, if necessary, with S7.

The second 8083 (IC5) is connected in
parallel to IC3. This generator, however,
is preset one octave higher and is shifted
slightly in frequency (a few Hertz) by
presetting P'v. This greatly improves
the end product, for when IC5 is in-
cluded, with the aid of S8 and S9
('soft' or 'powerful') respectively, — just
like S6 and S7), a very slight phasing
effect is obtained and the tones sound
fuller. The frequency determining ca-
pacitor Cl 5 which is connected to
pin 10 of IC5 has half the value of
Cl 1.

The wagnephone's is also largely depen-
dent on the vibrato generated by IC4.
An ordinary 555 has been used for this.
Cl 4, R20 and P3 determine the fre-
quency, being several Hz. The output
of the vibrato circuit is fed to the modu-
lation input (pin 8) of IC3 and IC5.
S5 enables the effect to be switched on
and off.

In order to amplify the signal suf-
ficiently to drive a loudspeaker an
integrated power amplifier (IC1 =
TDA 1 905, SGS-ATES) has been in-
cluded. This provides about 5 watts
maximum output power. By inserting
a springline reverb in the feedback
network of the output amplifier the
sound is further improved in quality.
This effect can be adjusted in volume
with S1 1 . If required, a microphone
can be connected to the input of IC1.
If this is done in the manner suggested
in figure 5, P2 can be used to mix the
mike signal to the music. The type
used in the prototype was an electret
which included a FET preamplifier,
not expensive nowadays and fairly
easy to obtain.

Since IC1 fortunately is not sensitive
to variations in supply voltage, only the
voltage for the tone generators has to
be stabilised and a simple 1C stabiliser
(IC2) will take care of this. As the

Wagnephone has been designed to be
battery powered, current consumption
is, of necessity, as low as possible.
Quiescent current, or when headphones
are used, is about 50 mA. This will of
course after according to the volume
preset with PI. The supply voltage must
be between 12 and 18 V. The speaker
used in the prototype was a special
high frequency horn with a 15 watt
rating. Together with the rest of equip-
ment it constitutes a neat until.

Expansion possibilities
The high quality sound (surprising
considering its size) can be improved
by connecting a good power amplifier
and a couple of decent loudspeaker
units, to the circuit. The improvement
will be quite amazing, however, if a
graphic equalizer is added. The equalizer
input is then connected to the wiper
of PI and the signal processed by the
equalizer is fed back to the input of
IC1 or that of an external amplifier.
This enables the Wagnephone to render
very good imitations of instruments like
the saxophone, clarinet and oboe. In
fact the Wagnephone is almost a synthe-
sizer. When a vocoder is connected to
it, a whole range of special effects can
be explored. This can include robot or
Donald Duck type voices and serious
musical effects. What's more the singer
does not even have to be in tune, as
the sound of his/her voice will be
'reformed' by the build-in micro-
phone. M

bookshop

It is not often that a book is reviewed in
Elektor but when a book arrives that fulfils a
need by our readers we will comply in Book-

We receive repeated requests for a book that
is suitable for the electronics enthusiast or

The approach is one of simplicity and practi-

THEART OF ELECTRONICS,
Horowitz and Hill,

Cambridge University Press. £ 12.50,
ISBN O 521 29837 7

Sharp's special shapes

In addition to conventional tubular
lamps. SHARP CORPORATION otter
of special shapes. Designed for flush f
front panels, the shapes presently
include round 'point' indicators, eq
and isosceles triangles, square and flat

Telephone: Luton 105821 41 1085

HF digital frequency counter

High-quality 8-digit frequency counters for
line or square wave inputs from B. Davis
Electronics, start at only £ 1 16 for a 60 MHz
instrument with a resolution down to 10 Hz. Power FETs

Four devices presently available have thi
lowing key specifications:

V DSS (min) R(on) (
HPWR-6501 450 V .85

HPWR-6502 400 V .75

A new series of matched, low-cost ultrasonic
transducers from Impectron Limited are
small, light and highly sensitive. They offer
excellent performance in applications such as
industrial control and intruder detection
systems.

The EFR-OCB25K5 and EFR-RCB25K5 are
transmitter and receiver respectively, with
centre frequency of nominally 25 kHz.
Sensitivity is around -65 dB /V/pbar with
minimum bandwidth of 3 kHz. Overall dimen-
sions are 1 inch long (body length 0.37")
by 0.95 inch diameter for both receiver and

All lamps can be supplied with wire termin-
ations or popular base styles, including pro-
prietory all-glass integral pin versions. Conven-
tionally. their leads are tin plated to MIL
specifications, though versions with the leads

Voltage, current and light output specifi-
cations for the range cover an extremely
broad spectrum of application requirements,
and life expectancy extends beyond 100.000

burn-in period of 16 hours at the rated
voltage, and so offers a very high degree
of reliability.

Diamond H Controls Ltd.,

Vulcan Road North,

Norwich NR6 6 AH.

Telephone: Norwich 106031 45291/9

sockets which can accept pre-programmed
E-PROMS containing an assembler, debug
program or BASIC
Chip tech Limited,

Tewin Court,

Welwyn Garden City.

Hertfordshire AL7 1AU.

Telephone: (07073) 33260

ABS plastic keyboard enclosure

West Hyde Developments Limited have added
a purpose-designed keyboard enclosure to

This extremely lightweight yet robust
enclosure provides a cost-effective and simple
approach to mounting a full alpha-numeric
keyboard which will blend with any associ-
ated hardware such as VDU screens. The
unit is produced from textured ABS plastic
and is available in black, or in a wide variety

Portable E-PROM Programmer

100 CPS bidirectional printer

The MPI Model 88T bidirectional impact

labels, varying from 1" to 9.5" in width.
An easily inserted long-life ribbon cartridge
Bliminates messy ribbon changing.

Selectable character densities allow formatting
of output in either 80, 96, or 132 character
lines. Double width characters are software
selectable for any of the three character

Diamond H Controls Ltd.,

and single or multiple connector tabs secured Vulcan Road North,
by either a hollow or solid rivet. Norwich NR6 6AH.

Designed to provide ease of accessibility to Telephone: Norwich (06031 45291 /9
the connector tabs, the 250 D series is well ( y g

suited for high-speed or mass assembly work
in both domestic and commercial equipment.

Their connector terminals are of the standard
6.3 x 0.8 mm size, and offer the possibility
of up to six terminations per cavity. The
compact design of the terminal block, which
is only 13 x 13 mm in section, means that it

**•«•• dip ">«•»

characteristics of the series include a p - Caro and Associates Ltd have introd
of binary coded DIP

be either serial IRS232C

loop) or parallel (Centronics Keyboard switches
ASCII) with a standard two Two ranges of keyboard

Foundry Lane,
Horsham,

W. Sussex. RH13SPX.
Tel: 0403 501 1 1

Low-cost terminal blocks

from Diamond
The Series 400 and 475 keyboard switches,
with their low-profile equivalents, the 415

and feature contact ratings of 5 V d.c.
resistive, at 100 mA with a breakdown voltage
in excess of 500 V a.c.; Contact resistance is
less than 100 milliohms throughout the life of
the switch, and contact bounce is less than
3 milliseconds. Both normally-open and
normally-closed versions are available, and
each version has a total key travel of 0.1 25 in.
Both Series are fitted with plug-in printed
circuit terminals, and embody a snap-in

ct pins are compatible with conven-
DIP switches and the sealing features
i that they are gas and solvent proof. A
lerable amount of mechanical and

les which are designed to withstand

m

UiMJupdj:g

'Scope for servicemen, industry 240 v mi
and Hobbyists 150 mm i

The new Model SB 3 M oscilloscope from high. Pricr
Albol Electronic will serve most purposes Albol Elec
required bv industrial and service and hobby 3 Crown l
engineers, and yet manages to keep on the Crown St.
right side of the significant £100 price LondonS
barrier. With a bandwidth of 0 to 3 MHz at Telephone
—3 dB (extending to 6 MHz at -6 dB). the

in the range 10 Hz to 500 kHz. If internal
synchronisation is used the range extends
from 10 Hz to 3 MHz. The internal trigger
threshold is 5 mV. and the external 100 mV.
The SB 3 M takes about 20 watts from the
240 V mains, weighs 4.5 kg. and measures
150 mm wide by 340 mm deep by 280 mm
high. Price: £ 99.00 plus VAT.

Albol Electronic & Mechanical Products Ltd.,

■in

Digital heart beat monitor
(Summer Circuits '80)

In the circuit diagram pin 10 of IC7 is shown
incorrectly connected to pin 3. The correct
link should be between pins 10 and 6. The
printed circuit board, however, is correct.

□f;

• I

The TOUCH TEST 20 is a 3 VS digit multi- |„ addition, t
meter with 0.55" LED display and front missing. The
panel touch keyboard for selection of func- circuit should
tion and range. On selecting the desired |f the bu | bs c
function the least sensitive range is automati- th i s can be r6
cally displayed. Optimum display reading is f| op between
obtained by touching the decade touch pads.

The selection of function and range is indi-
cated by an audible tone and illuminated
LED at each pad. In addition to the usual
multimeter functions, the TT20 measures
capacitance (1 pF -200pF); temperature
(—40 to +150°C); conductance (0.01 nS — I

1.999 n Siemens); diode test and audible I

Voltage ranges are 10 pV - 1 kV DC (0.2%);

lOpV - 750 V RMS. T V to I

Current 0.01 pA - 10 A DC and lOpA- \

10 A AC.

Resistance 10 Mohms — 20 Mohms .

The TT20 is available as mains only or re-
chargeable battery mains, it measures only
2.9 x 6.3 x 7.5" and weigths less than 31 bs

l overlay of the article pub-
ser 80 IE66, p. 10-05). These
d 180° degrees.

If the bulbs do not fade on and off smoothly,
this can be remedied by placing a D type flip-
flop between N6 and IC3, as follows:

r I...

L

■O

:

. i •

'4 74 74

i

i

■L