aerial amplifiers yes or no? and, if so, how? digisplay logic levels on a scope. 58 February 1980 up-to-date / v-i - iVU- 1 <■ ■ for U.K.55p. U.S.A | Can. $1.75 february 1980 volume 6 number 2 selektor page 2-04 The aerial booster can be provided with a frequency selective input filter, giving this result. Alternatively, omitting this filter enables amplification over the entire 80 . . . 800 MHz range. aerial booster The design targets for this novel aeria noise, ample gain, wide dynamic range. narrow p FET opamps in the Formant At several points in the Formant mu FET source followers are used as h buffers. This type of stage is not alway an alternative is well worth consid amplifier with an FET input stage. page 2-34 . . , . , . . TV interference suppression Analogue delay Annoying interference orT a TV rec technology is becoming among other things, by local transmit! ever more complex and ,his can be dealt wi,h in a ,air| v sim P |e ever more 'powerful'. This chip, for instance, coming soon contains four split-elec- trode CTDs. Together, elektor vocoder (2) they perform a mathe- Last month, we explained the basic pi matical operation that shouw^be^ie™ how tlwunft^orks* can prove of fundamental And that is what this article is aboi importance in speech boards, full constructional details and t recognition and synthesis. aerial amplifiers Aerial amplifiers are frequently used tusic synthesiser circuit, high-impedance output ays the best solution and irinciples of the Elektor and circuits given, it [, many advocates insist improved performance, vestigate the problems digisplayiA. One of the wii digisplay We pi page 2-44 The integrated circuit used in the digital thermo- meter was originally intended for use in freezers. Only a few other components are required for thi unit, and mounting the 1C underneath the LCD display makes for a very compact construction. analogue delay technology Bucket brigade delay lines are nt mown to Elektor verb unit and for a :hed the surface of This month's cover clearly illustrates one of the better-known uses of a vocoder: making musical instruments 'talk' or 'sing'. However, this by no means exhausts its possibilities, as those who build the Elektor vocoder will soon discover! elektor extending the % GHz counter counter (Elektor. June 19781. The two here - leading-zero blanking and perioi independent circuits. This means that t digital thermometer The AY-3-1270 (General I tended for use in ct digital thermo- iid crystal (LCD) market advertisers index elektor february 1980- ARE BACK 1 I INI BUSINESS' I DORAM Electronics Ltd., a name well known in the home electronics market, are back in business under new management. We aim to combine our many years of experience supply- ing kits and components worldwide with personal service to our new customers. ELEKTOR PROJECT PACKS Touch Tuning (79519) Gate Dipper (79514) Digifarad (79088) Variable Fuzz box (9984) Topamp OM931 (80023) Topamp OM961 (80023) Topamp Power Supply (OM931) 8 ohm Topamp Power Supply (OM961) 8 ohm Ring Modulator 80054 Guitar Frequency doubler 80065 £17-80 Colour Generator 80027 £15-95 Sound Generator (80009) £25-10 Simple Sound Effects (79077) £ 6-55 Digital Tuning Disp 80021 £16-85 Oscillographics (9970) £31-75 Sinewave Generator (79019) £22-45 Cackling Egg Timer (9985) £20-80 Electronic Nuisance 80016 £10-00 Guitar Preamp (77020) £ 5-45 Metal Detector (9750) £19-70 £1845 £ 5-90 £36-30 £11-50 £ 8-90 £ 7-20 £ 3-85 £ 6-22 £11-45 Consonant (9945) £34.65 Preconsonant (9954) £5.75 Luminant (9949) £20.70 Elektornado + H/S (9874) £19.50 Moving Coil preamp (9911) £19.50 Touch Dimmer (78065) £6.80 Ioniser (9823) £9.55 Pools Predictor ( 79053) £8 15 Kirlian Camera (4523/9831) £18 40 Magnetiser (9827) £4.90 Elekdoorbell (79095) £22.00 Video biofeedback (9825-1-2) £19.75 Simple Function Generator (9453) £27.70 Equaliser ♦ Slider pots (9832) £17.70 Steam Train 80019 £ 6-50 Mini Counter (9927) £25.65 Nicad Charger (79024) £14,25 Lab Power Supply 2.5 Amp (79034) £33,45 Lab Power Supply 5 Amp ( 79034) £41.65 Piano * Keyboard £259.00 SC/MP MICROPROCESSOR KITS RAM I/O <9846-11 CPU Board (9851) Memory Ext. <98631 BUS Board * Sockets <98571 HEX I/O (9893) 4K RAM 19885) Power Supply <9906) Cassette Interface (99051 ASCII Keyboard (9965) Elekterminal (99661 ELBUG E proms Basic Micro (790751 Micro Interface (79101) TV Modulator (9967) £27.60 £57 45 £49.45 £11.65 £52.65 £104.45 £14.45 £46.30 £69.55 £37.00 Topamp preamplifier £34-30 Shortwave Converter 49 or 11M £9-95 Auto light Dimmer (80029) £10-35 Parametric Equaliser Filter (9897-1) £645 Parametric Equaliser Tone Control (9897-2) £5-20 Universal Digital Meter (79005) £14-65 SEWAR 80009 £18-45 Audio Analyser (9932) £14-80 Matrix Printer 80066 £57-90 Matrix Printer Unit EP702A £68 65 Timer Controller 79093 £29-50 +SEND FOR OUR FREE PRICE LIST+ Buying one of our PROJECT PACKS will save you the frustration of tracking down those evasive com- ponents that hold up the completion of your project. We can usually supply for all new projects as soon as they are published. Our packs include EPS circuit boards and all the components listed in the article together with sockets and solder. Cases can be supplied as extra items if required. TV GAMES COMPUTER This TV Games Computer is designed around the Sig- netics 2650 microproces- sor and enables you to de sign and play games of your own invention rather than rely on pre-programmed modules. Once a game is designed it is stored on cassette tape for future use. Features of the system include: -f-Joystick and Keyboard control 'Sound Effects ♦Score display Full kit includes all electrc ♦PAL Video output +8 Colours ♦Object size control tic components and V ACCESS ordering service now available. Place your order by telephone or ring our 24 hour Answering Service. HOW TO ORDER Send a cheque or postal order to Doram Elec All our prices include VAT. please add 40p fc ham (0760) 21627. Telex 817912 Doram G. a de boer company RCA Satcom ill Satellites can provide a better quality, more reliable and more economical alternative to land-based facilities for long haul and multipoint voice, data, facsimile, radio and television communi- cations. These communications can be transmitted not only to remote areas but to a virtually unlimited number of locations at the same time — all at service rates lower than existed before the development of domestic satellite technology. The RCA domestic communications satellite system is the first of its type to provide such a broad range of commer- cial satellite services in the United States. On December 1 2, 1 975, RCA launched the first of its satellites - Satcom I - beginning a new generation of commu- nications spacecraft. Satcom II followed on March 26, 1976, and Satcom III on December 6, 1979. Each RCA satellite is capable of serving the 50 states with a wide range of communications services for government, business and the media. Satcom III. however, is fully utilized by the cable TV industry. The spacecraft are controlled from tracking, telemetry and control earth stations at Vernon Valley, N.J. and South Mountain, California. The RCA Satcoms are basically repeater stations, receiving signals from various earth locations and beaming the signals back down to about 1,400 receiver antennas. Without the spacecraft, thousands of miles of ground cables and microwave links would be required to perform the same task. If not used for TV, each of the satellite's 24 channels can carry 1,000 voice circuits or 64 million bits per second of computer data. The RCA Domestic Communications Satellite (RCA Satcom III) is a 24-chan- nel spacecraft to provide commercial communications to Alaska, Hawaii and the contiguous 48 states. Each channel carries 1,000 voice grade circuits, one FM/color TV transmission, or 64 million bits per second of computer data. The spacecraft was placed into a 22,300 mile geosynchronous orbit by a Delta 3914 launch vehicle. With solar panels deployed, the satellite spans 37 feet. The spacecraft main body measures 5'4" x 4’ 2" x 4'3”. The three-axis stabilized spacecraft is equipped with the power, attitude con- trol, thermal control, propulsion, struc- ture and command, ranging and tel- emetry necessary to support mission operations from booster separation through eight years in geosynchronous orbit. Spacecraft life, with continuous full power, is designed to be eight years. Communications payload The RCA Satcom III communications capability is provided by 24 powered TWTA (Traveling Wave Tube Amplifier) channels and four redundant TWTA channels. RCA Satcom I and II provided only the 24 powered TWTA channels. RCA Satcom Ill's four redundant TWTA channels can be switched to replace any of the 24 powered TWTA channels which may become unusable over the life of the spacecraft. The 24-channel communications satel- lite payload consists of a fixed, four- reflector antenna assembly with six offset feedhorns, lightweight transpon- ders, high efficiency TWTAs, and low density microwave filters. Rigid mount- ing of the antennas maintains alignment and eliminates risks associated with deployment. The RCA-developed, 2-02 - ele graphite-fiber, epoxy -composite material for microwave filters, waveguide sec- tions, and antenna sections achieves ultra-light weight while retaining stan- dard electrical designs for critical elements. Frequency and polarization interleaving of the separate channels is employed with the transponder and four antennas to achieve 24 channels, each having a 34-MHz usable bandwidth within the 500-MHz allocation. The dielectric antenna reflectors employ orthogonal conducting grids such that the em- bedded wires provide cross-polarization isolation which doubles the channel capacity by permitting frequency spec- trum reuse within the permissable band- width. The four-reflector antenna assembly provides general coverage of the lower 48 states and Alaska, with a spot-beam coverage of Hawaii. The narrowband command and telemetry channels use the edges of the allocated 500-MHz band on both the 6-GHz uplink and 4-GHz downlink. Structure The spacecraft mainbody, measuring 64" x 50" x 51 ", mounts all electronic boxes, batteries, propulsion and attitude control equipment on three, honey- combed, structural pallets. All transpon- der components are mounted on a south pallet (that side of the spacecraft oriented parallel to the orbit plane pointing south in the operational mode), and all the housekeeping equipment on the opposite north pallet. A third earth-facing pallet provides a mounting surface for four communi- cation antenna-reflectors with their separate composite feed assembly, two command/telemetry antennas, and the earth sensors for attitude sensing. The two sides between the equipment pallets and earth-facing pallet provide shear stiffness for the mainbody struc- ture. Integrated with these assemblies are four spherical propellant tanks. The 915-pound kick motor is housed in the center column of the spacecraft through the sixth side of the mainbody. A coni- cal adapter attaches the motor to the cylindrical column and also provides transition support from the launch vehicle interface to the baseplate struc- Attitude control The attitude control subsystem employs a sealed, high-speed (4,000 rpm) wheel with a separate earth sensor and closed- loop magnetic roll control. The RCA- designed Stabilite attitude control system provides three-axis control by virtue of the gyroscopic rigidity of the wheel and its servo-controlled exchange of angular momentum with the space- craft mainbody. The inertial stability permits attitude determination by a single, roll/pitch, earth-horizon sensor without the com- plexity of a yaw gyro or star sensor. Continuous control of the pitch axis alignment to the orbit normal is achieved by magnetic torquing with no expendables or moving parts. The system maintains orientation during normal orbital operation, orbit adjust, and the acquisition and injection maneuvers. The pointing capability during normal operation is ± 0.21 degree about roll, ± 0.30 degree about yaw, and ± 0.19 degree about pitch. The spacecraft has 12 hydrazine thrusters in a closed-loop system for North/South, East/West station-keeping. During a period of approximately 7 minutes every 3 weeks, this loop with its rate gyro will be energized to modu- late the North/South stationkeeping thrusters and compensate for residual thrusters misalignment or mismatch to maintain attitude control. Thermal control subsystem A Thermal Control Subsystem provides control of heat absorption and rejection to maintain all components of the spacecraft within safe operation tem- peratures, which range from 10 to 30 degrees Centigrade. Space-type mirrors and thermal blank- eting insulation are employed to provide passive heat control. Layers of aluminized insulating material offer high resistance to radiant heat flow. The highly reflective mirrors maximize heat rejection and minimize heat absorption. Under changing conditions of season and array life degradation in orbit, the battery temperature is maintained be- tween 0 and +10 degrees Centigrade for maximum battery life. The Power Subsystem consists of two bi-folded solar array panels and three nickel-cadmium batteries. The sub- system delivers a maximum output of 740 watts of regulated 35 volts at begin- ning of life and 550 watts after eight years. During the two eclipse periods that are experienced each year, power will be supplied by the batteries. Sun- oriented solar arrays and a direct array- to-load connection maximize the efficiency and minimize the weight of the electrical power generation, storage and regulations subsystem. With the spacecraft mainbody always aligned vertically, a single-axis clock- controlled drive shaft maintains the array toward the sun. Solar cells, which convert the sun's energy into electrical power, cover an area of 71.5 square feet. Input converters in each subsystem con- vert the 24.5 to 35.3 voltage range to their specific requirements at constant power and efficiency. These converters, including one in each of 24 Traveling Wave Tube Amplifiers, are designed to preclude a major single-point failure Propulsion subsystem An on-board propulsion subsystem is designed to maintain the spacecraft on RCA Satcom domestic communications satellite (RCA Satcom III) mission objective: Provide commercial communications to Alaska, Hawaii and the contiguous 48 states launch information: launch site: Air Force Eastern Test Range Cape Canaveral, Florida launch vehicle: Three-stage Delta 3914, with nine castor, solid-propellant, strap-on motors orbital elements: circular: Geosynchronous, 22,300 miles above the equator period: 24 hours inclination: Equatorial, zero spacecraft information: 37 feet with solar panels extended main body measurements: 5'4" x 4'2" x 4'3" weight: 2,050 pounds stabilization subsystem: Three-axis stabilized, earth oriented spacecraft design life: Eight years station throughout its eight-year life. The RCA Satcom III carries 216 pounds of hydrazine monopropellant in four tanks for in-orbit use. Upon command from the ground, selected thrusters can be fired to provide spin-axis control in the transfer orbit, as well as velocity control in synchronous orbit. The hydrazine reacts with a catalyst to pro- vide the energy thrust from the twelve reaction engine assemblies. The passive surface-tension propellant feed ensures operation with no risk of bladder deterioration. Two independent, cross-connected half-systems are de- signed to maintain control, even in the event of failure of any thruster, valve or Maintenance of the station longitude and equatorial orbit inclination to 0.1 degree requires about 21 minutes of thrusting once every 3 weeks. An apogee kick-motor uses a solid- propellant fuel to provide the 2,000 pound transfer orbit thrust capability. A dual-squib igniter is designed to ensure reliable in-orbit firing. Command, ranging and telemetry subsystem The functions of command, reception. decoding and distribution, along with automatic and manual telemetry and transponding range tones are handled by the command, ranging and telemetry subsystem. Command signals are modulated on a 6.425-GHz carrier and received by one of the spacecraft's two omni antennas. Each of the two command receivers produces three isolated outputs con- taining the Frequency Shift Key (FSK) command tones. Two outputs from each receiver are sent to the dual com- mand logic demodulator for further processing and conversion to a digital bit stream. Logic level commands are distributed to the spacecraft from the demodulator. Other commands, such as thruster driver, relay closure and pyrotechnic firing, are generated in the central logic processor. The processor has the capa- bility to implement 160 redundant com- mands. During attitude maneuvers, the pro- cessor provides an interface between the thruster firing commands and the actual operation of the thrusters. The ranging function involves the use of the two command receivers and two beacon transmitters. The telemetry function is performed by the dual telemetry mode. This unit samples each of 128 analog telemetry points at 64 frames per second. The sampling is controlled by counters within the module. The telemetry points are available for storing house- keeping data, sync and spacecraft identi- fication. The two beacon transmitters with car- rier frequencies of 3701 and 4199-MHz can operate at two selectable output power levels. The high-power output level is used continuously during launch and transfer orbit operations and prior to earth orientation in synchronous orbit. The low-output power level is used during geosynchronous mission operations. (521 S) 2-04 - elektor february 1980 jrial booster An aerial booster that scores on all of the above points is rarely found, as some of the performance requirements are conflicting. For most contemporary transistors it is necessary to make a compromise between low noise content and good power handling; all too often, the large currents involved also produce high noise levels. However, while the Elektor design team were searching for a solution to this problem, semiconductor R & D engineers have come up with high frequency transistors that remain sufficiently 'noiseless' when subjected to large currents. The devices around which the circuit of this article is built, namely the Siemens BFT 66 and BFT 67, are particularly efficient in first amplifi- cation stages such as aerial boosters - say no more! The favourable character- istics of these transistors will be further exploited by giving the booster a dual function; for wideband operation the booster is made to work with a high current to prevent overload at high input levels, whereas a lower current is employed for narrowband operation. a high performance amplifier, their outlay does not weigh so heavily on the constructor's budget. To be able to profit fully from the improved characteristics of the BFT 66 and BFT 67, the makers publish data sheets including application examples which can be used as a starting point for the amateur and which considerably simplify construction. The circuit for a single stage amplifier is shown in figure 1 , while figure 2 shows a two-stage circuit of extended bandwidth. Respective performance details are given in the graphs of figures 3 and 4. The latter circuit gives more uniform figures for noise and gain over the full frequency band from 25 MHz to 1 GHz. For the single stage amplifier, gain decreases and noise increases with rising frequency. Around the 100 MHz mark, however, its noise content is clearly lower and the gain higher than for the amplifier of figure 2. Measure- ments taken for the circuit of figure 1 at a frequency of 800 MHz proved to be 15 dB for gain and less than 2 dB for noise, verifying that the single stage aerial booster effective frequency range 80 . . . 800 MHz The design targets for this novel aerial amplifier were; low noise, ample gain, wide dynamic range, wide frequency range and last, but by no means least, the possibility of using the same printed circuit board for both wide and narrow passband versions. 80022 - 1 Aerial booster performance largely depends on the characteristics of its active elements It is self-evident that high quality transistors are absolutely essential in the design for a high quality booster, to skimp on this specification is false economy. Although passive elements are equally important in the construction of amplifier will behave satisfactorily in most applications. The BC 177 transistor in figure 1 serves merely to stabilise the collector supply of the BFT 66 to a working level of approximately 6.5 V to give a collector current of 3.7 mA. The graphs of figures 5 and 6, taken from the aforementioned data sheets, show the noise and intermodulation I behaviour respectively. Figure 5 , 80022 2 ge wide band aerial signal booster circuit aerial booster eleklor febr 1 980 - 2-05 represents noise level at frequencies of 10 and 800 MHz as a function of source resistance, with collector current as a parameter. With a resistance of between 50 and 75 S2 and a collector current of 10 mA, the noise level is below 3 dB - even at 800 MHz. The graph of figure 6 shows the intermodulation ratio as a function of the collector current. For this measurement, two input signals capable of giving an output of 180 mV are applied. Intermodulation ratio is then defined as the difference in level, expressed in dB, between the input signals and the amount of intermodu- lated signal product at the output. In the range 2.5... 10mA this ratio continuously increases with current and finally maximises to about 60 dB at 10 mA. Further increase in current will not improve on this figure, which is an indication of the large signal handling capacity of the circuit. An output level of 180 mV can hardly be expected anywhere but in close proximity to a transmitter and is, at any rate, considerably more than most receivers can handle. In narrow bandpass or single channel amplifiers using a BFT 66 the collector current can, therefore, be set to below 10 mA. For wide band amplifiers, however, it is recommended that a collector current of 10 mA be used to obtain the full 180 mV (105 dB pV) output level. Circuit description The circuit consists of a single stage amplifier (a BFT 66 transistor) and will operate anywhere in the 80 to 800 MHz band. Its gain and noise characteristics approach those indicated by the graph of figure 3. Since it was initially intended to function over a restricted bandwidth, the standard version of the circuit as shown in figure 7 features a frequency selective input network (C6,C7, LI and C8). The actual values of the filter components for five different bands are listed in table I. Without this network the circuit will function as an aperiodic amplifier over the entire 80 to 800 MHz range. This standard circuit can be powered from a 16 ... 21 volt DC source via the co-axial cable centre conductor. The HF signals are blocked by inductor L3. The supply voltage is stabilised by IC1 to between 11.5 and 12.5V thereby fixing the transistor working point. Quiescent collector current is deter- mined by resistor R3 while L2 consti- tutes the HF collector load. Capacitor C3 provides HF decoupling. Transistor current is set by the base bias resistors R1 and R2, and stabilised by negative DC feedback via R2 from the junction of R3/L2. Wiring and inductor construction Due care and cleanliness are essential for mounting the components to the printed circuit board (figure 8). As is the rule for all HF circuitry, live HF conductors like those joining C6, Cl, transistor T1 and C2 must be kept short as possible. The construction of the inductc are identical and are wound on a ferrite bead of ferroxcube material, as used for HF suppressors, 5 mm long and 3.5 mm in diameter, with a 1.5 mm bore. Five turns of 0.2 mm diameter enamelled copper wire are led through the hole and wound toroidally around the bead as illustrated in photograph 1. Inductor LI is of the air core type; an Setting up Trimming capacitors C7 and C8 are used to tune the input filter to the passband required. Initially, C7 is set for minimum capacitance and C8 approxi- mately midway. The receiver is then tuned to a weak transmission, preferably halfway across the band. C8 is then adjusted for best reception; this can be achieved by obtaining a maximum reading on a signal strength meter, a minimum of noise in the audio output, or a good quality TV picture. Reception is then tuned to perfection by means of Photograph 2. Typic Precise tuning is carried out gradually by fine adjustment of C8 for possible improvement. If C8 requires a new setting, C7 will also require some correction. These alternate steps should be continued until no further improve- ment is noticeable. Final criterion for correct tuning is signal-to-noise maximisation in audio reception or optimisation of picture Inductor LI number of turns silvered copper wire on 8 mm dia mandrel Trimming capacitors C8, C9 FM (lOOMHzl wirnSla 1 mm green marker 2.. 22 pF 2m 1144 MHz) wire dia 1 mm yellow marker 2. . lOpF VHF (200 MHz) wire dia 1 mm grey marker 1.2.. 6 pF 70 cm (432 MHz) wire dia 2 mm grey marker 1 .2 . . 6 pF UHF (600 MHz) wire dia 2 mm (4 mm dia mandrel) grey marker 1.2 . .6pF Photograph 1 . Looking at finished L2 to L5 inductors: five turns of 0.2 mm enamelled copper wire toroidally wound on a choke type ferrite bead. quality for TV reception. A signal strength meter, if available, can be used for the initial settings, but final tuning can best be accomplished by adjusting for a minimum of noise. Modifications and further applications So far we have dealt only with narrow band operation. For wide band operation, components C6, C7, LI and C8 become redundant. The function of input capacitor is now assumed by Cl. Input connection can now be made where LI joined Cl. With these modifi- cations the amplifier can be used over the entire 80 to 800 MHz range. It can be made to operate as low as 10 MHz, however, by substituting all 1 nF capacitors with 10 nF ones. As previously mentioned, the booster can be powered via the aerial cable. A suitable circuit to adapt the power supply at the lower end of the cable is shown in figure 9a. High impedance inductor L4 prevents HF signals from being grounded, capacitor CIO serves as a HF decoupler, while C9 separates the power supply line from the tuner input circuitry. Inductor L4 is identical to L2 and L3: 5 turns of 0.2 mm enamelled copper wire through a ferrite bead. Figure 9b shows the circuit for a separate DC supply if the receiver power unit can not be used. This supply will feed up to six boosters. If it is decided to mount the power supply near the booster(s), inductor L3 becomes redundant, and the supply is connected directly to R4. The booster is intended to match inputs and outputs rated at an impedance of 60 S2 (not less than 50 S2. not more than 75 £2). Cables with impedances of 240 ft, such as flat twin, will require some method of impedance matching. Commercial balanced - to - unbalanced adapters can be used, of course, but home made constructions will perform just as well and are less expensive. The construction of such a 'balun' is illustrated in figure 10a. A 240 ft balanced aerial can be matched to the booster input with the aid of a co-axial cable loop whose actual length is half the wavelength of the required signal, multiplied by a reduction factor of about 0.7. The loop lengths for various frequency bands are given in table II. A booster-to-downlead unbalanced-to- balanced transformer construction is shown in figure 10b. Inductor L2 is now made to function as a 1 to 4 impedance transformer. Two 0.2 mm diameter enamelled copper coils are wound through a single ferrite bead, 3 turns for the primary and 6 for the secondary. Inductor L4 makes the DC ground connection for the twin lead. When connecting the downlead to the booster, correct DC polarity must be observed. Figure 10c shows the circuitry required to power the booster via a 240 f l flat twin cable. With a power supply incor- porated, these components, together with C2, L3 and L4, are unnecessary. M At several points in the Formant music synthesiser circuits (described in Elektor, May 1977 .... April 1978), FET source followers are used as high- impedance output buffers. This type of stage is not always the voltage, with respect to the input top view voltage. This must be compensated in one of the following stages. Figure 1c. Pinning of the LF 356H. 4. The dynamic range is relatively small. 5. The gate-source bias voltage is tempe- rature-dependent and therefore the drift. The other disadvantages affect the output voltage tends to drift. construction (making it more compli- These disadvantages are not so serious cated and time-consuming) rather than - they don't limit the Formant's the quality. When the Formant circuit was devel- oped, FET Opamps and especially the ~\q MLY 'high-speed' versions were practically V non-existent. The only economical bf 245 alternative was to use standard Field- a ,/T v g effect transistors in the well-known V \v3ns source follower circuit (figure la). . — »r - As those who have heard the Formant Cl will know, this solution works. How- j rj ever, there are certain disadvantages. (5)— • 4— 1. The amplification achieved is not “ A precisely 1 , but slightly less (approxi- mately 0.9). 2. Because of tolerances in the FETs, Figure la. A source folk the source resistor has to be selected N-channei-fieid effect tr carefully. This stage serves as a hig 3. The gate-source bias voltage (UqsI the Formant Interface, V causes a certain 'offset' in the output FET (vamps In die i K>niinnt best solution and an alternative is well worth considering: an operational amplifier with an FET input stage. This article will examine the use of these more up- to-date components and will give a description of ways to adapt the VCO's. potential as a musical instrument. Nevertheless, it is better to avoid them altogether by replacing the FET source follower circuit (figure la) by a voltage follower circuit, using an FET opamp (figure 1b). All source followers in the Formant (in the Interface, VCO and VCF circuits) can be eliminated in this way. When is it worth it? One of the FET source follower's greatest drawbacks is its temperature In a VCO, in particular, the temperature drift should be reduced to a minimum, because when several VCO's are used together any mistuning is immediately audible. As far as the Interface is concerned, temperature drift may cause the entire circuit to be out of tune, which can be heard when it is played together with other instruments. In practice, this is rarely a problem and so there doesn't seem to be much point in converting this module to FET opamps. Only if you're dealing with a keyboard compass of more than 5 octaves (and 2-10 - elektc 1980 FET opamps in the Formant therefore a greater dynamic range), it may be advisable to use FET opamps instead of source followers. The low slew-rate requirements in the Interface mean that economical FET opamps (TL084 and TL074) can replace source followers T1, T3 and T4. The fourth opamp can take over the function of one of the 741s (ICS or IC6, for instance). All these changes will involve a lot of 'flying wires'! Once the circuit has been modified, the offset adjust- ment |P4) must be repeated. In the VCF, the FET's have no real effect on the temperature stability, so that little would be gained by modifying FET opamps in the VCO VCO's which are already in use can easily be converted. However, their oscillator and curve shaper will have to be realigned. For this reason, the modi- fication is only advisable if the fre- quency stability is still not good enough — even though it is high compared with many other synthesisers. Figure 2a shows the original circuit, which has two source, followers. Of these, only T2 affects the oscillator's frequency stability; the simplest con- version, therefore, will entail replacing this FET by a voltage follower using an LF 356H. The rest of the circuit can remain unchanged, as shown in figure 2b. However, the oscillator and the curve shaper must be re-adjusted, since this modification will alter both the amplitude and the DC level of the sawtooth. Figure 3a gives the modified com- ponent layout for the circuit shown in figure 2b. Connections 1 and 5 of the metal-case version of the LF 356 (IC12) are not used and these wires can be cut short. Connection 6 is soldered to 2. R17 and T2 are unsoldered from the VCO board and IC12 is mounted, as shown in figure 3a. If a new VCO module is to be built from scratch, more extensive changes may be considered. Figure 2c shows the new circuit: FETs T2 and T3, source resistors R17 and R20, gate resistor R16, and trimmer P10 have all been removed. The voltage follower opamp (IC12) replaces both source followers. Resistor R18 is changed to 470 ft. A metal film resistor is not strictly neces- sary. If the 'ultimate' in temperature stability is required, though, a 470 W2% metal film resistor should be used. Figure 3c shows the modified com- ponent layout for new VCO's. IC12 substitutes T2/R17; a wire link to the left of it replaces T3/R16. The FET opamp in the new oscillator circuit, as shown in figure 2c, not only improves the frequency stability, but also reduces the component count in comparison with the original circuit. Furthermore, the removal of preset PI 0 makes the adjustment that much easier. M Figure 2a. The oscillator circuit in the VCO contains two FETs used as source followers; only Figure 2c. When building a new VCO module, it is worth considering this simplified oscil- lator circuit. The FET opamp now replaces both source followers, and preset potentiometer P10 can be omitted. 3a 2-12 - elektor 1980 TV interfer In actual fact, it is not always the fault of (amateur)transmitters that they cause interference on TV sets. As a rule, it is the 'broad-band aerial amplifier' in- cluded in the TV set's aerial system which is at the root of the problem. Broad-band amplifiers have the dis- advantage of being rather indiscriminate. They pick up and amplify everything, including signals which are not meant for them at all. When powerful broad- cast, amateur or mobile transmitters are around, the voltage in the aerial ampli- fier rises to such an extent that the amplifier becomes completely 'jammed' and this makes a clear reception of TV signals very difficult. the broad-band amplifier, is stripped at a certain point and connected to one end of a piece of coax. This coax, believe it or not, is the filter. It should be exactly ’A wave length of the signal that is to be eliminated. The other end of this piece of coax, which is known as 'A X (quarter-lambda) stub, remains open. This is how it works: Radio waves reaching the open end of the Va X stub are reflected. For the un- wanted signal, the stub is exactly %X long, so that the reflected waves have travelled a distance of 2 x Vt X = 'AX by the time they get back to the beginning of the stub. Consequently, the reflected wave is in exact phase-opposition with TV interference suppression Nearly everybody will agree that interference on TV can be extremely annoying. Interference can be caused, among other things, by local transmitters. Usually, however, this can be dealt with in a fairly simple and effective way. So what do you do? Well, after reading the above, it would seem an obvious conclusion that it is probably better to do without an aerial amplifier altogether. For that matter, very often one is included in the aerial system ‘just to be on the safe side', without it being strictly necessary. It is a much better (and cheaper!) idea to simply use a good TV aerial which is a powerful 'amplifier' anyway (and will have a more accurate directional effect and an improved front-back ratio - both important factors). If, on the other hand, you cannot manage without an amplifier, it is advisable to use tuned aerial amplifiers (also known as channel amplifiers). These, being narrow-band, do not pick up unnecessary signals and so interference is no longer a problem. However, if you already have an aerial system which is fitted with a broad- band amplifier, it is rather frustrating to talk about the kind of aerial you should really have. Quite a few interference problems can be dealt with in an inexpensive way by simply inserting a band-stop filter in the broad-band amplifier's input. This elim- inates the interfering signal (produced by an amateur transmitter, for example) before it reaches the broad-band ampli- fier. The so-called %X-filter is a good choice: it is easy to make — all you need is a piece of coax cable! The % X-f ilter Figure 1 shows what the filter looks like. In passing, it should be noted that this filter can be used for all kinds of purposes — not only eliminating inter- ference in broad-band amplifiers! As the drawing demonstrates, the (coax) aerial cable, leading from the aerial to Figure 1 . The filter is a piece of coax, connected in the lead from the aerial to the broad-band aerial amplifier. In practice, it is often best to connect the 54 \ stub at the input of the amplifier. Figure 2. The filter works as follows: The voltage reflected in the stub (2b) is in exact anti-phase to the input voltage (2a), so that the resulting voltage (2c) is nil. TV interference suppr the input signal, so that the resulting voltage is nil. This is illustrated in fig- ure 2. Figure 2a shows the input voltage, figure 2b shows the reflected voltage and figure 2c gives the result. Everything always sounds marvellous in theory, but often turns out differently in practice. Here too, unfortunately this is the case. What happens is that the V* X stub attenuates the reflected wave. Figure 3. A spectrum-analyser photo of a coax Va X-filter for the 2-metre band. The attenuation is approximately 36 dB. Figure 4. The rejection filter intended for the 2-metre band can also be used for the 70-centimetre band, with marginally poorer Figure 5. A spectrum-analyser picture over a much wider frequency range (100 MHz per division) shows that there are many more frequencies at which the input signal and the signal reflected by the filter are in anti-phase. so that the resulting voltage is not completely nil, as shown so optimisti- cally in figure 2c. It doesn't have to be! A reduction by about 30 dB (32 times) is usually achieved with the aid of the filter and nine times out of ten that is enough. Furthermore, the filter not only blocks interference on the wave length which is four times as long as the % X stub, but it also works for wave lengths corresponding to %X, %X, Y«X etc. The input signal and the reflected wave are in anti phase at these fre- quencies as well! coming soon In practice As far as the exact length of the filter is | concerned, simple theory is one thing, practice another. The speed at which radio waves travel along coax is not the same as that in air. For this reason, the wave length inside the cable is shorter than that outside: a radio wave may have a wave length of 3 ft. outside and as little as 2 ft. inside the coax cable. The reduction factor, in that case, is: % = 0.67. Let us consider a rejection filter for a 2-metre amateur transmitter. Amateur transmitters on the two-metre and 70-centimetre bands seem to be prime targets for complaints about inter- ference. On the two-metre band % X corresponds to V* x 2 = 0.5 metres. In order to find out what the exact length of the %Xstub should be, this figure must be multiplied by the reduction factor of the coax. Every manufacturer (and reliable retailer) will be able to supply this information. It is advisable to make the cable slightly longer than the calculated length, so that once the stub has been connected, it can be trimmed for maximum suppression of the interfering signal. This can be done by cutting off small bits at a time. When you have found the correct length, the %Xstub can be rolled up. It looks neater, that way. One of the characteristics of this type of filter, as mentioned earlier, is that it will eliminate several frequencies. This can be an advantage: a filter for the 2-metre band can be used for signals on the 70-centimetre band as well. The spectrum-analyser photo's (figures 3 and 4) illustrate this. Figure 3 shows how the filter attenuates interference at the frequency for which it was originally intended: 144 MHz (the 2-metre band). Figure 4 illustrates the effect at 432 MHz (70-centimetre band). Since the damping of the coax cable is greater at higher frequencies, the attenuation achieved is less than that at 144 MHz. As the photo's illustrate, the difference is approximately 6 dB. The spectrum-analyser photograph in fig- ure 5 gives an idea of the attenuation over the whole frequency range (hori- zontally 100 MHz per division). M March — Chorosynth: a low-cost string synthesiser — Printer for microprocessors April: car special! — electronics in cars — fuel consumption meter — intelligent wiper delay — electronic ignition — and many others! May — cassette interface for BASIC pP — a practical intercom • 55 - construction and alignment Last month, we explained the basic principles of the Elektor vocoder. From the block diagrams and circuits given, it should be clear how the unit works — once you've built it. And that is what this article is about: the printed circuit boards, full constructional details and calibration procedures. Every effort has been made, at the design stage, to make this a straightforward project for the home constructor; the extensive explanation of the construction given here is intended as the necessary 'software backup! ' let's put one thing right. Last we stated that there were to be printed circuit boards. Wrong: ire fourteen now. The wiring between the twelve original boards was getting so extensive that it was decided to plug them all into a so-called 'bus board' that runs along the back of the case. This board turned out to be so long that it had to be cut in two, for postal reasons. All other boards, with the exception of the power supply, are plugged into connectors on the bus board. This is a great help, both for construction and 'service' - so we hope no-one will complain about the two additional boards . . . Power supply Before getting to the p.c. board layouts, we must first provide the power supply circuit, as promised. As shown in figure 1, this circuit is so simple that it is hardly worth talking about. The symmetrical +/-15V supply is obtained in the easiest possible way, using two integrated voltage regulators (IC19, IC20I. The total current con- sumption is only 200 mA, so the 400 mA mains transformer will be more than adequate. Obviously, a larger transformer could be used, provided it fits in the case: future extensions, if and when they come, can then be powered from the same supply. For biasing the OTAs, a further symmetrical +/-5 V supply is also required. As shown in figure 1b, these voltages are derived from the (stabilised) +/— 15 V supply, by means of another pair of integrated voltage regulators (IC21, IC22). The two tantalum electro- lytics, C86 and C87, and the lOOn capacitors C84 and C85 are essential for this type of regulator: they suppress its annoying tendency to break into spontaneous oscillation. A printed circuit board for the supply is given in figure 2. To be more precise, it only accommodates the circuit shown in figure la; the +/- 5 V supply (figure 1b) is mounted on the bus board. A new feature We owe an explanation, although it is doubtful that many readers will have noticed it! Just before going to press last month, our esteemed 'boffins' came up with a small but very useful extension. It was included in the circuits for the high-pass filter and the input/output module (parti, figures 5 and 6) at the last minute, but we didn't quite get around to explaining it in the text — mainly owing to the fact that we were chasing around, trying to find out whether we were allowed to include it! The trouble was that our beautiful 'find' turned out to be patented — by Bode. We were still trying to find out how this effected us (fortunately, it doesn't) when the issue went to press, with the result that there were a few details in the circuits that remained completely unexplained in the text. This is common practice in industry, of course, but we feel that it is rather below-standard for a self- respecting technical magazine. Our apologies! What extension? In figure 3, part of the high-pass filter is repeated. There's a potentiometer, PI 7, with a series resistor ( R 1 1 7) . When we point out that the lower end of the series resistor is connected to the second input, 'K', of 2-16 - elektc 1980 >der (2) Figure 3. Part of the high-past filter circuit. P17 and R117 are added so that a smell amount of the original speech signal can be added to the final output. This liigh-frequency blend' can be particularly useful when the carrier signal is lacking in high frequencies. the summing amplifier (part 1, figure 6), the basic idea may suddenly dawn. Some of the signal at the output of the high-pass filter (A11/A12) is taken off by PI 7 and added, without 'vocoding', to the final output. In this way, the lack of a voiced/ unvoiced detector and associated noise generator can be camouflaged to some extent. More than 'some extent', in fact: the results can be surprisingly good! When the carrier signal is lacking in high-frequency content, there is not enough 'replacement signal' for the unvoiced 'hissing' sounds in speech (the 's', for instance). In this case, the high frequency components of the original speech signal can be added to the output signal; the correct 'blend' is set with PI 7. In many cases, this vastly improves the intelligibility of the vocoded signal. Provision is made for mounting the potentiometer, PI 7, on the p.c. board for the filter modules. The ground connection and that for the wiper ('f') are both at the edge of the board; the 'hot end' of the potentiometer is connected to a copper pad marked 'x' on the copper side of the board. Resistor R1 1 7 is mounted on the bus board. The connection from the lower end of this resistor to the input of the summing amplifier (points 'k') is included as a copper track on the bus board. Input/output and filter boards We can now do one of two things. Either repeat all the circuits already published last month, in part 1, or else ask you to dig out that January issue and refer to it as required. The latter option seems to be the most sensible. All right, so now we've got part 1 in front of us. A general block diagram of the filter units is given in figure 2, and complete circuits for the band-pass, low-pass and high-pass filter units in figures 3, 4 and 5. respectively. In the accompanying text, it was explained that a modular construction was to be used: one printed circuit board for each complete filter unit. No wild guess, this; in fact, our printed circuit board designer had already come up with a single, universal design for the filter board, suitable for all types of filter: low-pass, band-pass and high-pass. The layout of this universal filter board is given here, in figure 4. Figure 5 shows the com- ponent layouts, with accompanying parts lists, for mounting a band-pass filter unit (figure 5a), low-pass filter (5b) and high-pass filter module (5c). The values for capacitors Cl ... Cl 1 in the eight band-pass filter units are listed in Table 1. This table was also included in part 1, but it is repeated here with the rest of the parts lists. Observant readers may notice that the supply- number frequency BPF 1 265 Hz BPF 2 390 Hz BPF 3 550 Hz BPF 4 800 Hz BPF 5 1200 Hz BPF 6 1770 Hz BPF 7 2650 Hz BPF 8 3900 Hz 210- 320 320- 460 460 - 640 640 - 960 960 - 1440 1440-2100 2100 - 3200 3200 - 4600 Table 1. The values of capacitors Cl ... Cl 1 for from this table. decoupling capacitors (C73 . . . C76, 8 x C77 and 8 x C78, shown in figures 3, 4 and 5 in part 1) are missing in the layouts given in figure 5. Not to worry: they are included on the bus board. Then there's the board for the in- and output module (the circuit shown in parti, figure 6). The copper and component layouts are given in figure 6. This p.c. board is exactly the same size as the filter unit board (70 x 168 mm). For that matter, the supply board (figure 2) is also the same size, even though it is not the intention at this time to mount it as a plug-in module. As before, the decoupling capacitors for the input/output module (C79 and C80) are mounted on the bus board. Now for a closer look at the boards. Mounting the components shouldn't be a problem — provided you don't get the various component layouts for the filter board mixed up. And don't forget the wire links; although they're not mentioned in the parts list, they do play an essential role. All connections to the boards are along the two ends. At one end, the connections associated with front-panel components; at the other end, the connector plug. On the filter boards, this means that the 'front' of the board contains the control voltage connections, U Ci0u t and U c j n (points d and e in the circuits), the LE:D output and the connections for the U c ,in level control (8 x P3, P7, P1 1 ). The 'rear' of the board contains all 'internal' connections: the speech and carrier inputs (points a and b), the vocoded output (point c), the supply connections and, for special applications (to be described later), a second set of control voltage connections (U c ou t and *-*c,in)- Similarly, on the input/output board, the front panel connections are at one end: input and output jacks with associated level controls (PI 3. P14, P15). The 'connector' end is for the supply voltages and the internal in- and outputs a, b, c and k. This system means that each board can easily be built as a separate, plug-in module. A 21 -pin connector is mounted on the ’inner’ end of each of the filter- unit boards and the input/output board (one suitable type is made by Siemens). The front panel is mounted at the other Cl . . . C8 C9 82 n 220 n 56 n 1 50 n 39 n 100 n 27 n 68 n 18 n 47 n 12 n 47 n 8n2 47 n 5n6 47 n CIO 22 n 10 n 6n8 6n8 6n8 the eight bend-pass filter units must be selected end; it contains the control(s). jacks and LED. This construction is illustrated in figure 7: a sketch of a complete filter- unit module. The small (3 mm) earphone jacks shown are a good choice for the input connections. If the 'high-frequency blend' feature shown in figure 3 is to be added in the high-pass filter unit, this will obviously call for a second potentiometer on its front panel. The input/output module also has a more densely populated front panel: it contains three potentiometers and three large-sized headphone jacks for the speech and carrier inputs and the vocoded output. Final assembly Now we come to the job of combining all the separate boards (or modules) into Resistors: R1.R17.R30 = 10 k R2.R18 = 680 n R3.R7.R19 ■ 100 k R4.R20 = 8k2 R5.R21 = 560 n R6.R22 = 82 k R8.R26 . . . R29.R31.R32 = 47 k R9.R10 = 150 SI R11 = 4k7 R12 = 1 M R13.R33 = 22 k R14.R15 - 33 k R16- 15k R23.R24.R25 = 3k3 R34 = 120 k R35 = 1 k R36 = 68 k Capacitors: Cl ... Cl 1 : see Table 1 Cl 2 = 33 p C13 = 180 n C14 > 22 n Semiconductors: T1 = BC 547B T2 = BC 557B D1.D2.D4 = 1N4148 D3 = LED IC1.IC2 = TL084 IC3= 741 IC4 = CA 3080 Sundries: PI = 100 k preset P2 = 25 k preset P3= 10 k lin. P4 = 10 k preset m\m R129.R132.R13 = R 1 26 = 1 M R131 = 47 k R134 = 150 n C47.C56.C66 = 220 n C48 = 100 n C49.C50.C52.C57.C61 ,C62 C64.C65.C67.C68 = 33 p C51 .C53.C60.C63.C69, C70= 10 p/16 Vtantalunr C54.C55.C58.C59 = 39 n C72= 22 p/16 V tantalum Semiconductors: IC13 = TDA 1034NB, N IC14.IC15.1C16. IC18 = TDA 1034B IC1 7 = LM 301 one complete 10-channel vocoder. The constructional block diagram (figure 81 illustrates the principle. It shows all the plug-in modules and the power supply; as can be seen, the bus board is a great help. Without it, the wiring would become rather messy. The letters a, b, c, d, e and k, shown in figure 8, are also included on the various p.c. boards; they correspond to the indications in the circuits given in part 1 . For simplicity, the supply is shown in figure 8 as a single board. In practice, as explained earlier, the +/— 5 V supply is actually mounted on the bus board. PI 7 and R 1 1 7 are also included in the block diagram; they are only required if the high-frequency blend option is to be Also shown in figure 8, enclosed in dotted lines, are the supply connections and two mysterious connection links. Sra£«9l These refer to nine connections on the bus board, into which connector pins can be inserted. At a later date, they will provide an easy way to add a voiced/ unvoiced detector with its associated noise generator. All supply voltages are available in this group, so that the unit can be powered from the main vocoder supply. The connection links between two pairs of contacts are actually those shown in the circuit of the input/output module (part 1 , figure 6) . at the outputs of A31 and A33. The links are already included as copper tracks on the board; when a voiced/unvoiced detector is to be added, these tracks are scratched away so that the speech and carrier signals run through this module. Having said so much about the bus board, it's time to take a look at it — or them, actually: as mentioned earlier, it is supplied in two sections that must be joined by means of wire links. Figure 9 shows the two p.c. boards and their component layouts. As can be seen, there was plenty of room between the eleven 21-pin 'female' connectors to mount the 5 V supply, decoupling capacitors and one or two other odds and ends. One point has not been mentioned yet (nor shown in figure 8, to avoid confusion): beside each connector. there are two connections for the U c ,in and U C O ut control voltages for each filter module. These are included with an eye to possible future extensions. For instance, in a complete system it may prove useful to route the control voltage interconnections through a plug- in matrix board, instead of using loose cables on the front panel. The various modules and the bus board are designed to fit neatly into a module case, as shown in figure 10. A standard 19 inch case can be used, with guide strips to hold the boards. This type of case is available from various manufacturers. The 19 inch width is just right for mounting the eleven modules at the spacing dictated by the bus board — no coincidence, this! The mains transformer and supply board can be mounted on the back plate, as shown in figure 10. A neat way to make the connections between the supply board and the bus board is by using so-called flat cable. For the various signal and control voltage in- and outputs, jack plugs are a good choice; the smaller (3 mm) type for all U c in and U c ,out connections and a larger version (6 mm) for the signal in- and outputs. Flexible cables with a small plug on each end can then be used to make all desired control voltage connections on the front panel. The mains switch, and an LED for power on/off indication, can be mounted on the front panel of the input/output unit. An alternative can be seen in figure 10: a potentiometer with built-in mains switch can be used for the output level potentiometer, PI 5. One word of warning, however: sometimes, the electrical screening between the switch and the potentiometer may prove inadequate — giving rise to an annoying hum. Alignment procedure We assume that everybody still has the original circuits, given in part 1 , to hand; in any event, we will be referring to them regularly . . . There are three preset potentiometers on each filter-unit module that must be correctly adjusted. This means that three separate adjust- ments must be performed for each board, as follows: 1. First the preset that sets a DC bias voltage for the inverting input of the OTA in each unit. In the eight band-pass filters, this is P2; on the low-pass filter board it is P10 and for the high-pass filter it is P6. The purpose of this adjustment is to ensure that the varying DC bias voltage, derived from the )der (2) elektor february 1980 - 2-21 8 Figure 8. Block diagram of the complete vocoder. The indications a. b, c, d. a and k correspond to those given in the circuit diagrams (part 1 ). They are also included on the p.c. boards. control voltage output of the analyser section when a speech input is present, cannot 'break through' to the 'vocoded' signal output. In simple terms: a signal present at point 'e' should not appear at output 'c'. This adjustment is carried out as follows: • The U c _out and Uc, in sockets on the front panel are interconnected by means of patch cords. • All control voltage level poten- tiometers on the front panels (8 x P3, P7 and P1 1 ) are set to minimum, with the exception of the one on the module that is to be set up; that control is set to maximum. • A steady noise signal is applied to the 'speech' input. One simple way to do this is to blow gently into the micro- • The bias potentiometer on that module (P2for a band-pass filter, say) is adjusted for minimum output signal from the vocoder. If measuring equipment is available, a more precise alignment procedure can be considered. Instead of blowing into a microphone, a test signal can be applied direct to the U c j n input of the module; a suitable test signal is a low-frequency sinewave (500 Hz or less), superimposed ! on a fixed DC voltage. The output signal i from the vocoder can be observed on an oscilloscope, and the preset is adjusted for minimum LF output. In some of the modules, it may prove impossible to reduce the break-through to an acceptably low level. In this case, the OTA is almost certainly the culprit: in any batch there will always be a few that have too high a leakage from the control input to the output. The only solution is to replace them. 2. The next step is the preset in the voltage-to-current converter for the OTA: P4 in the band-pass filter units, PI 2 in the low-pass filter and P8 in the high-pass filter module. This adjustment is intended to set the initial point of the control characteristic to the same level for all modules. The procedure is as follows: • A suitable test signal is applied to the 'carrier' input - white noise is a good • A very low DC voltage (approxi- mately 200 mV) is applied to the U c ,i n input of the module that is to be adjusted. This calibration voltage can be derived from the +5 V supply by means of a 25:1 attenuator (a 22 k resistor in series with 1 k, for instance). • The control voltage level control on the front panel of the module (P3, P7 or P1 1) is set to maximum. • The preset potentiometer (P4, P8 or PI 2) is now adjusted so that an output signal just appears at the main output. • If the test voltage proves to be outside the adjustment range of one or more of the modules, the whole procedure can be repeated with a slightly higher or lower test voltage. 3. Finally, the easiest adjustment: PI, P5 and P9 in the band-pass, high-pass and low-pass modules, respectively. These presets determine the DC offset of the active low-pass filter that is the last stage in the analyser section of each module. With no (speech) input signal, each preset is adjusted for minimum U c out voltage of the corresponding module'. In conclusion We've got an interesting photo for you, saved to the last. With a spectrum analyser and a lot of patience, we succeeded in plotting each of the filter characteristics separately and combining them in a single photo. The result of our efforts is shown in figure 1 1 . At the left in the display, the characteristic of one of the two (identical) low-pass filters; this is followed by a neat procession of band-pass filter characteristics and. ttfttwSl finally, the high-pass filter. The minor differences in peak amplitude are caused by inavoidable component tolerances. Not that they have any real effect, in practice, since they can be compensated for by means of the control voltage level controls on the front panel. As can be seen, the filters provide a nicely regular division of the audio spectrum. Their Q is virtually identical, as is apparent from their equal band- pass 'widths' on this logarithmic frequency scale. This is by no means our last word on the subject of vocoders. Exactly what is to come, and when, has not yet been finally decided — so we won't make any promises. Anyway, for the time being all enthusiastic constructors should have plenty to do . . . M are they worth it? amplifiers Aerial amplifiers are frequently used to try to improve the sensitivity of an existing receiver. All too often it is found that any increase in sensitivity is accompanied by an increase in the noise content of the resultant signal. For this reason it becomes apparent that such an amplifier needs very careful design. If the amplifier is only required to compensate for the losses in an aerial distribution network, or the like, then the problems become less severe. Contrary to the opinion of those who claim that aerial signal amplification is of no use whatsoever, many advocates insist that amplification will often lead to improved performance. For a well-founded appraisal of the various 'for and against' arguments it is, therefore, interesting and even important to investigate more deeply into different aspects of the problems involved. This article deals with reception on VHF/FM and UHF/TV wavebands. With equipment that performs satisfactorily on these frequencies there should be no need for further atrial signal amplifi- cation. With a system that consists of a high quality receiver, an effective aerial and a short low-loss cable between them, even the best of aerial boosters will not improve the performance. Not everyone, however, has such an optimum combination. In many in- stances the aerial lead can spoil results by attenuating the signal to an extent depending on the cable quality and length. A coaxial cable of average quality and a length of, say, 20 metres may attenuate the signal by as much as 6 dB. This means that a mere 25% of the signal picked up by the aerial actually arrives at the receiver with consequent deterioration of reception, especially in fringe areas. The above example illustrates the princi- pal justification for the use of aerial boosters: to make up for signal losses between aerial and receiver, such as cable damping and mismatching. Aerial signal amplifiers are sometimes used, or rather abused, to compensate for low sensitivity in existing receivers. In this case they will function as un- tuned receiver input stages. This appli- cation does, however, have its hazards, the most objectional one manifesting 2-28 - ele itself as cross-modulation — offsetting any increase in signal strength. 1 Ways and means The logical application for aerial signal amplifiers is to overcome transition losses between aerial and receiver. A few requirements must, however, be fulfilled to achieve the best results. For one thing, the amplifier must be masthead positioned. It can be powered either by an incorporated supply unit or, via the coax cable itself, from a power unit at the lower end of the cable. Obviously, the best results are obtained by tuning the masthead amplifier. In practice, however, this method is usually ruled out because of the compli- cated layout and the necessity for a separate tuning control. Second best is a bandpass amplifier which operates over a limited band of channels. Incoming signals outside its specific frequency band will be rejected, thereby elim- inating hazards such as intermodulation and preventing powerful radiations from outside the band from squelching or otherwise impeding reception of the desired transmission. These arguments may explain why wide band amplifiers are not the first choice for single band aerials, such as VHF/FM types. Wide band amplifiers may be put to good use in multiple band systems where a number of aerials, each with its own bandpass amplifier, are followed by frequency selective 'splicing' networks. In this instance, a wide band amplifier could be inserted in the common down- lead to compensate for any cable losses (see figure 1). Gain versus noise It is not enough that the aerial booster has a certain gain; its self-generated noise must be appreciably lower than that of the receiver. In order to evaluate this comparison, the magnitude of self- generated noise in an amplifier or receiver is defined by the symbol F. This is the relation between the signal-to- noise power ratio at the input and the S/N power ratio at the output of the amplifier in question. The relation can be algebraically expressed as: P s j = input signal power level Pni = input noise power level P S o = output signal power level P n o = output noise power level. In the ideal case, a 'noiseless' amplifier, the F number is unity. In all other cases it is higher. It is expressed as a number without dimension or in kT 0 units, the numeral in both expressions being the same, for example F = 4 = 4 kT 0 . It is often convenient to express the relation logarithmically in decibels, in view of Figure 1. Aerial signal amplification and splicing network. Each aerial has its own amplifier (A1 . . . A3) active over the selected aerial band. Cross-over network B splices the three aerial signals and applies the sum to a further amplifier C. This last amplifier is a wide band type and makes up for losses in the coax cable and the signal splitting network D. Other losses such as mismatching and plug-to-socket connections are also overcome. Depending on reception conditions, aerial gain and cable efficiency, some or all of the amplifiers may be redundant. Some makes of amplifier have built in splicing networks. the common practice to define power ratios in dB, thus: F(dB) = 10logF( kTo ) F numbers for good receivers are often less than a factor of 5 (7 dB) and for high quality tuners they can vary between 3 and 4 kT 0 (4.8 and 6 dB). To justify their use, the F number of aerial amplifiers must be much better to be of any advantage. For a cascade of ampli- fiers (see figure 2) this works out as fol- G1 • G2 • G3 in which G stands for power gain. This formula shows that the F number of the first amplifier represents the main con- tribution to the overall noise; the effect of the second stage amounts merely to its F2 number divided by the gain of the first stage. Since high gain in the first amplifier stage practically nullifies the influence of noise in the second and third stages, sensitivity and noise of the entire re- ceiving equipment are largely dependant on the quality of this first stage. This means that the performance of a receiver with insufficient sensitivity and noise characteristics can be considerably 'tuned up' by a good aerial amplifier. On the other hand, no improvement can be expected from an amplifier whose F number is the same as or worse than that of the receiver, or whose gain is not high enough to overcome the effect of noise in the receiver. These considerations can be illustrated in the following example. Let us assume that a given receiver has an F number of 5 and that it is preceded by an ampli- iplifiers ary 1 980 - 2-29 fier with an F number of 3. The overall F number will then depend greatly on 2 the amplifier's gain. For an amplifier with a gain of 2 (3 dB) the overall noise still works out to be F = 5 — no improve- ment at all. For a gain factor of 10 (10 dB), the improvement amounts to Ftot ” 3.4. An amplifier gain of 100 (20 dB) brings the overall noise down to 3.04, a figure which practically equals that of the amplifier itself. FI-01 F2-G2 Gain and losses — noise and sensitivity Improvement in noise characteristics does improve receiver sensitivity. The question still remains: is it really worth all the trouble and expense? The usual way to define sensitivity is to state the signal input voltage for a certain demodulated (or for stereo, decoded) output at a given audio signal- to-noise ratio. The signal at the aerial input terminals of the receiver depends not only on the receiver noise figure but also on demodulation method, modu- lation depth, audio frequency band- width and receiver input impedance. Only if all these remain the same, will any improvement in sensitivity and noise characteristics bear any effect. The improvement can be calculated from the formula F r = receiver noise number (kT 0 ) Ftot = overall noise number (kT 0 ) G = improvement in power ratio g = improvement in voltage ratio. Transformation of these equations gives us; improvement in dB = 10logG or 20 log g. Having got this far, what is the effect of this improvement on the final audio signal? The equations show improve- ment in the high frequency S/N ratio with respect to the demodulator input. However, the audio output signal from the demodulator also has a S/N ratio. For amplitude modulated signals this S/N figure corresponds closely to the the input level for both mono and stereo. Figure 3 shows such a graph, in which it can be seen that the S/N ratio at low input levels (around 1 pV) suddenly decreases with an increase in input level. Firstly in a linear proportion and afterwards, from a certain level on, it remains constant. In the given example the upper limit appears to be some 200 pV for mono and 300 to 400 pV for stereo. How can these figures be translated into actual receiver performance? When reception of an FM signal is weak, any small improvement in signal level from aerial or amplifier can result in an appreciable improvement in the audio signal-to-noise ratio. This improvement will not be quite so spectacular with high FM signal levels. This means that the S/N increase in high quality equip- ment is hardly worth the additional amplification; in mediocre equipment the S/N increase is more effective. It is, however, very likely that the latter type of receiver falls short in other respects too, such as selectivity and fidelity. Under these circumstances it seems a better proposition to invest effort and money into better equipment. If an existing receiver or tuner performs and the addition of an amplifier would appear to be a more convenient and effective solution. Compensation for cable losses Losses in coaxial cables determines their quality and may differ between various makes. As a rule-of-thumb the larger its diameter, the better its charac- teristics. As can be seen from figure 4, coax cable attenuation increases with frequency. For commercially available types, the attenuation at 200 MHz may be anything between 4.5 to 45 dB/ 100 m, a figure of 25dB/100m being typical for inexpensive run-of-the-mill general purpose coax. Special quality cables marketed as ‘low-loss' types may show attenuations of some 1 2 or 15 dB/ 100 m. To these coax cable attenuations a figure must be added for inevitable (small) mismatches. Obviously, the sum of these losses adversely effects the sensitivity and noise characteristics of the whole receiving system, which can not be made good by increasing the receiver gain only. This effect can be put down in figures by considering a number of amplifiers, represented by the well known 'black boxes', connected in cascade. The black box which represents the cable has a noise figure of nearly unity and a 'gain' figure 'D' that is negative and stands for the attenuation. From this, it follows that the equation for cable plus receiver can be repre- The overall noise for the cascaded masthead positioned amplifier, cable and receiver is given by the equation F r — 1 F tot = F a + qJTd where F a is the noise figure of the amplifier and G a its gain. This equation demonstrates that in the masthead con- figuration the overall noise is deter- mined by the noise and power gain of the amplifier, just as in the case without HF S/N ratio. This is not so in the case of FM signals, especially when the input signal is on the high or low side. Technical data sheets of high quality stereo receivers often include a graph to indicate the S/N ratio as a function of satisfactorily, especially as regards S/N behaviour, but its IF gain is not quite up to scratch, then optimum sensitivity can be achieved by employing a good ampli- fier. In spite of slightly reducing the overall S/N ratio, this will supply the higher incoming signal level required to 'fill up' the demodulator. Although, in this particular case, it may be possible to increase the IF gain, this procedure may be a relatively clumsy undertaking 2-30 - 1 aerial amplifie -igure 3. This graph sho\ aval, as a function of th< igura of 3.5 kTo. Tha gr cable losses. The overall performance differs merely in that the effective amplifier gain has suffered because of cable attenuation; it now amounts to G a • D, If amplifier noise is less than receiver noise, and the overall gain is sufficient, cable losses can be com- pletely eliminated, the overall noise figure dropping below that of the With the amplifier positioned at the lower end of the aerial cable, its ben- eficial effect will be considerably inferior. The noise equation then becomes F a - 1 F r - 1 F tot = 1 + o + G a • D which shows that the detrimental effect of cable losses is maintained to the full. Numerical examples Figure 5 compares different configur- ations of the same components, namely: • an FM stereo receiver with factor of 3.5 and a sensitivity in accordance with figure 3, measured with a sweep of ± 40 kHz and a bandwidth of 180 Hz to 16 kHz; • an aerial amplifier with a r ber of 1.5 and 20 dB (100 times) power gain; • a cable with 6 dB (factor 0.25) attenuation. The following configurations are shown; 1 : receiver without cable or amplifier, 2: receiver without cable but with amplifier, 3: receiver with masthead amplifier and cable, 4: receiver with cable and amplifier at lower end, 5: receiver with cable but without amplifier. Table I lists the figures for overall gain in dB, aerial signal level for 60 dB stereo S/N, and S/N ratio for 100/iV aerial signal for each configuration. The conclusion is that in the absence of an aerial cable the amplifier is good for a 5 dB improvement in S/N; with a 6 dB cable loss the improvement can be as high as 10 dB. Even though these figures cannot be completely realised in practi- cal applications, due to unavoidable mis- matching etc., layout 3 shows a clear superiority over layout 4 and is decid- edly close to the ideal of layout 2. Overload problems Overloading of the amplifier or receiver is a possible adverse result of aerial amplification. Most modern types of amplifier are reasonably free from this effect, so the first one to suffer would be the receiver itself. Severe over- loading may even result in complete squelching, especially if the amplifier is of the untuned type and not fitted with automatic gain control. Overload conditions manifest themselves by the production of harmonics, un- wanted demodulation and intermodu- lation. These spurious signals can result in multispot tuning to the same trans- mission, swamping weaker transmissions, images and beat frequencies. Strong and weak signals on neighbouring wave- lengths could be demodulated together, especially in receivers with deficient AM suppression. Other harmful phenomena are chirping 'birdies' in FM stereo demodulation as well as chatter and whistles in AM reception. When one or more of these troubles manifest themselves, the best advice would be to substitute tuned amplifi- cation or else discard it altogether and install an aerial with directional charac- teristics and high gain. Another solution might be to insert a tuned pre-ampli- fying stage with automatic gain control or invest in a superior receiving system. The best HF stage is a good aerial This adage is given new life by the 'happy circumstance that aerials for the very and ultra high frequency ranges can be designed to give considerable 'gain'. This 'something for nothing' can be achieved from the directional character- by which an aerial array can be made to concentrate the field energy of a transmission and so permit a much higher pick-up efficiency. The 'passive' gain so realised isexpressed as the aerial output level for a given field strength in relation to the output 1 1 dB gain; 6d.an UHF array with as many as 91 (!) of a simple dipole. It is usually ex- pressed in dB; an aerial of 8 dB gain picks up 6.3 times the energy of a dipole. This gain in turn results in a clean 8 dB improvement in the signal- to-noise ratio, which verifies the truth of the statement heading this chapter. An aerial, no matter how high its gain, cannot be overcontrolled - and it needs no power supply. In spite of these arguments there may be instances where it is absolutely necessary to use an aerial signal ampli- fier, due to circumstances beyond the listener's or viewer's control. In these cases the design for an efficient one may prove welcome and so, true to form, Elektor have come up with the goods. Such a design is described elsewhere in this issue. N p.c. board for the It was probably the simplicity of the circuit's design which helped it to score such high points. It extends the uses of an oscilloscope considerably and at a relatively low cost. Usually one can only observe several TTL levels at once by taking notes at the same time. If the levels are static there is no problem, but when levels vary it becomes more difficult. The digisplay can be used to observe static levels as well as slowly changing ones (for signals which change rapidly, other methods will have to be sought), so that an overall picture of everything happening in the circuit is obtained at a glance. digisplay The lay-out The original circuit (as published in the Summer Circuits issue) has been modified slightly, as shown in figure 1. The oscillator around N1 and N2, in particular, needed some attention. The result is an oscillator which starts up without any trouble at all — unlike the older version. The complete circuit works as follows. The logic levels which are to be displayed on the oscilloscope screen as 'noughts' and 'ones', are fed to the IC1 inputs and are passed in sequence to the output (pin 10) in inverted form. Which input signal reaches the output depends on the information at the inputs A, B, C and D. This is provided by a hexadecimal counter (IC2), driven by the oscillator (N1 and N2). As the oscillator always runs (as long as voltage is applied!), the binary information will keep changing from 0 to 15 and then start all over again. The input signals to IC1 are therefore scanned, appearing in sequence at pin 10. The circuit around T1 is a sinewave generator. The sinewave produced passes through R7 to the Y output. C4, C5, R5 and R6 constitute a phase-shifting network. If the W output of IC1 is low, this network is blocked. If, on the other hand, pin 10 is high, the phase-shifted sinewave reaches the X output through R9. The binary information given by IC2 is also used to determine the position of the 'ones' and 'noughts' on One of the winning circuits of the Eurotronics competition was circuit no. 68 in the 1979 Summer Circuits issue: the digisplay. We promised p.c. boards for winning circuits . . . elektor ■ — R1.R2.R3> 3k9 R4 = 820 k R5.R7- 100 k R6.R11 = 10 k R8 - 47 k R9 “ 120 k RIO - 330k R1 2.R13 - 3k3 R14.R15.R16.R17 = 6k8 R18 = 680 f! the oscilloscope screen. N3 . . . N5 super- impose various DC voltages on the X output. In this way, each input signal is given its own position on the screen — at least, as far as the first eight input signals are concerned. There are, how- ever, sixteen signals altogether, so that in order to get a clear overall reading two different DC voltages must be applied to the Y output. This has been done by using the D output of IC2. Two rows of eight signals are displayed on the screen, corresponding to the sixteen pins of a DIL-IC. If the output of IC1 is low, a sinewave appears at the Y output and a DC voltage at the X output. The spot on the oscilloscope's screen is therefore horizontally fixed, whereas vertically it moves up and down with the sinewave. The result is a short, vertical line on the screen. When the output of IC1 is high, the phase-shifting network is no longer blocked, so that a sinewave is fed to the X output as well as the Y output. Since these sinewaves are shifted in phase with respect to each other, a lissajous figure in the shape of an ellipse appears on the screen. The ellipse's position is also determined by the DC voltage at the Y and X outputs. PI is one component which was not part of the original circuit. It has been added as an adjustment point for the horizontal position of the ones. If the IC1 output is low, the output of N6 will be high. This is the case when the input of IC1 being scanned at that moment is also high. This voltage is applied to a voltage divider consisting of RIO, PI, R1 1 and one or more of the other ’ resistors, depending on which of the gates between N3 and N5 is (are) low at that particular moment. Using PI, the DC level at the X output can be slightly altered. When the output of N6 is low (caused by a '0' at the 'active' input of IC1 ) , RIO merely constitutes a minor load for the DC bias at the X output; PI now has very little effect on this voltage. The construction A suitable p.c. board is shown in figure 2. With its low current consumption (approximately 20 mA), the digisplay can often be fed from the circuit being . tested. If required, a separate supply can be added as shown in figure 3. There is no room for it on the p.c. board, but it shouldn't be too difficult to construct on a piece of Veroboard or similar. Finally, it should be mentioned that only TTL levels can be displayed on the screen with the aid of the digisplay and that an oscilloscope which has a connection for an external time base (X input) must be used. Unconnected inputs cause ‘ones’ to appear on the screen. The best way to make the > necessary connections between the signals to be measured and the digisplay _ is with the aid of a TTL test clip. K Bucket brigade delay lines are not unknown to Elektor readers. We've used them in an analogue reverb unit and for a TV scope. At that, we've really only scratched the surface of what promises to be a new area in electronics: analogue delay line technology. New types of delay line are appearing at regular intervals, and new applications are even more commonplace. Not only reverb, phasing and similar 'musical' applications: filtering, scrambling and real-time spectrum analysis are also possible. There are two main types of electronic components that can be used to delay an analogue signal. Although their oper- ating principles are completely different, they can be used in very similar appli- cations. The first group are the so-called charge transfer devices: CTD for short. Bucket brigade delay lines belong in this category, as do charge coupled devices (CCDs). Both of these types of CTD can be used in virtually identical appli- cations. Charge transfer, using a CTD, is not the only way to delay an analogue signal. Another way is to convert the electrical signal into mechanical vibrations. These vibrations can cause mechanical 'waves' aiuilognc delay technology in a solid, for instance; at some distant point you pick up these waves again and convert them back into an electrical signal. With a little care — making sure that the mechanical wave can only travel along one path, for instance, so that the path length is constant - it is possible to make the electrical output signal identical to the original input. Delayed, of course, but that is the whole object of the exercise. This principle is used in surface acoustic wave devices, or SAWs. A 'SAW filter', for instance, is fairly well-known. Little capacitors, all in a row If you want to delay an analogue signal, have to store it somewhere for a One solution is to 'sample' the signal at regular intervals and store the samples. This is what happens in a CTD. One of the simplest CTD arrangements is shown in figure 1. Basically, it is than a chain of little One 'plate' of the capacitor is a gate electrode; the other is the corresponding part of the semi-con- ducting P-silicon layer. The dielectric for the capacitor is silicon oxide. Each group of three capacitors (gl . . . g3, for instance) form one step in the delay line. The analogue samples of the signal are moved down the chain as charge packets: one packet for each sample. Starting with the situation where the first charge packet is 'under- neath' the first gl, the procedure is as follows. The voltage on g2 is made more positive, and that on gl more negative. This 'pushes' the (negative) charge from gl to g2. Then g3 is made more positive and g2 more negative (but not as nega- tive as gl I), squeezing the charge packet on to g3. Finally, it is moved in the same way to gl in the next set of three; simultaneously, the next charge packet 1980 - 2-35 — corresponding to the next sample — moves underneatch the first gl. In all, 1 three 'transfer pulses' are needed to move the charge packet up one step in the delay line. The rate at which the charge packets are moved along the chain depends on the frequency of these transfer pulses; this, in turn, determines the total delay time. The first step in the CTD is a normal PN junction. The analogue input signal is applied (with a positive bias voltage) to the N-silicon embedded in the sub- strate. This pulls a negative charge (electrons) to the P-side of the junction; the higher the input voltage, the greater the charge. A short pulse on the J*?' 'sampling electrode', g s , pulls this charge exam 'underneath' g s — ready to start down the line. Note that another way of looking at this is to consider the whole 2 input circuit (input electrode, sampling electrode, and first 'gate', gl) as a MOS transistor. The three electrodes can be considered as source, gate and drain of this device, respectively. The last step in the CTD is simplicity itself: the output signal is taken off from the last electrode. Since this is a capacitive source - and a very small capacitance, at that - a very high impedance buffer amplifier must be It should be noted that three capacitors per step is not essential. It can be done with two (if you're very skillful!) or with more than three. In practice, three capacitors are often used, however, since this is the minimum required to keep the charge packets from 'running into each other' when using straight- forward technology. There are other ways of making a CTD, Fi9U , as mentioned earlier. However, the basic The t principle — moving charge packets along 'cieai some chain — is always the same. __ Blay technology Two advantages Charge transfer devices have two signifi- cant advantages: the total delay can be varied by altering the frequency of the transfer pulses - offering easy external control. Furthermore, a CTD is fairly simple to make - for an 1C manufac- turer, that is! The manufacturing process is the same as that used for normal integrated circuits. For this reason, it is also an attractive idea to combine a CTD with some other semiconductor device, on the same chip. The buffer amplifier in figure 1, for instance, and the 'clock' generator that provides the transfer pulses. It is also possible to incorporate a charge transfer device in a circuit for one particular application, and integrate the whole lot on the same chip. A hundred-step CTD (using 300 capacitors) can be squeezed onto an area of only 2.5 x 0.25 mm. This is only 2.5% of the total area on a 5 x 5 mm LSI chip! So what do you do with them? The sampling rate determines the highest frequency that can be delayed by a CTD. Devices are commercially available that can be used at sampling rates of up to 20 MHz, which means that signals up to 10 MHz can be handled. In the lab stage, devices exist that work at a sampling rate of 1 30 MHz (signals up to a good 60 MHz). Speed is one thing, length is another: CTDswith more than 1000 steps in the chain are quite common already! 'Normal' CTDs, as described so far (special types will be dealt with later), have been used in Elektor more than once. For sound effects, in particular: phasing, flanging, vibrato, chorus, reverb and even echo; all of these effects, and more, can be obtained with CTDs. There is little to be gained by going into all this again; the list at the end of this article refers to all previous articles. Another application is for measuring instruments. For instance, CTDs can be used for timebase expansion or com- pression. This possibility is utilised in the extended version of the Videoscope. 3 Figure 3. Basically, a Surface Acoustic Wave (SAW) device looks something like this. The conducting electrode fingers' are deposited on a piezo-electric substrate; this transmits the signal as a mechanical ‘wave' from input to output transducer. 4 transducer A signal is first 'stored' in a CTD using one sampling frequency, and then 'played back' using a different fre- quency. The result is that the output signal is either a 'stretched' or a 'com- pressed' version of the input signal. Another 'measuring' application of CTDs is so-called 'transient recording'. Transients are defined, quite succinctly, in the Oxford dictionary, as 'not perma- nent' and 'of short duration'. Very true. This type of signal - interference pulses, say - is not easy to examine on an oscilloscope: it’s gone before you realise that it was there. To view a transient on a 'scope, you have to store it. In a CTD, spacing of the 'fingers' in the output >rt signel 'bursts'. for instance; that way, it can be 'played back' once for each sweep and at a different speed, if necessary. One important application for this is medical electronics: irregular heart-beats, brain- waves, and so on. The application shown in figure 2 is intended for video recorders - and audio recorders too, for that matter. The idea is to eliminate the effect of rapid variations in the tape speed (flutter). For video recorders, especially, even the least trace of flutter is notice- I 4. This type of SAW device, with ever-decreasing ran be used for spectrum analysis of she ZnO Si-MOSFET SURFACE WAVE TRANSVERSAL FILTER DIGITAL SHIFT REGISTER BUCKET BRIGADE DEVICE -jaaa^ssSiiSuBassssssasssssassss!! mil } SI-MOSFET DETECTORS lalogue delay technology 1 980 - 2-37 able in the reproduced signal. During recording, a reference or 'pilot' tone is also recorded on the tape. At the playback side, this pilot tone is filtered out and compared with a stable refer- ence, using a phase detector. The output from the phase detector is used to vary the output frequency of a clock gener- ator that provides the 'transfer pulses' for a CTD; the whole signal, pilot tone and all, passes through this CTD. Pro- vided the delay time is long enough, the loop can be designed to maintain a constant pilot tone frequency at the output of the CTD. This, in turn, means that any 'flutter' in the main signal is eliminated. Surface acoustic waves A surface acoustic wave device works on an entirely different principle than a CTD. The reason for discussing both in the same article is that they are both suitable for a very similar (and very extensive!) range of novel applications. The basic construction of a SAW device 5a is even simpler than that of a CTD — see figure 3. Its operation is based on the piezo-electric effect. Piezo-electric materials alter their shape when a volt- age is applied across them, and vice versa: when they are 'mechanically deformed' a voltage appears across the material. A sudden, sharp blow on a piece of piezo-electric material can pro- duce a brief pulse of several thousand volts. More than adequate for a very nice spark, as can be seen in one type of 'electronic' lighter. A somewhat less obvious application of the same effect is in 'crystal' microphones and some 'tweeters'. A SAW device consists of a slice of piezo-electric material, with conducting electrodes on its surface. At one end, the electrodes are used for an 'input transducer' that converts an electrical input signal into mechanical vibrations; at the other end, a similar set of elec- trodes converts the mechanical vibrations back into an electrical signal. The mechanical vibrations travel as a kind of shock wave, mainly along the surface of the material; the amplitude of this 'VI 'surface acoustic wave' is very small, in the order of a few nanometres (ID* 9 m ) . The SAW filter If the piezo-electric material is suf- ficiently pure and regular of structure, the speed with which the mechanical wave travels across the surface is vir- tually constant over a wide (input) frequency range. It is in the order of 3000 metres per second, or one hundred- thousandth of the speed of an electro- magnetic wave in vacuum. The means that the wavelength is also shorter, in the same proportion. For example, the wavelength of a 30 MHz signal in air is 10 metres; in a SAW device, the corre- sponding wavelength is only 0.1 mm. This fact can be used to construct a SAW filter: a selective device. If the electrodes, for both the input and the output transducer, are spaced at 0.1 mm intervals, signals with this wavelength will be boosted - whereas signals at a different wavelength will tend to cancel. As more and more electrodes are used in parallel (so-called 'fingers'), for both in- and output transducers, the filter becomes more and more selective. In practice, SAW filters are manufactured that are almost incredibly 'sharp'. At present, these devices are used for signal frequencies from 5 MHz up to a few GHz. They are already in use as selective filters in the IF strip of some television receivers. The advantage is that the assembly is simplified, since no 'alignment' is needed; the disadvantages (unimportant in TV sets) are high damping and the fact that the 'reson- ance frequency' is completely fixed in the manufacturing process - there is no way to alter it afterwards. Other possibilities Using surface acoustic wave devices as highly selective bandpass filters is one possibility. But there are others. In the SAW device shown in figure 4, for instance, a different electrode layout is used. The input transducer (at the left) consists of only two 'fingers’, so that it is relatively broad-band - not at all selective. The output transducer consists of several 'fingers' at decreasing distances. Initially, the fingers are widely spaced, making that section of the transducer especially sensitive to relatively low-frequency signal com- ponents; as the spacing decreases, the transducer becomes more sensitive to higher frequencies. Now, let us assume that a short signal 2-38 - 1 technology analogue delay 'burst' is applied to the input. The input transducer converts it into a 'wave' that travels across the device. After a very short delay, signal components start to appear at the output: first the low- frequency components and then, as the wave passes under more closely spaced fingers, the signal components at higher frequencies. The total input signal is split up, in other words, with its various frequency components appearing con- secutively at the output. As sketched in figure 4, the output signal from this device can be rectified and displayed on an oscilloscope. This forms the basis for a high-frequency spectrum analyser! If the scope is triggered at the same moment that the signal burst is applied to the SAW device, the amplitude of the lower frequency components will be displayed first — followed by the amplitude of ever higher frequency components, as these appear at the output of the SAW. Obviously, this system can only 'analyse' the signal one burst at a time. Even so, it can be the basis of a one-chip spectrum analyser. This would require quite a bit of additional electronic circuitry on the same chip as the SAW device, of course; in this connection, it is interesting to note that monolithic devices have already been manufactured that include both a SAW device and 'normal' semiconductors on the same chip. This is not as impossible as it may seem at first sight. On part of a silicon substrate, an integrated circuit can be manufactured in the normal way; on another part of the same substrate, a layer of piezo-electric material (zinc oxide, for instance) is deposited, as the basis for the SAW device. The electrode 'fingers' can be added at the same time as the conducting strips for the rest of the integrated circuit. 6a Figure 6. A lass regular 'chirp' filter pair. As before, one filter can be used to 'recognise' the signal transmitted by the other. Figure 7. The basic principle of a ’transversal' filter. Virtually any filter characteristic can be selected, by choosing the correct combination of delay time (rl and weighting factors (w). Chirp! The same SAW device used in figure 4 is drawn again, twice, in figure 5. This is not just an easy way to fill magazine pages (!): we are interested in another useful application. A brief, 'spikey' pulse is applied to the input of the SAW device in figure 5a. This type of pulse contains a whole range of frequency components; as the corresponding wave travels down the SAW device, a so-called 'chirp' signal appears at the output: a sinewave at rapidly increasing frequency. If this signal is applied to the input of the second SAW device, as shown in figure 5b, the inverse characteristic of this device (high frequencies first, low frequencies last) pulls the various components together again - re-creating the original 'spike'. Chirp signals of this type are used for radar. There, the whole idea is to transmit a short pulse and listen for the echo — in other words, listen for your pulse coming back after bouncing off some object. The problem is to know whether it really is your pulse that you're hearing - there are plenty of others around! However, if the pulse is converted into a 'chirp' before trans- mission and the received signal is converted back, your very own chirp is the only signal that will produce a nice sharp pulse at the output. An additional bonus is that the transmitter doesn't have to put all its power into one short pulse. To put it another way: with a given peak output power rating, a radar transmitter can put a lot more energy into a chirp signal than it could into a Matched filters The chirp filters described above are one example of so-called matched filters. One filter converts a spike input into a particular output signal; the other will only produce a spike output when that particular signal is applied to its input. In other words, the second filter is 'selective' for that particular signal (note that has nothing to do with 'normal' filter selectivity, for one particular frequency!). An almost unlimited number of vari- ations are possible, on the same theme. The transmission filter can be designed to create almost any output 'tune', and the corresponding second filter will reconvert this into a spike. A further example is given in figure 6. It should also be noted that the two filters in a matched pair can be interchanged: instead of transmitting, the filter shown in figure 6a can be used to 'decode' a signal transmitted by that shown in figure 6b. The only difference is that the 'tune' will now be played 'backwards'. elektor