January 1980

up-to-date electronics for lab and leisure

colour genera

for a lovely light show

constant speed und<

1980 - UK 03

january 1980
volume 6
number 1

$

n=

-TL

page 1 -06

The colour generator
produces a fairly
constant amount of light,
shifting slowly through
the spectrum. When used
for indirect lighting, it
produces a beautiful
effect.

selektor

the eighties

COlOUr generator (VI/.H.M. van Dreumel)
digital tuning scale

mini drill speed control

Miniature electric drills have been available for some time.
Most of them are battery powered. For precision work, it is

maintained, independent of the load, so much the better.
Both of these objectives can be achieved fairly simply, using
an integrated voltage regulator.

VSWR meter

To get the maximum power output from a transmitting
aerial, it must be properly matched to the transmitter.
Amateur radio operators often make use of a so-called
voltage standing wave ratio meter.

frequency doubler

This frequency doubler circuit for guitars produces an output
that is one octave higher than the original input signal.

elektor vocoder (1)

talk funny on board

1-01

1-04

1-06

1-16

1-18

1-22

1-24

1-30

page 1 -08

The electronics inside radio receivers have changed
radically over the years, but tuning scales remain
basically the same. The 'bent wire on a string'
principle is still the most popular. In this article, a
digital tuning scale is described.

page 1 -24

Here it is at last! A 10-channel vocoder. The design
offers good performance at a very reasonable cost.
Ideal for musicians with a lot of enthusiasm but
insufficient funds to back it up!

new bus board for microprocessors 1-31

modern methods of voltage regulation 1-32

In this article we take a look at the advantages and dis-
advantages of switching regulators from a practical point of

ejektor: PWM applications 1-36

market 1 -39

advertisers index UK-30

Now, the complete MK 14 micro-computer
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PROM PROGRAMMER.
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□ Cassette interface modi

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Specification

The Acorn consists of two

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1. MPU card

6502 microprocessor
512 x 8 ACORN monitor
1 K x 8 RAM

16-way I/O with 128 bytes
of RAM
1 MHz crystal
5 V regulator, sockets for
2K EPROM and second
RAM I/O chip.

2. Keyboard card

25 click-keys (16 hex, 9
control)

8 digit, 7 segment display
CUTS standard crystal
controlled tape interface
circuitry.

Keyboard instructions:
Memory Inspect/Change
(remembers last address
used)

Stepping up through
memory

Stepping down through
memory

Acorn Microcomputer

Price £65 plus VAT in kit form

This compact stand-alone microcomputer is based
on standard Eurocard modules, and employs the
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been to provide full expandability, versatility and
economy.

Set or clear break point
Restore from break
Load from tape
Store on tape
Go (recalls last address
used)

Reset

Monitor features
System program
Set of sub-routines for use
in programming
Powerful de-bugging facility
displays all internal registers
Tape load and store
routines

Applications

As a self teaching tool for
beginners to computing.

As a low cost 6502 devel-
opment system for industry.
As a basis for a powerful
microcomputer in its ex-
panded form.

As a control system for elec-
tronics engineers.

As a data acquisition system
for laboratories.

START WITH SYSTEM 1 AND CONTINUE AS AND WHEN YOU LIKE

Acorn VDU

£88 plus VAT (kit form)

Acorn Controller

£35 plus VAT (min config.)

Acorn Memory 8 k
£95 plus VAT (kit form)

Acorn Computer! Ltd.

Cambridge (02231 312772.

l""order Form

| Please send me the following:

j □ (qty) Acorn Microcomputer kit @ £65 plus £9.75 VAT.

a (qty) Acorn Memory kit @£95 plus £14.25 VAT.

' ZZ (qty) Acorn VDU kit @£88 plus £13.20 VAT.

I Hj (qty) Acorn Power Supply (for System 1 only) @ £5.95 plus
£0.89 VAT.

~Z (qty) Acorn Microcomputer assembled and tested @ £79
plus £11.85 VAT.

i (qty) Acorn VDU assembled and

I tested @£98 plus £14.70 VAT. Acorn Computers L

the CPU card of System 1, it allows for up to 4V& k EPROM,
1% k RAM and 32 I/O lines. It has on board 5 V regulator
and optional crystal control. Custom programs may be
developed on System 1 and the card makes an ideal
dedicated hardware module.

A fully buffered memory card allowing up to 8 k RAM
plus 8 k EPROM on one eurocard, in an Acorn system
both BASIC and DOS may be contained in this module.
Static RAM (2114) is used and the card may be wired into
other systems.

A memory mapped seven colour VDU interface with
adjustable screen format. Full upper and lower ascii and
teletext graphics are features of this module which along
with programmable cursor, light pen, hardware scroll etc.,
make this the most advanced interface in its class.

Acorn BASIC - a very fast integer BASIC in 4 k
Acorn COS - a sophisticated cassette operating system
with load and save and keyboard and
VDU routines in 2 k

Acorn DOS - a comprehensive disc operating system in
4k

Post and packing free on all orders.

1 enclose a cheque for £

(indicate total amount) made out to Acorn Computers Ltd.
Please send me further details of this and other Acorn options

WlCORN
COMPUTER

a. Cambs. (0223) 312772. Regd. No. 1403810

1980 - 1-01

Cries of stress

Interpreting the tiny noises that ma-
terials and structures emit when they
are loaded is the fastest -growing tech-
nique in non-destructive testing. It is
already being applied to find developing
cracks in mechanical structures, leaks in
nuclear reactor systems and. at the
other end of the scale, to detect loose
bits of solder or other debris that might
short-circuit electronic equipment.

It is common knowledge that when we
bend a piece of wood to its breaking
point, it makes a noise before it frac-
tures. It has even been said that early
miners were soon sorted out into the
'quick and the dead' by how well they
could assess the creak ings in the rock
above their heads. Audible warnings of
movements in the ground used to come,
too, from wooden pit-props, especially
if they were cut from larch. And many
eye-witness accounts of disasters to do
with mechanical structures such as
bridges, boilers and pressure vessels refer
to a loud creak or crack being heard
some time before the breakdown
eventually occured.

So it is hardly to our credit that en-
gineers have only recently come to
exploit the phenomenon of what we
now call acoustic emission, to give us
valuable information about materials
under load. This is particularly true
when we reflect that Robert Hooke, the
great English experimental physicist,
drew attention as long ago as 1681 to
"the possibility of discovering the
internal motions and actions of
bodies . . . whether animal, vegetable
or mineral, by the sound that they
make'. Two centuries later, the medical
profession had adopted the stethoscope
to help discover the health of patients,
and by the turn of this century the first
experiments were under way at
Cambridge to detect the electrical
signals from organs within the human
body. Seismology has been well
established for many years, making it all
the more remarkable that it was only in
1924 that Ehrenfest and Joffe, in Russia,
first reported hearing 'ticking' noises
from a crystal of rock sheared at 450 C.
This was followed by a report in 1927
from Claussen-Nekludova in Germany
of a correlation between tiny jumps in
the lengths of stretched specimens of
brass and aluminium, held at tempera-
tures above 400° C, and weak 'crackling'
noises the speciments gave off. But it
was not until further work had been
done by Kaiser, in Munich in 1950, that
anyone began seriously to consider that
we could learn something useful by
listening to the sounds coming from
metal or wood under stress.

— Component under test

P-

'^inoMfie

4

Figure 1. Simple j
added, as shown.

Today we know that most materials,
including wood, metal, plastics and
rock, give out such noises, and that by
listening to them we can find out three
things. First, we can detect that some-
thing, an 'event', has happened in the
material. Second, by measuring the
difference between the sound arriving at
two or more sensors, and knowing the
speed of propagation of the sound in
the material, we can calculate where the
event took place. Third, by studying the
signal we can sometimes infer the nature
of what has happened. We may, of
course, need to amplify the sounds and
listen beyond the audible frequency
range to hear them. But even more, we
have discovered recently that not only
mechanical and thermal loading produces
noise, but electrical stressing too:
insulators and dielectric materials give
early, audible warning of electrical
breakdown.

Need for Research

Few engineering structures fail during
use; but the risk is always there, so we
must guard against it. The more serious
the consequences of failure, the more
searching the precautionary measures
must be and the more often they have
to be taken.

When a structure such as a pressure
vessel fails, it does so either with a hiss
or a bang, and in the latter instance
people may be killed and extensive
damage may be done to property, so the
most important aim in design, manufac-

2

ture and inspection is to ensure that the
risk of either sort of failure, and es-
pecially of an explosion, is as small as
possible. The various compulsory
measures, codes of practice, design
procedures and inspection routines are
now very good, as is shown by the
extremely low incidence of failure; but
demands on integrity are becoming
more stringent, through the use of
higher pressures and the need to store
more dangerous materials than hitherto.
As the vessel ages, the risk becomes
greater that there is some insidious
deterioration going on somewhere in the
structure.

It is, of course, reasonable to suppose
that experience guides us to the part of
the vessel where deterioration is most
likely to occur, and that appropriate
non-destructive inspection can be
concentrated there at regular intervals in
the hope of detecting cracks before they
grow too big. But sometimes the
available inspection techniques do not
have the resolution or reliability needed
to find the sort of flaws that have to be
detected if they are not to become
dangerous.

One of the most serious limitations of
established non-destructive test pro-
cedures, which include ultrasonics,
radiography, and magnetic and eddy-
current techniques, is that it is not
always easy to apply them precisely at
the place to be examined. This is where
the so-called acoustic-emission technique
has much to offer, because it enables us
to listen to the whole of the volume of
the material in the structure from only
one or two places, which are likely to be
easily accessible. Nevertheless, confi-
dence in an ability to hear defects
growing, without missing any, depends
upon first proving that the kinds of
defect we are looking for always emit
detectable noises under specified
conditions. Research is now going on in
many countries to define what can and,
rather more important, what cannot be
heard with this new technique.

Detecting and Processing
A sensor mounted on the surface of a
component can easily detect the stress
waves beneath it. The most common
sensors are pieces of piezoelectric (PZT)
ceramic which, when vibrated, produce
small voltages. Detecting, amplifying
and processing systems vary from
something as simple as that shown in
figure 1 (no more complicated than a
gramophone pick-up, amplifier and
loudspeaker) to multi-channel,
computer-on-line defect-tracing systems
such as that outlined in figure 2, which

Figure 3. The

single stress wave.

automatically print information about
the place and extent of an event, within
a second or two of its happening, on a
diagram of the structure under test.

An idealized picture of the signal from a
PZT transducer excited by a single stress
wave is shown in figure 3. To analyse
such signals it is usual to count the
number of times the signal crosses the
threshold level, which is set above the
noise level, to give what is known as the
ring-down count. It is also common to
measure the amplitude of the signal or
the repetition rate at which such signals
are received, or to analyse the fre-
quencies contained in the complex
wave. Various combinations of measure-
ments may be made, depending on the
structure under study.

So far, the widest application of
acoustic-emission techniques, now
known as AE, is in non-destructive
testing. NDT is carried out on all bridges,

boilers, boats, aircraft and nuclear
reactors, to name only a few structures,
to find out whether they are safe to use.
AE has one enormous advantage over all
other NDT techniques in that it is able
to monitor an entire structure when,
perhaps, only a few points may be
accessible. This makes it particularly
attractive for monitoring large process
or pressure vessels in dangerous environ-
ments, allowing a watch to be kept from
a safe place.

Theoretically, any abrupt relaxation
anywhere within a material gives out a
pulse of stress waves which can be
detected and used to find and study the
source. In practice, not all such events
can be detected. A simple way of
looking at this is to think in terms of a
partition process in which any event,
which may be the growth of a crack, a
change of state (phase transformation),
local yielding, electrical breakdown and

Figure 4. Any 'event', or relaxation in a material re
one of three forms, through a partition process.

rgy which we may regard as taking

so on, releases energy which in the end
takes one of three forms. This is shown
diagrammatically in figure 4, where it is
seen to end up as energy stored in the
atomic lattice of the material, as a new
crack surface or as heat and sound. A
transitional form of the last of these is
stress waves and, provided they are
detected, they may enable us to infer
something about the other two, but one
of the fallacies in the early hopes for AE
was always to assume that detected
energy meant some sort of crack growth.
Though that is often true, there are
many instances of noise being made
without a crack occurring (for example,
the well-known 'cry of tin' is plainly
audible when we bend a stick of tin; it is
caused by bending of the crystals in the
metal). If follows that it would be
wrong to assume a unique relation-
ship between the number or amplitude
of the stress waves counted and the area
of crack produced, without knowing a
great deal more about the circumstances
under which the crack is growing and
unless calibration experiments have
been done; such experiments are by no
means easy.

An even more important point is to
note the difference between AE and all
other forms of NDT. In any other NDT
technique it is possible to go back again
and again to check the indications of a
flaw in the material, but noise is
generated only by movement of the
flaw, that is, while the event is taking
place. This means that the material or

firs! prize winner

As our regular readers will know, the
articles published in the July/August
summer circuits edition this year were
the entries for the Elektor competition
in which you, our readers, voted the
winners. Against stiff opposition (with
competitors from 14 countries) the first
prize went to Mr. John Mitchell of
Balham, London. Mr. Mitchell, pictured
here with his prize, was a clear winner
with a total of 58,158 points for circuit
number 106, the Chorosynth. His choice
of goodies was a Roland MP 700 electric
oiano, Roland RS 202 string synthesiser
and a Roland JC 120 amplifier. Readers
■** . * be pleased to know that a printed
circuit board for the Chorosynth is
nearing completion and will be published
together with a full length article in two
or three months time.

A final word on the competition: out of
the 20 prizes, the U.K. won six - with
three of them inside the first four places!

structure has to be suitably stressed
mechanically, thermally or electrically
to stimulate the sort of event we are
trying to detect, and the AE monitoring
equipment must be switched on and
thoroughly reliable at the time the
sound is emitted.

Potential

Metals, composite materials, ceramics,
ferro-electric materials and dielectrics all
make a noise when stressed. Studies
have shown how we can use the noises
to gain information about the quality of
the material, the growth of cracks and
other defects within structures, and
whether the cracking is caused by
tensile loading, fatigue, stress corrosion
or some other factor; the technique also
enables the sounds associated with
certain kinds of phase transformation
within the solid material to be detected.
So AE can be used as a tool for materials
research. It has already proved able to
detect changes as diverse as deterioration
in the condition of concrete, a growing
crack in an aircraft wing, the formation
of ice crystals in ice-cream and unre-
liability of thermistors in electronic
systems. Use of AE for process control
in welding and to keep a lookout for
loose parts in equipment being assembled
is already established and it is being
used in mines to monitor micro-seismic
activity around places where the rock
formation is hazardous.

SPECTRUM No. 165

(512 S)

Computing at the speed of light

A 'superfast' computer working at the
speed of light to make calculations
1000 times faster than even the speediest
of today's electronic brains could be the
result of a radical break-through in
thinking machine technology made by a
team of scientists at Heriot-Watt
University in Scotland's capital city,
Edinburgh.

Harnessing space-age laser technology,
the Heriot-Watt team have produced a
unique and revolutionary device called a
'transphasor', which they claim, is the
key to a totally new generation of
optical computers whose switches are
operated not by electronic components
but by high intensity light beams.
Research has shown that an optical
computer using laser controlled switches
instead of conventional electronic
devices is literally just around the corner.
This would represent a breakthrough of
the same magnitude as the transistor,
the component on which the post-war
electronics revolution was based.

The technique developed by the Heriot-
Watt team involves the direction of a
laser beam onto a flat crystal of a
semiconductor made of the elements
antimony and indium. An adjustment of
the laser's power causes the crystal to
start resonating, in turn suddenly
boosting the brightness of the laser
beam being transmitted through it. That
sudden jump in output is the basis for
an incredibly fast optical switch.

It is a remarkable development that it is
now possible to switch, control and
even delay the progress of something as
rapid as a beam of light within a
timescale that can only be measured in
picoseconds - that is an incredible one
millionth of a millionth of a second.

(513 S)

the riglilk^

Goodbye, seventies!

January 1980: the beginning of a new
decennium. It's always interesting to try
to guess what will happen in the next
few years. Now, especially - so many
predictions have already been made for
this period, ranging from rosy optimism
to the most gloomy pessimism. Every-
body has heard of George Orwell's book
'1984' -even if they haven't actually
read it. We don't expect to have a
'Big Brother' within the next ten years,
but what about all the other pessimistic
prophecies? Are we to be killed by
radiation when the ozon layer in the
upper atmosphere disappears, or are we
to freeze to death when our energy
supplies run out just in time for the
next Ice Age, or will we be roasted alive
due to the 'greenhouse effect' of all that
CO2 we're putting out?

It may seem strange, but we just don't
feel like being pessimistic. Technology is
developing. The direction it takes is for
us to decide; if we're even half as
intelligent as we think, we should be
able to solve our problems in time.

Let's make a few guesses about what
may well become technically feasible
within the next ten years. First off: easy
access to all kinds of information, for
everyone who wants it. Teletext and
Viewdata are still in their infancy, but
we can expect that these services will
expand rapidly. At the same time, a lot
of the 'learning by rote' is disappearing
from education — this type of 'ballast'
can just as well be stored in computer
memories. If we are to really utilise this
type of 'data bank', however, it is
important that we learn to interpret
facts and judge results correctly. This
need will be reflected in education:
there will be more emphasis on teaching
how to find the information you need,
and how to use it once you've found it.
At the same time, it is to be hoped that
higher education will become more
truly 'universal' (why else call those
institutes 'Universities'?) and less
specialist-oriented. One of our main
problems at present is that specialists
from different fields often cannot
communicate on anything above the
lowest level.

We seem to be getting off our subject:
what will become technically feasible
within the next few years. Even if we
restrict ourselves to technical prophecies,
however, we've got a problem. It is to
be expected that a completely new
technology will appear within the next
few years — with far-reaching conse-
quences. However, without knowing
what that technology is, it's difficult to
guess at the results!

This is no wild assumption. Think back
over the last century. It started with
'passive' electronics - well, just
'electricity', really — but the telephone,
telegraph and electric light were in use.
Then, in 1907, the first 'valve': an active
component! With it, radio, radar and
television became possible, to give a few
examples. Even the first computer, in
the mid-forties. But things were
becoming big and bulky . . . until the
transistor appeared, in 1947: the first of
the semiconductors. Using this new
technology, we are now making things
that would be virtually impossible with
valves: portable radios, pocket calcu-
lators, digital wrist watches, heart
pacemakers, powerful computers . . .
From thermionic valve to semiconductor
took forty years. The semiconductor
has been with us for over thirty years
now, and everyone maintains that
progress is becoming ever more rapid.
Isn't it time for another radically new
technology? With the same impact as
valves and semiconductors, in their day?
It would seem likely. But what is it to
be? Molecular electronics? Optical elec-
tronics — or 'optronics', to coin a

Whatever it is, it will almost certainly be
used to make life easier. Further
developments of semiconductor tech-
nology are also moving in that direction.
This means that it is possible to make
some educated guesses about the
future — new technology or no. Here we
go:

Goodbye record! Goodbye tape!

For sound recording, gramophone
records and tapes are nearing the end of
their useful life. Even the brand new,
almost revolutionary Philips 'Compact
Disc' is not likely to last. Admittedly,
sound recording will go digital. As far as
that goes, the Compact Disc is a step in
the right direction. But all that fiddling
with moving, mechanical parts! No,
vinyl disc and tape will follow the wax
cylinder - into the museum.

In a digital audio system (say 16 bits
and a sampling rate of about 50 kHz),
one hour in stereo corresponds to some
6000 Mbits. Six thousand million bitsl
Where can you store them? At present,
the answer is: on a Compact Disc. But
integrated circuit memories also exist.
There's even a 'bubble memory' with a
storage capacity of 1 Mbyte, but it's still
rather expensive . . .

Currently available ROMs haven't
enough storage capacity for digital
audio, but things are improving rapidly:
see figure 1 . The coming ten years

should see ROMs with enough capacity
to replace the vinyl disc and pre-recorded
cassette. Even the ‘home recording' days
of tape and cassette are numbered,
EAROMs (Electrically Alterable ROMs)
can be programmed by the user, and it
won't be long before they, too, have
enough storage capacity.

Audio will soon be 'all solid state')

Goodbye, organs!

Not the musical instrument variety, The
human body contains a large number of
vitally important organs. Those who
have the misfortune to be forced to do
without one of these, find themselves in
a very awkward situation. Some artificial
organs do exist — artificial kidneys, for
instance - but they are terribly clumsy
things.

As electronic circuits become ever
smaller and our understanding of how
the human body works becomes ever
greater, the chances of building artificial
replacements are increasing. As a first
step, we might have artificial limbs that
are controlled by the existing nervous
system, and have a sense of 'touch'.
Later on, vital organs. Suitable trans-
ducers could translate all relevant
biochemical and mechanical data into
electrical signals. These would go
to a Central Processing Unit (a
'bioprocessor'?) that is programmed to
take the correct action — again via
suitable transducers. The whole unit
would work on the following lines: if
the concentration of substance X in the
bloodstream is higher than value Y,
convert it into substance Z.

A pipe dream? No, it'll come. Not in the
next ten years, perhaps, but we should
certainly see the first steps in that
direction.

Goodbye, home constructor?

Will electronics last, as a hobby? We
don't mean do-it-yourself repairs of
factory-made equipment: that's not a
hobby, that's bitter necessity . . .

More and more interesting circuits are
becoming available ready-made, often
using custom-designed integrated circuits
or 'hybrid' technology. And cheap, too:
they often cost less than the components
needed for a home-construction job. So
what's the future in our hobby?

In the first place, building your own
equipment has the advantage that you
can include all the most recent develop-
ments — before they reach the shops,
even! Also, you can build it exactly the
way you want it, make it do exactly
what you want — not what some
designer thinks you want.

Then there's the challenge to your
ingenuity. Designing something that
nobody has ever made before. Or using
components for some job that they
were never intended to do. As an
example, think of the speech distorter
(T alk funny') we described last month.
EXAR introduced the 2206 as a
function generator 1C — but it makes a
very good ring modulator for sound
effects.

Furthermore, as with any hobby: it's
something to do! Collecting bottle-tops
has no economic value, but a lot of
people enjoy doing it. Electronics, as a
hobby, has the added advantage that it
always has something new to offer.
Admittedly, that's also the frustrating

New components are being introduced
almost every day. The manufacturers are
delighted when one of their new
products is discussed at length in a
magazine like Elektor. But when it
comes to actually supplying it to
retailers, their enthusiasm often seems
to evaporate. Then, by the time it
finally does become available, some
other manufacturer announces some-
thing even newer and better. Once
again, of course, the home constructors
come right at the bottom of the waiting
list.

Not that we have much ground for
optimism, but the least we can do is
ask: Please, all you manufacturers out
there, please start reserving a reasonable
proportion of the first batch for the
retail trade! We would be most grateful.

Goodbye, bulk!

Electrical equipment is bulky. TV sets,
loudspeaker cabinets, even lighting
fixtures - they're all ugly obstacles,
when you come to think of it. Oh yes,
we know that 'industrial designers' are
doing their best to fit all those bits and
pieces into boxes that don't look too
bad — but, if you ask us, they're fighting
a losing battle.

In the coming years, things should
improve. Components are becoming
smaller and flatter, so that it is becoming

easier to design enclosures that look
neater. For television sets, the 'flat
screen' is just around the corner. In fact.
National has already introduced a
'pocket-sized' version. Philips had the
technology within its grasp fifteen years
ago, if only they'd realised! Now,
suddenly, people are taking a new look
at electro-luminescent panels.

Flat loudspeakers have been tried.
Remember the 'Isoplanar'? It looked
like a picture, hanging on the wall, but
music came out of it. It was a pity that
the quality of both sound and picture
left a lot to be desired. Electrostatic
loudspeakers are famous, but you're not
supposed to place them flat against the
wall. For that matter, they're not
particularly pretty ornaments in a living
room. However, a lot of very clever
people are still working on the problem.
Within the next ten years, a truly 'flat'
loudspeaker for high fidelity music
reproduction may well become available.
Flat 'lamps' should become available in
the near future. Already, so-called panel
lighting is well-known: the lamps are
mounted behind flush-fitting translucent
panels in ceiling or even floor, giving
evenly distributed light. The next step is
a flat panel that gives enough light —
without mounting a conventional lamp
behind it.

We could go on, giving more examples,
but the general message is clear: the
'functional units' in our homes will
become smaller, less obtrusive. This
means that there will be more room for
things that are just 'nice to look at' —
whether they serve any 'functional'
purpose or not. Plants or art objects,
maybe, or even just room to move
around in without feeling cramped!
Goodbye, drudgery!

Automation at work: some people are
afraid it will cost them their jobs.
Automation at home: very welcome.
The effect of automation in the next
ten years will depend on our own
wishes, to a very large extent. One thing
is clear: work will be valued differently,
and it will have to be distributed more

Working hours will be shorter, as more
and more of the work is done by
machines. This is nothing more than a
continuation of a process that has been
going on for a century or more: the
'working week' has halved, from eighty
hours to forty, and are we any the
worse for it?

The main result is: more 'free time'.
Which brings us to the main question:
what are we going to do with it? A lot
of people spend most of their 'spare
time' doing things that they'd really
prefer not to do. Mowing the lawn, for
instance, takes a lot of time; but there
are very few people who consider
mowing lawns an ideal hobby. We are
not the first to predict the introduction
of an 'intelligent lawn mower': a gadget
that does the job without you having to
run behind it.

Automated kitchen? Yes, definitely, by
overwhelming majority vote. We've
already got 'programmable' ovens, that
can almost cook a meal on their own.
This trend is bound to continue.

A 'house-keeping computer'? Why not?
In both meanings of the word. If we're
going to have a programmable lawn
mower, we may as well have a pro-
grammable vacuum cleaner. Controlled
by a central 'domestic computer', that
can also take care of central heating and
turning on the lights and closing the
doors, in a few spare milliseconds. At
the same time, it can be used as an
invaluable aid for balancing the house-
hold budget.

While we're at it, let's go a step further.
If we link our home computer to a
bigger, urban computer, it can keep us
posted on the price of meat, do our
banking, replace our twenty-volume
encyclopedia, replace our letter-box,
even? And still have time to spare.

All in all, a lot of the chores that eat up
our spare time at present will be taken
off our hands in the next few years.
Which leaves us with truly 'free' time.
So what do we do with it? Something
creative?

What about modelling in clay. You
can't? How do you know, have you ever
tried? Maybe an electronic tutor would
help. Using a laser, a three-dimensional
image can be created 'in thin air'. This
can be used as a 'mould': you shape the
clay until it fits the image. After gaining
some experience this way, the step to
'free modelling' should be relatively

Computers can also be used to design
patterns for dress-making or needle-
work. They can make an ideal 'scratch-
pad', showing the finished article from
various angles; when you're satisfied,
they can draw the necessary patterns.

For that matter, computers are a hobby
in themselves. A lot of people enjoy
building their own computer, and then
working out programs for it. Which
brings us back to the present, and
Elektor. We will do our utmost, in the
next ten years, to provide you with an
interesting magazine! 14

colour

The circuit consists of three almost
identical sections. The heart of each
section is a Siemens 1C, type S 566B",
that was originally intended for a touch-
activated light dimmer (see Elektor,
Summer Circuits 1978, p. 7-90). When
the control input of this 1C is operated
continuously, it will provide a drive
signal to the triac that causes the lamp
to fade up and down periodically. The
complete cycle time is seven seconds.

By using three of these ICs and off-
setting their cycles by 2'h seconds.

generator

Figure 2. Block diagram.

using coloured light
for an effective display

Circuits for producing light effects
are well-known - but this one is
different. It generates a fairly
constant amount of light, shifting
slowly through the spectrum.
When used for indirect lighting, it
produces a beautiful effect that
can be useful in all kinds of
applications: parties (both indoors
and outside), shop windows, or
even as a permanent 'light play' in
the garden.

W.H.M. van Dreumel

three overlapping cycles are obtained as
illustrated in figure 1 . Each 1C is used to
control a lamp that gives one of the
primary colours: red, green and blue. If
all three lamps are aimed at the same
white background, the total colour will
sweep through the colour spectrum in
7 seconds.

A block diagram of the circuit is given
in figure 2.

The circuit

The complete circuit is given in figure 3.
The basic circuit configuration is similar
to that for the touch dimmer mentioned
earlier. The circuits around T4 and T5
provide the correct initial time delays
when the circuit is first switched on. PI
and P2 are used to set these initial
delays to 2‘/j and 4 2 h seconds, respec-
tively. The design of the ICs themselves
ensures that this offset will be main-
tained indefinitely, once they have been
started at the correct moments.

As in the original circuit, R1 and Cl are
used to derive the 15 V supply from the
mains. A zener diode, D1 , does the
actual stabilising.

The printed circuit board and com-
ponent layout are given in figures 4 and
5. Note that the complete circuit is
connected to the mains. This means that
the circuit must be built into an
insulating case; particular care must be
taken when adjusting the initial time
delays. M

•Note that the NE566 cannot be used in this
circuitl That 1C is a function generator - it
has nothing in common with the S 566B.

R1 - 330 n/1 W
R2.R5.R10 = 1M5
R6.R7.R11 = 4M7
R3.R4,R8,R9,R12,R13 - 470 k
R14.R16.R18 * 10 k
R15,R17,R19 = 1 20 H
R20.R21 = 100 k
PI ,P2 = 470 k preset

Capacitors:

Cl = 220 n/400 V
C2 = 47 u/25 V
C3.C6.C9 = 470 n/400 V
C4,C5,C7,C8,C10,C1 1 - 47 n
C12,C13,C14 ■ 150 n/400 V
C15.C16- 100 n

Semiconductors:

IC1.IC2.IC3 - S 566B (Siemens)

T1 .T2.T3 - BC 1 07B. BC 547B or

T4.T5 = BC 177B, BC 557B or
equ.

Tril ,Tri2,Tri3 = 2 A/400 V triac
le.g. TIC 226D,

D1 “ 15 V/1 W zener diode
D2 = 1N4001

03,04 - 5.6 V/250 mW zener

Sundries:

LI .L2.L3 - 50 mH/ 2 A (ring core)
F 1 ,F2,F3 = 2 A sloblo

(p.c.b. mounting)

Lai = blue lamp, max. 400 W
La2 = green lamp, max. 400 W
La3 = red lamp, max, 400 W

O— K»2*>

’ oHHj“

iponent layout.

1-08 - i

It's not so easy to add digital frequency LED display direct — i.e. without any
indication to an existing receiver. The additional transistor buffers,
design must be small enough to fit in The counter is crystal-controlled. It can
the available space; it must work on all be programmed to correct for different
wavebands, it must be easy to install; IF frequencies, so that the same unit
the total cost price must be kept within can be used for all AM and FM re-
reasonable limits. All in all, quite a tall ceivers. Furthermore, channel number
order! indication can be selected on the

The trouble with most designs of this VHF-FM band instead of frequency
kind is that they use an expensive indication,
frequency divider (a 95H90, for in-
stance), to bring the high oscillator
frequency down to a value that can be
handled by normal frequency counters. The system

What you really want is one 1C, specifi- The basic principle is quite straight-
cally designed for this kind of work, forward. The oscillator frequency of the

no strings attached Recently, Valvo introduced two ICs: receiver is measured, and the IF fre-

digital Inning scale

The electronics inside radio receivers have changed
radically over the years, but tuning scales remain
basically the same. The ‘bent wire on a string' principle
is still the most popular. A more 'sophisticated' system
in the most expensive of modern receivers is digital
frequency indication. In this article, an 'add-on'

quency is subtracted from this to
obtain the actual transmitter frequency.
As such, the IF frequency is not available
in the receiver, of course. The SAA1070
derives this frequency from a 4 MHz
crystal oscillator, by means of a fre-
quency divider. A whole range of IF

digital tuning scale is
described.

frequencies can be obtained by selecting
the correct division ratio.

In practice, it is not the oscillator signal
that is fed to the 1070. The first step in
the chain is the SAA1058; the internal
block diagram of this 1C is
given in figure 1. The outputs
from the AM and FM oscil-
lators are fed to the two pre-
- amplifier inputs. This is fol-
lowed by a six-stage divider,
that can be 'blocked' by an
external gate signal (derived
from the SAA 1070). An output
buffer stage boosts the signal to a
comfortable level for the 1070.
The SAA 1070 is a rather more compli-
cated 1C, as the internal block diagram
(figure 2) shows. It is a complete
frequency counter with LED drive
capability — and some more besides. To
limit the power dissipation (and the

GATE CM32

1-12 — ele

january 1980

number of pins needed on the 1C!) the
display is subdivided into two sections,
and these are driven alternately. This is
called 'duplexing'. The necessary control
signal is derived from the mains by two
simple half-wave rectifiers.

The data to be displayed are stored in
the display register. These data can only
be modified once every three counts;
even then, they are only updated if a
new value must be stored. This system
reduces 'display flicker' to a minimum.
During each count cycle, the dividers in
the 1058 are enabled via the 'gate
control' section in the 1070, and the
internal counter receives the output
signal from those dividers. When the
count is complete, the 'IF preset'
(stored in ROM) is subtracted and the
result is compared to the existing
display data. If necessary, the new data
are now stored in the display register.
The 'IF preset ROM' contains the data
for a whole series of different Inter-
mediate Frequencies. The correct data
are selected by applying corresponding
logic levels to the 'waveband select'
inputs.

A block diagram of the complete circuit
is given in figure 3. The oscillator
signals (from, the AM and FM oscillators
in the receiver) are fed to the SAA1058,
for initial frequency division and gating.
The output from this 1C goes to the
SAA1070. A 4 MHz crystal and the
'50 Hz duplex' provide the necessary
reference frequencies. The five-way
selector switch determines the wave-
band, and 'IF frequency preset’ resistors
determine the corresponding IF fre-
quency that must be subtracted. The
seven-segment displays are driven by the
SAA1070.

Figure 6. In some cases, input preamplifiers may be required. The two versions given here are
for VHF-FM and AM use, respectively. Resistors R/\ and Rg can be given other values if a
different supply voltage is available in the receiver.

The circuit

The complete circuit of the digital
tuning scale is given in figure 4. The two
oscillator signals are fed through coupling
capacitors Cl and C2 to the correspond-
ing inputs of the SAA1058. The DC bias
for the preamp is derived from the
internal reference voltages, via R3 and
R4; the positive supply is decoupled by
means of R7 and C5. The CM32 control
input' to the dividers is grounded,
setting the division ratio to 1 : 32.
The 1C has an open collector output
(pin 8), so R9 is included as collector
resistor.

The output signal from IC2 goes to a
voltage divider (R10, R11 and R12),
and from there to the signal input of
IC3 (pin 12). The positive supply to the
SAA1070 is decoupled by L2 and C8.
Resistor R14, from positive supply to
the gate control circuit, is included to
ensure a reliable start of the internal
'process control' when the circuit is
switched on.

The bulk of the crystal oscillator
circuit is included in the 1C; the only
external components are capacitors C9,
CIO and Cl 1 , trimmer C19 and the
4 MHz crystal. The oscillator frequency

must be set to exactly 4 MHz. This can
be measured at pin 18; note, however,
that capacitive load at this point
detunes the oscillator by -4 Hz per pF.
In other words, if a 10 pF probe is
used, the frequency should be set to
3.999960 MHz (i.e. 4 MHz - 40 Hz);
when the probe is removed, the oscil-
lator frequency will be 'spot on'.
Alternatively, of course, the receiver can
be tuned to a known frequency, after
which the oscillator is adjusted until the
correct display is obtained. In this case
the trimming screwdriver will detune
the oscillator slightly, so that some
patience is required . . .

Switch SI is the wave-band selector. If
you're lucky, the existing selector
switch (or pushbutton block) in the
receiver may have a set of spare contacts;
if not a separate switch will have to be
added. Switch positions 1 and 2 both
apply to the VHF-FM band: in position
1 the frequency is displayed, whereas
position 2 gives a display of the channel
number. Position 3 is for the short-
wave band and position 4 for long and
medium waves. Finally, position 5 is for
testing the displays: ail segments should
light in this position.

The correct IF frequencies are selected

Table 1

frequency

R45 R42 MHz

0 0 10.70

0 0 10.60

0 0 10.6125

0 0 10.625

1 0 10.6375

1 0 10.65

1 0 10.6625

1 0 10.675

0 1 10.6875

0 1 10.70

0 1 10.7125

0 1 10.725

1 1 10.7375

1 1 10.75

1 1 10.7625

1 1 10.775

ary 1980 - 1-13

by means of resistors R37 . . . R45, as
will be discussed later (see 'calibration').

Construction

Two printed circuit boards are used, as
shown in figure 5: a 'main board' and a
'display board'. The two boards are
mounted at right angles and inter-
connected by means of short wire links,
as shown in the photo.

The position of the decimal point in the
display is fixed; two LEDs are used to
distinguish between kHz and MHz.
The two coils (LI and L2). for supply
decoupling of the ICs, each consist of
three turns of 0.3 mm enamelled
copper wire on a 5 mm ferrite bead.

Care should be taken when mounting
the voltage stabiliser 1C (IC1): its metal
'back' should be towards R6. In other
words, if it is bent over as shown in the
component layout the plastic 'front'
will face the board. A small heatsink is
sufficient (1.5°C/W).

The total current consumption is quite
small; an 8 V/600 mA transformer will
be adequate.

Connections to the receiver

In some cases, the oscillator signals can
be taken direct from the receiver
without any special precautions. For
this to be possible, two conditions
must be met: a point must be found in
the existing oscillator circuit where the
impedance is (considerably) less than
1 k. and the oscillator must be able to
drive this additional load without
noticeable detuning.

An alternative solution is to use a
pick-up coil, so that no direct (soldered)
connection is required. Even then, care
must be taken to avoid undue loading
of the oscillator. In most receivers, the
complete tuner is contained in a
screening box - both to reduce the
sensitivity to stray fields and to 'protect'
the rest of the receiver from the strong
field produced by the oscillator. Holes
are provided in this case, for trimming
purposes; one of these holes will corres-
pond to the oscillator coil, and it can
be used for inserting the pick-up coil. A
suitable coil diameter is chosen: small
enough to fit through the hole, and
certainly not larger than approximately
6 mm (%") diameter. The coil consists
of three turns of enamelled copper
wire; any wire diameter between 0.3
mm and 0.6 mm can be used.

The oscillator coil in the receiver can be
located by inserting a screwdriver in the
various holes (without turning any-
thing!). The hole where maximum
detuning is found will correspond to
the oscillator. In some tuners (a few of
the TOKO types, for instance) the
oscillator coil can be recognised as the
only one with an aluminium core (in-
stead of ferrite).

The connection between the pick-up
coil and the digital tuning circuit should
be made with 50 ... 75 coax cable.
The pick-up coil is lowered through
the hole in the tuner until a stable
frequency display is obtained, without
serious detuning of the oscillator.
Should this prove impossible, an ad-
ditional preamp will be needed. The
circuit given in figure 6 boosts the input
sensitivity to better than 3 mV. More

find the car electrical system that runs
on 50 Hz AC! The LED display 'duplex-
ing' is based on AC drive, so this must
be simulated if the circuit is to be used

A suitable supply circuit is given in
figure 9. No p.c. board has been designed
for it, but it shouldn't be too much
of a problem to mount it on Veroboard
or something similar. The letters A, C
and D at the various outputs correspond
to the three supply inputs to figure 4.

The last word . . .

The 'oscillator' inputs of the digital
tuning circuit are quite sensitive, so they
must be adequately screened.
Furthermore, the circuit itself produces
quite a bit of interference, so it must be
mounted in a screened box. In the
prototype, this was made up from
sections of copper laminate board. M

Before going into the actual circuit, it is
a good idea to take a brief look at how
these little DC motors work. Why does
the speed drop when the motor is
loaded?

Normally, a fairly constant voltage is
applied to the motor. Off load, the
speed increases until the power
consumption is exactly sufficient to
cover the electrical and mechanical
losses in the motor. When the motor is
loaded, the speed drops. This reduces
the back EMF, so the current through
the motor increases; a new equilibrium
is reached when the increased power
consumption equals the reduced electri-
cal and mechanical losses plus the power
delivered to the load. In other words,
the motor supplies the power required
by the load — but at reduced speed.
Obviously, there is a limit: if the motor
is loaded too heavily, it will stop.

If the speed is to remain constant, the
voltage across the motor will have to be
increased when the motor is loaded. In
this way, the current (and the power
output) can increase without affecting

more power is supplied to the motor —
counteracting the tendency for the
speed to drop.

Basically, this is a feedback system -
and positive feedback, at that. For
correct operation, the amount of
feedback must obviously be set accu-
rately. One solution would be to use a
preset potentiometer for R2. This is not
very practical, however: where do you
find a 4.7 J2 pot that will happily
tolerate a current of up to 1 A? Adding
P2 is an infinitely better solution. With
its slider turned right up, the circuit
becomes identical to that given in '
figure 1, as far as the regulator is con-
cerned; the voltage across the motor is
held constant. As the slider of P2 is .
turned down, more and more positive
feedback is added. With P2 set correctly,
the motor speed will remain almost
constant, independent of load.

Construction

A suitable printed circuit board is given
in figure 3. The only components not |

mini drill s|mmhI control

Miniature electric drills have been In the circuit described here, the it

former, fuse, and potentiometer PI .

available for some time. Most of
them are battery powered. For
precision work, it is useful to have
a speed control; if constant speed
can be maintained, independent of
the load, so much the better.

Both of these objectives can be
achieved fairly simply, using an
integrated voltage regulator.

active component is a voltage stabiliser
1C, the 79G. This is a negative voltage
regulator; it was chosen because its
output voltage can be reduced to as low
as —2.23 V. The minimum output
voltage of its positive voltage counter-
part, the 78G, is approximately 5 V.
The extended control range at the low
voltage end is important, since the
motors in miniature drills are all fairly
low voltage types - they are intended
for battery use. This circuit can be used
to power 2.5 ... 12 V motors, with any
current rating up to 1 A.

As figure 1 illustrates, the basic regulator
circuit using this 1C is very simple. The
output voltage is determined by the
ratio between the two resistors, as
follows:

.. _R1 + R2 ..

u out ~ — — — x ^control

For the 79G, U CO ntrol is -2.23 V.

As can be seen, the output voltage of
this regulator is determined by the
voltage on the control input - i.e. that
at the R1/R2 junction in figure 1. To be
more precise, it is the voltage between
the control input and the 'common'
connection that sets the output voltage.
Knowing this, the actual circuit
(figure 2) is not so difficult to under-

When the motor is loaded, its speed will
tend to drop. The current through the
motor increases, producing a larger
voltage drop across R2. The 1C will now
try to restore the original voltage
difference between the 'control' and
'common' connections, by increasing its
output voltage. This, in turn, means that

Having built the circuit, and before
connecting the motor, an initial check is i
advisable. The slider of P2 is turned
fully clockwise. Power is then applied, j
and PI is set to maximum resistance —
corresponding to the maximum output
voltage. This voltage (between the '+'
and '— ' output terminals) is measured.

It should be safely below the maximum
permissible motor voltage — say about
20% down. If it is too far off this value,
the value of R1 will have to be modified:
increasing R1 reduces the voltage, and
reducing R1 brings the voltage up,

PI is now turned back about halfway,
and the drill is connected. Preset P2 is 1
carefully adjusted so that the motor
speed is on the verge of increasing. The
idea is that too much feedback will
cause the drill speed to run right up, out
of control; too little feedback, on the
other hand, will make the circuit less
effective. It is possible that, with a given
motor, even the lowest setting of P2 is *
not low enough: the speed still drops
when the motor is loaded. In that case,

Figure 1. In the basic regulator circuit, the 1C
adjusts the output voltage to maintain a

constant -2.23 V between its control input
and the 'common' connection. This means
that the output voltage is determined by R1
and R2.

the value of R2 will have to be increased
and the calibration procedure repeated.
Obviously, this circuit is no miracle
worker. If the motor is loaded further
when it is already running flat out, at
maximum voltage, the speed will drop.
Which is just as well - a higher voltage
than the maximum permissible will burn
out the motor. This is why it is so
important to select the correct value for
R 1 — it determines the maximum
voltage that can be applied to the motor.
For that matter, it is a good idea to
check this again once P2 has been

loaded more and more. It should not
run up to more than 20% above the
nominal motor voltage; if it does, the
value of R1 will have to be increased
further. Alternatively, a resistor can be
included in parallel with PI - reducing
the maximum resistance value that can
be set by this potentiometer.

There is no need to worry about
damaging the 1C — it is internally
protected against output short-circuits
and thermal overload. M

Semiconductors:
IC1 - 79GU
D1 - 1N4001
B1 = B40C1500

Sundries:

Tr- 18 V/1 A trf
F » 100 mA fuse
Heatsink for IC1

Capacitors:

Cl = 2200 p/35 V
C2 = 2p2/35 V tantalum
C3 ■ 100 p/16 V
C4.C5 = 1 m/ 25 V tantalur

1-18 - elektc

VSWR

VSWR

meter

To get the maximum power output from a transmitting aerial, it must
be properly matched to the transmitter. For that matter, a really
serious mismatch can spell disaster for the (expensive) output stage of
the transmitter. For these reasons, amateur radio operators often make
use of a so-called voltage standing wave ratio meter - an 'SWR' meter.
In this article, we will not only describe the theory — what it is, what
it does, how it works; a practical circuit for an SWR meter with a very
wide frequency range is also given. With a p.c. board, of course.

Nearly everybody who has any experi-
ence of electronics has heard of
impedance matching. But how many
people really know what it's all about?
Not that it is so important, in most
cases — in fact, one of its prime purposes
would seem to be that it provides audio
fanatics something to discuss when
they've exhausted every other possible
topic . . .

There is, however, a group of electronics
enthusiasts that rightly considers im-
pedance matching to be of prime
importance. Amateur radio operators!
For them, an impedance mismatch can '
have disastrous results. At best, their
range will be drastically reduced; at
worst, they may blow up the output
stage of their transmitter.

Fortunately, it is not too difficult to
avoid mismatching. Provided you know
what you're doing, that is! If you buy a
transmitter ready-built, the output
impedance is usually specified. The
same is true for transmitting aerials: the
manufacturer will normally specify the
impedance. The idea is that you use an
aerial with an impedance that matches
the output impedance of the transmitter,
and that you use connecting cable with
the same characteristic impedance. For
instance, if the transmitter output is I
specified as 75 SI the obvious thing to
do is to use 75 £1 coax cable and a 75 SI
aerial.

However, life is not always so simple. I
Many amateurs not only build their own I
transmitter; the transmitting aerial, too,
is often the result of personal exper-
iment. In this case both impedances are
unknown, so that optimum energy
transfer from transmitter to aerial can
only be obtained by experimenting.

Why?

The simplest possible equivalent circuit
for a transmitter with its load is given in
figure 1. An ideal AC voltage source, U,
supplies power to the load, Z 0 , via the
internal impedance, Zj. The power
supplied to Z 0 can be calculated as
follows:

°”(Zi + Z 0 ) 2

In this formula, U is the (open-circuit)
voltage supplied by the voltage source;
P 0 is the power supplied to the load Z 0 .
For a given value of Zj, the maximum
power is supplied to the load when Z 0
equals Zj. In other words, if the output
impedance of the transmitter is 75 SI, a
cable with a 75 SI characteristic
impedance should be used. Similarly,
the power transfer from the cable to the
aerial is highest when the aerial im-
pedance is equal to that of the cable. In
that case, all the power supplied by the
transmitter is pumped into the aerial
(the losses in the cable will normally be
negligable). What happens when the
aerial impedance does not match the
characteristic impedance of the cable?

' The energy transfer from cable to aerial
j is not ideal in this case, and a standing
wave appears along the cable. This can
( be clarified as follows.

Normally, the aerial is not mounted on
top of the transmitter. It will be on the
roof, or at the top of a high mast, or in
some other suitably high position. In
most cases, it will be some distance
away from the transmitter — the latter
being at a more comfortable location
inside the house. The transmitter and
aerial are then linked by a cable. In
other words, the power output from the
transmitter runs down the cable to the
aerial. All very straightforward, you
would think, but let's take a closer look
at what happens in the cable.

The electrical signal moves down the
cable at high speed: 200,000 to 300,000
I • km/s (i.e. well over 100,000 miles per
; second!). This seems very fast, but let's
see how long it takes the signal to pass
l , down a 60-foot cable (20 metres). A
simple calculation (using the metric
figures it's simple, anyway!) shows that
i it will take about 100 ns. For a radio

c amateur, 10 MHz is not such a high

, frequency — but the period time at this
} frequency is also 100 ns. This means
that the AC voltage source in figure 2
i will have produced one complete period
5 before any signal appears at the other
I end of the cable, 20 metres further on!

3 This means that the voltage source can't
3 see' whether or not the cable is termin-
ated with the correct impedance. The
I transmitter only 'sees’ a small part of

the cable, and recognises its 'character-
istic impedance'. The current pumped
into the cable is therefore determined
by this cable impedance. When this
current arrives at the aerial, the results
depend on the aerial input impedance.
If the whole system is properly matched,
the complete power output from the
transmitter goes into the aerial: the
same voltage across the same impedance
corresponds to the same current — i.e.
the current coming down the cable. If
the aerial impedance is too high, how-
ever, it will 'reflect' some of the power.
Put it this way : the same voltage across
a larger resistance corresponds to a
smaller current. Some of the current
coming down the cable is 'left over', and
it bounces back down the cable towards
the transmitter. The same sort of effect
occurs if the aerial impedance is too low.
The power that is reflected back to the
transmitter interferes with that coming
the other way. The result is a standing
wave. This can actually be detected by
passing a field-strength meter along the
cable: at some points a maximum is
found, and at others the field strength is
at a minimum. The maxima and minima
occur at regular intervals. If the aerial is
incorrectly matched, the field strength
at that point will below - corresponding
to low output from the aerial.

If the aerial impedance is unknown, the
degree of mismatch can be determined
by measuring how much energy is
reflected. By using directional couplers
that only pass power in one direction,

the reflected power can be separated
from the power fed to the aerial. The
ratio between these two is a measure for
the accuracy of the impedance match.
To be more precise, the ratio between
sum and difference of the forward-
going voltage (Uf, towards the aerial)
and the reflected voltage (U r ) is taken,
as follows:

VSWR is the Voltage Standing Wave
Ratio.

It will be obvious that the VSWR equals
1 if the reflected voltage is zero; it
becomes infinite if the complete signal
is reflected. This would occur if the
aerial impedance is zero or infinite.
Note that the aerial impedance referred
to is that at the transmitted frequency.
If the aerial is correctly designed for this
frequency, it will be in resonance and
its impedance will be real.

The VSWR meter

By now we have some idea of what we
want to measure. The next question is:
how?

The VSWR meter circuit given in
figure 3 can be used for transmitting
frequencies between 2 MHz and 30 MHz.
The unit is connected in series with the
cable, close to the transmitter. The
current flowing from transmitter to

1-20-

aerial and vice versa passes through the
primary of a transformer. Both of these
currents produce a current in the
secondary; the direction of the current
flow in the secondary is obviously
determined by the direction in the
primary. By combining the total voltage
(Uf + U r ) with the correctly chosen and
half-wave rectified secondary voltage,
U r and Uf can be obtained separately.
These voltages (across C2 and Cl,
respectively) can be measured with a
simple meter circuit, consisting of
low-pass filters (R1/C3 and R2/C4) and
a meter with preset series resistor.

At frequencies above 30 MHz, no
transformer is needed. The same job can
be done by adding two secondary 'strips'
that run parallel to the main through
feed. This is shown in figure 4. The
directional characteristics of this circuit
are best when the following conditions

p 1 _ Z 1L . P2 ZlL

zir 2r'" d zi:--zr

where Z a is the impedance for which
the unit is intended.

Obviously, electrical waves can run in
both directions along all the strips;
however, if the above conditions are
met, waves in one direction will decay
quite rapidly. With the diodes only
conducting in one direction (as all good
diodes should do . . .), the forward-
going wave will build up a voltage across

C2 and the reflected wave can be
measured across Cl . As before, the two 6
voltages can be measured with a simple
meter circuit. The frequency range of
this VSWR meter is approximately
100 MHz . . . 300 MHz.

The p.c. board

There is no need for two separate p.c.
board designs for the two circuits. There
are so few differences between the two
that the same design can be used for
both, as shown in figure 5.

If the circuit given in figure 3 is to be
mounted (for measurements at up to
30 MHz), the centre stripline must be
divided into two halves. This is done by
scratching away some of the copper
between the two central holes. The
transformer is wound on an Amidon
ring core. The secondary winding
consists of 30 turns; the primary is only
half a turn, which is equivalent to passing
a wire link through the ring. This link is
soldered into the two holes in the
central strip, effectively fastening down
the transformer at the same time. This
construction is clearly illustrated in the
photo.

Mounting the high-frequency version
(figure 4) is even easier. The central
strip is left intact, of course. The only
important point to watch is that C5, C6
and the transformer are omitted.

■aKHMIHIKHajiHi

R1.R2- 1 k
P1,P2 = lOOnpre
P3* 10 k preset

Capacitors:

Cl ,C2 - 330 p (ceramic)
C3,C4 = 100 n
C5*,C6", - 10 p (ceramic)

Sundries:

Ml = meter, lOOpA f.s.d.

Trl * - Amidon ring core,
type T50-6;

primary: 0.5 turns, 1 mm CuAg;
secondary: 30 turns. 0.5 mm Cul
SI = single-pole change-over

• These components are only
required for the low-frequency
version (up to 30 MHz).

For both circuits, calibration is quite
easy. The unit is included in the cable,
close to the transmitter. First, the
switch is set to the 'Uf' position, and P3
is adjusted so that a fairly high reading
is obtained on the meter. Note that
these three components are not mounted
on the p.c. board. Now, with the switch
set to the 'U r ' position, P2 is adjusted
for minimum deflection.

After this adjustment, the circuit is
connected 'backwards': the 'aerial
output' is connected to the transmitter,
and the 'transmitter input' goes to the
aerial. The same calibration procedure is
repeated, but this time P2 is left
untouched and PI is adjusted for
minimum deflection in the 'U r ' position.
After restoring the original connections,
the adjustment of P2 can be checked;
then PI again, and so on until no

further improvement can be obtained.
Having found the optimum settings for
PI and P2, the switch is set to position
'Uf' and P3 is adjusted for full scale
deflection. In the 'U r ' position, the
meter will now indicate the strength of
the reflected wave. If all impedances are
correctly matched, the meter deflection
will be zero. This corresponds to
VSWR = 1, as explained above; in other
words, the VSWR scale on the meter
runs from 1 to °°.

If a different aerial is tried, the meter
will now indicate how accurately it is
matched — or how badly ... If the
meter originally has a linear scale, it is
easy to fill in the correct VSWR values:
full scale is '<»', three-quarters of full
scale is '7', the mid position is '3',
one-quarter is '1.6', one-tenth is '1.2',
one-twentieth is '1.1' and zero is T. In
practice, a VSWR of less than 2 is quite
good.

frequency doubh

raise your guitar one octave

This frequency doubler circuit for guitars produces an output that is
one octave higher than the original input signal. An uncommon, but
particularly useful, feature is that the original signal and the frequency-
doubled version can be mixed in any desired proportion.

As can be seen in the block diagram
given in figure 1 , the signal from the
guitar is amplified and then fed along
two separate paths. The lower path
carries the basic signal; in the upper
path, full-wave rectification is used to
obtain frequency doubling. After a
'balance' control, the two signals are
summed; the output is at the correct
level todrivea suitable 'guitar' amplifier.
The complete circuit is given in figure 2.
In practice, it is rather more compact
than it looks: the four opamps are all
contained in a single 1C. The first stage,
A1, is an input preamplifier/buffer. The
gain can be varied between x 50 and
x 1, by means of PI. The values given
for R1 and Cl in the circuit may have
to be modified to suita particular guitar.
The correct input impedance is deter-
mined almost exclusively by the value
of R1; Cl will have to be modified
accordingly (if R1 is decreased, Cl must
be increased and vice versa) unless a
different low-frequency cut-off point is
desired.

The DC output voltage from A1 is 0 V
- in other words, it is biased mid-way
between the positive and negative
supply rails. This output is connected
direct to the non-inverting inputs of A2
and A3, so that these two opamps are
also biased to the mid-point. This
ensures that the maximum (symmetrical)
AC voltage swing is available throughout
the circuit.

Opamps A2 and A3 are used in a
full-wave rectifier circuit. For the type

doubler

of signals that we are dealing with,
full-wave rectification is equivalent to
frequency doubling - which is what the
whole exercise is about! The output
from A3 is fed to one half of a stereo
potentiometer (P2a); the 'basic' signal,
from the output of A1, goes to the
other half of this potentiometer. By
wiring one of the (linear!) poten-
tiometers 'upside down' (when the
slider of P2a is connected to the C4 end,
the slider of P2b is at supply common)
the desired 'balance control' is obtained.
The output signals for three possible
settings are given in figures 4, 5 and 6.
Figure 4 is the frequency-doubled signal
(P2a turned right up, so P2b is down);
figure 5 corresponds to the half-way
situation — original and doubled signal
in equal amounts; figure 6 is the original
signal only.

The final opamp, A4, is the summing
stage. It is actually a virtual-earth mixer,
with unity gain for both signals.

Construction

A printed circuit board design for the

After all the articles describing
the theory of vocoders, there must
be a lot of enthusiastic readers just
itching to build one. This is just
the sort of challenge that Elektor
designers love: a lot of people
want it, but nobody has come up
with a suitable design until now.
Now, here it is at last! A 10-
channel vocoder, designed
in collaboration with Synton
Electronics - an acknowledged
specialist in the field. The design
offers good performance at a very
reasonable cost. Ideal for
musicians with a lot of enthusiasm
but insufficient funds to back it

Certainly for those who would rather
wield a soldering iron than a formula,
the theory of vocoders has by now
been discussed in more than adequate
detail. Two years ago we discussed the
'hows and whys' and described the
basic principles of a few commercially
available vocoders. Last month's article
'Vocoders' was intended as a brief
recap of the history and technology
of vocoders, and at the same time as a
'warming-up exercise' for the construc-
tion project described this month.
The difficulties associated with design-
ing a vocoder were discussed at length.
Obviously, these difficulties are even
more apparent when the design is
intended for home construction, as
opposed to commercial production: the
circuits must be absolutely reliable, and
the effect of component tolerances
must be reduced to a minimum. Fortu-
nately, the problems are not insurmount-
able. as we will see.

One more time . . .

We've explained what a vocoder is,
often enough . . . 'We didn't really
oughta repeat ourselves'. However. For
those who are still unsure, in spite of
all explanations given in earlier publi-
cations here is a brief definition:

A vocoder is a 'box' with two inputs;
one for a speech signal and one for a

'carrier' or 'replacement' signal (in
practice, this is usually some kind of
'music' signal). Inside the 'box', the
speech characteristics are superimposed
on the carrier signal. A single output
signal results. It contains all the charac-
teristics (and intelligibility) of the
speech input, but the basic sound pro-
duced by the speaker (vibrations of
vocal chords, resonances in the oral and
nasal cavities) are replaced by those of
the music signal. The result is something
that sounds like the music, but talks as

ar vocoder ID

January 1 980 - 1 -25

interest of providing a smooth transition
to the block diagram and circuits that
are to come, let us take a quick look at
what is 'inside the box'.

Most vocoders are so-called 'channel
vocoders'. Other systems do exist
(heterodyne-based, for instance) but
these are so complex that they are rarely
used in practice. The Elektor design
is also a channel vocoder, so we will
forget the other possibilities. Last
month's article gave block diagrams
that illustrate the basic principle. A
quick look at the block diagram of the
Elektor vocoder (figure 1) shows that it
is almost identical.

A channel vocoder consists of two main
sections: the analyser and the syn-
thesiser. These are very similar, both
consisting mainly of an identical set of
filters (two groups of ten in the Elektor
vocoder).

In the analyser section, the filters are
used to split the incoming speech signal
into corresponding frequency bands.
The output from each filter is rectified
and passed through a low-pass filter; the
total result is a set of varying DC
voltages, each corresponding to the
'envelope' of the speech signal within
that particular frequency band.

The synthesiser section splits the 'carrier'
signal into the same set of frequency
bands. The output level in each band is
varied by a voltage controlled amplifier
(VCA) that is driven by one of the
varying DC control voltages produced
by the analyser section. The result is
that the amplitude 'envelope' of each
frequency band in the speech signal is
superimposed on the corresponding

frequency band of the carrier signal.
The outputs of all VCAs are then
summed, to produce the total output
signal: basically, the tonal charac-
teristics of the carrier signal with the
articulation of the speech. Talking
music, in other words.

The Elektor vocoder
After we had already done quite a bit
of experimenting with vocoder circuits,
we happened to come into contact
with Synton Electronics — the manufac-
turer of the well-known Syntovox
vocoders. Some very profitable dis-
cussions with these specialists led to the
circuit described here: a vocoder, de-
signed specifically for home construe-

The number of channels (frequency
bands in the analyser and synthesiser
sections) is limited to ten, for several
good reasons. That number is adequate
for good music reproduction and good
'speech' intelligibility; furthermore, it is
a reasonable compromise between per-
formance and price. Admittedly, a
twenty-channel version sounds better,
more 'detailed'; but, in practice, the
improvement is not often worth the
vastly greater cost and complexity
required. Not only do you need twice
as many filters: they must also be much
'steeper' (approximately 50 dB/octave)
and this requires careful design and
expensive components. Usually, strict
selection of components is neccessary
for this type of filter — not a very feas-
ible proposition for the average amateur.
For a ten-channel vocoder, on the

other hand, 24 dB/octave filters can be
used. These are not nearly as complex
and — even more important — quite
reliable results can be obtained without
having to resort to exotic components
or test equipment.

For that matter, reliability was an
important factor in the design of the
whole circuit - not only the filters.
Wherever possible, the circuit is set up
so that component tolerances and
wiring will not affect the operation;
furthermore, a larger number of adjust-
ment points are included than is normal
in professional equipment. By this
means, good results can be obtained
without the component selection
normally required.

Two features are deliberately omitted
from the basic version: spectrum
analysis and a voiced/unvoiced detector.
The reason is obvious: although ad-
mittedly useful, these features are also
expensive! However, the design does
provide the option of adding them at a
later date, and it is quite likely that we
will be publishing suitable designs in
the near future. For the present,
however, we will do without. One little
gimmick is included: ten LEDs, one for
each channel, give an indication of the
varying spectrum of the speech signal.
Not that it has much practical use — but
it doesn't cost anything, either.

What does it all cost? An important
consideration, for most people! From a
quick look at figures 3 ... 6 it is
apparent that there are quite a few
components in a vocoder. In plain
language: it's crawling with opamps. To
make matters worse, a lot of p.c. boards

Block diagram of the Elektor vocoder.

2. Block diagram of one filter unit.

Table 1

BPF 1
BPF 2

BPF 4
BPF 5
BPF 6
BPF 7
BPF 8

265 Hz 210- 320

390 Hz 320 • 460

550 Hz 460 - 640

800 Hz 640 - 960

1200 Hz 960-1440

1770 Hz 1440-2100

2650 Hz 2100-3200

3900 Hz 3200 - 4600

C1...C8 C9 CIO

82 n 220 n 33 n

56 n 150 n 22 n

39 n 100 n 15 n

27 n 68 n 10 n

18 n 47 n 6n8

12 n 47 n 6n8

8n2 47 n 6n8

5n6 47 n 6n8

C11

330 n
220 n
150 n

68 n
68 n

elektor vocoder (II

go into a unit of this kind — and they
are not nearly as cheap as we would
like. All in all, our estimate of the total
cost works out at somewhere in the
region of £ 100. A lot of money for a
home construction project - but very
cheap for a good vocoder, like this one. '

What's in the box?

A block diagram of the vocoder is
given in figure 1. The upper half is the
analyser section, and the lower half is i
the synthesiser.

Let's take a look at the analyser first.

The microphone signal is passed to a
suitable preamplifier. Although not
shown in the block diagram, the sensi-
tivity of this input is adjustable over
a wide range, so that it can also be
used as a line input from an external
microphone preamp. The preamplifier
is followed by a buffer stage that
includes bass cut, with a roll-off below
approximately 30 Hz.

The output from the buffer stage is fed
to the filters that split it into fre- '
quency bands. Ten filters, correspond-
ing to ten bands. Not all equal, however.
Taken together, the filters cover the
whole audio band, from about 30 Hz
to 16 kHz, but the first filter (low-pass)
and the tenth (high-pass) take care of
a disproportionately large part of the
spectrum. The low-pass filter covers j
the range from 30 Hz to 200 Hz; the

Table 1. In figure 3, '

capacitors Cl ... Cl 1

be selected from this table.

high-pass is for everything above 4600
Hz. The central range, from 200 Hz to
4600 Hz, is most important for speech;
it is divided into eight bands by the
remaining filters.

Each filter is followed by a precision
rectifier and a low-pass filter. The latter
is not shown, as such, in the block-
diagram — it is taken as an essential part
of the rectifier stage. Obviously: for a
vocoder, we are not interested in rapid
fluctuations of the speech signal or
remaining half- or full-wave rectified
frequency components; what we want
is the general level trend for each
frequency band.

The first stage in the synthesiser section
is also a preamplifier, for the carrier
signal this time. Once again, it is followed
by a buffer stage - similar to the one
the analyser. From here, the signal
passed to the filters; these are ident-
al to the first group. The output of
each filter goes to a voltage-controlled
amplifier (VCA). Each VCA receives
its control voltage from the correspond-
ing filter and rectifier in the analyser
section. The output signals from all
ten VCAs are summed; the total signal
is passed to the output buffer stage.
Finally, about all those dotted lines.
In both the analyser and the equaliser
section, the link between the input
preamplifier and the following buffer
stage is brought out, to create the

possibility of adding a voiced/unvoiced
detector at a later date. What happens
is that both outputs and both inputs
are brought out to a connector; from
there, they run along a bus board to
a further connector (intended for the
detector); along the way, the copper
tracks are deliberately bridged so that
each amplifier output is connected to
the corresponding buffer input. When a
voiced/unvoiced detector is to be added,
the bridge between the tracks must be

Furthermore, the connection between
each rectifier output and the correspond-
ing VCA control input is shown as a
dotted line. These points are brought
out to sockets on the front panel. This
has the advantage that it is now possible
to deliberately connect some or all of the
outputs to the 'wrong' VCAs, for
special effects. This will be discussed
later, in greater detail, when we come to
'using the vocoder'.

For the moment, we are more interested
in the electronic details of the various
sections shown in the block diagram.
Time for the circuits.

The circuits

A modular construction was chosen for
the vocoder, as we will see later on. The
various circuit sections are mounted
on separate printed circuit boards.
Twelve in all: one for the supply, one

for the input amplifiers and buffers plus
the summing amplifier and output
buffer, and one so-called filter unit
board. This contains one complete
section, as shown enclosed in dotted
lines in figure 1 : two complete high-,
low or bandpass filters with the as-
sociated rectifier and VCA. A more
detailed block diagram of one filter unit
is given in figure 2.

Since the circuit of the complete
vocoder is rather too extensive to
swallow in one gulp - for that matter,
it would be virtually impossible to print
on a single magazine page — it is easier
to deal with each circuit section separ-
ately. First the central building block
of the vocoder: the filter unit. In
particular, the band-pass filter version,
as it appears eight times with only
minor component value changes.

The band-pass filter

The circuit is given in figure 3. Those
who may have felt that we were exag-
gerating when we said that the complete
circuit was so extensive, should be
having second thoughts by now. All of
these components represent just one
filter unit — and there are ten of them
in our vocoder.

The band pass version shown here is
required eight times. Each one takes
care of its own band in the total range
(200 Hz . . . 4600 Hz), and this is ob-

viously reflected in the component
values. In particular, the values of
capacitors Cl ... Cl 1. Table 1 gives the
correct values for the bandpass filters
BPF1...BPF8, with the resultant
centre frequency of each filter.

On talking a closer look at the circuit
given in figure 3, it is not too difficult
to recognise the various sections that
make up the block diagram shown in
figure 2. First, let's pin down the in-
and outputs. Points 'a' and 'b' are the
filter inputs for the analyser filter
(speech) and synthesiser filter (carrier),
respectively: 'c' is the signal output -
the output of the VCA, in other words.
Point 'd' is the control voltage output
from the rectifier (more properly, from
the final low-pass filter) in the analyser:
V c ou t; 'e' is the control voltage input,
V c in. for the VCA in the synthesiser.
Al’and A2, with associated components,
make up the band-pass filter in the
analyser section. An identical configur-
ation, using A5 and A7, does the same
job in the synthesiser. The precision
rectifier is constructed around A3 and
A4; it is followed by the low-pass filter,
using A9. Finally, A10 is the VCA.
Admittedly, there are a few more
opamps - but these will be discussed
later.

One thing is very obvious: there are a
lot of opamps in this circuit. Not only
in this one, for that matter — the whole
vocoder is opamp-based. The main
reason for this is to keep the circuit as
simple as possible — using transistors.

it would really become messy . . . For-
tunately, the high-quality opamps that
are readily available nowadays are quite
suitable for audio work.

Most of the opamps used in this filter
unit are JFET-input types. There are
four of them in a TL084. Another
possibility is to use a 4741 - with the
added advantage that its current con-
sumption is lower. Both of these types
have been used in previous Elektor
designs, with good results, and avail-
ability should not be a problem. They
cost about one pound each. A common-
or-garden 741 is also used in the circuit
and - for the VCA — an OTA, type
CA3080. Quite familiar to Elektor
readers!

The band-pass filters are of a fairly
well-known type: in both sections, two
so-called Rauch filters are connected in
cascade. The slightly different com-
ponent values for the first and second
filter in each pair ensure that a slightly
■flattened' top is obtained for the total
filter characteristic, instead of the sharp
peak that a single filter would give. Each
filter gives a slope of 12dB/octave, so
that two in cascade provide the desired
24 dB/oct. In passing, it is perhaps
interesting to note that the slope of any
properly-designed filter can be estimated
by counting the 'active' capacitors and
multiplying by 6. A single filter in this
circuit contains two capacitors, making
for 12 dB/octave.

Back to the circuit. In the analyser
section, the band-pass filter is followed

by two opamps in a full-wave rectifier
circuit (A3, A4, D1, D2) and an RC
network (R30 and C9) to take care of
the worst of the ripple. An active low-
pass filter (A9) does the bulk of the
smoothing. It is a good idea to tailor
the low-pass filter to suit the frequency
range selected by the preceding band-
pass filter. For this reason, C9, CIO and
C11 are given different values for each
section, as listed in Table 1 .

The no-signal DC component in the
V c out control voltage should be zero,
in ’the ideal case. For this reason, an
offset adjustment (preset PI) is included
for A9. The LED indication of the
'speech spectrum' that was mentioned
earlier is obtained by using the same
control voltage to drive a LED (D3) via
a transistor (T1).

In the synthesiser section, the first
two opamps (A5 and A7) are used in
the same filter configuration as that in
the analyser. Then the VCA, for which
an OTA (A10) is used. Since an OTA
(Operational Transductance Amplifier)
is basically a current-controlled amplifier
- not voltage-controlled - a minor cir-
cuit extension is needed. The control
voltage from the analyser section
(Vc,in) is buffered (A6) and then fed
to a voltage-to-current converter: A8
and T2. Basically, this is a voltage-
controlled current source; variations of
the control voltage, V c , are converted
into variations in the bias current for
the OTA (at pin 5 of A10). P4 is used to
set a threshold value for this current —

january 1980 - 1-29

vocoder (1)

6a

Figure 6a. Microphone/line 'speech' input amplifier.

6c

the calibration procedure will be de-
scribed later. The same applies for the
calibration of P2, this adjustment is
included to balance the input differential
amplifier in the OTA — a necessary
precaution to prevent the bias current
variations breaking through to the
output, in the absence of a 'carrier'
signal.

Low- and high-pass filters
Figures 4 and 5 both bear a strong
resemblance to the circuit given in
figure 3. This is hardly surprising: the
only real difference between the band-
pass filter units (figure 3), the low-pass
(figure 4) and the high-pass filter unit
(figure 5) is the actual filter circuit. And
even there, the difference is marginal.
Both the low- and high-pass filters are
standard variants on the well-known
Sallen & Key filter. As before, two
sections are connected in cascade to
obtain a total filter slope of 24 dB/octave
(four capacitors, remember?). The cut-
off point for the low-pass filter is set at
200 Hz; for the high-pass filter, this is
4600 Hz.

In- and output module

The remainder of the vocoder proper is
shown in figure 6: the in- and output
circuits. These are all mounted on one
p.c. board.

For these sections, good signal-to-noise
ratio and drive capability are extremely
important. The 'ideal' opamp for this
job is the illustrious TDA1034 (or
NE5534). Tf availability is a problem,
an LF 357 can be used as a (temporary)
replacement — although the signal-to-
noise ratio will suffer.

The speech input circuit is given in
figure 6a. Opamp A31 is used as a very
low-noise microphone preamp. The
voltage gain can be set between xl and
xIOOO, for any input sensitivity between
10 mV and 7.7 V. The input impedance
is roughly equal to 10 kJ2, and in prac-
tice microphones with almost any im-
pedance can be used. A line input is also
provided, suitable for signals from an
external microphone preamplifier; in
this case, the gain is set to about x12.
The output from A31 is brought out,
via the bus board, to a spare connector;
from there, it comes back to the sen-
sitivity control PI 3. As mentioned
earlier, this is done to offer the possi-
bility of adding a voiced/unvoiced
detector at a later date. The sensitivity
control is followed by a buffer/amplifier
stage, A32. By adding C54 and C55, this
stage also serves as an active rumble
filter. Output 'a' from A32 is connected
to all ten inputs 'a' on the filter units.
Figure 6b is the 'carrier' input circuit.
The sensitivity control, P14, is followed
by an input preamplifier with a gain
of approximately xIO (A33). As before,
the signal then loops around the spare
connector; finally, A34 is used as a
combined buffer/amplifier/active bass-
cut filter — identical to the one in figure

- elektor january 1980

6a. Output 'b' is again connected to all
ten inputs 'b' on the filter unit boards.
The outputs of all filter boards (point
'c' in figures 3, 4 and 5) are all connected
to input 'c' in figure 6c: the input of the
summing amplifier. The first stage
(A35, an LM301) is followed by an
output level control (PI 5) and an out-
put buffer stage (A36). A TDA1034 is
used for this final stage, for the same
reasons given earlier (low noise and high
output drive capability). The nominal
(line) output level of the vocoder is
approximately 700 mV; the output
impedance is very low (a few ohms) due
to the negative feedback: the effect of
R134 is cancelled (this resistor is
included for stability and short-circuit
protection).

What's to come?

The power supply circuit, printed
circuit boards and parts lists are still
outstanding. Then, of course, construc-
tional details and calibration procedure.
Quite a lot, all told, but we hope to
squeeze it all in next month.

What else? An article on 'using a
vocoder' is scheduled, and there are
plans for extending the LED indication
- little more than a gimmick in the
present design — so that the vocoder
can be used as a simple spectrum
analyser. A very useful extension.
The further plans are rather more vague,
but we certainly hope to do something
about the voiced/unvoiced detector and
associated noise generator in the not-
too-distant future. One thing is for sure,
we haven't heard the last of vocoders
yet — not by a long chalk! M

Lit.:

liilk funny
onboard

The electronic speech distorter, de-
scribed last month as 'Talk funny?', is
quite a nice little circuit. On second
thoughts, it seemed a pity that the
p.c. board was not available via the
EPS service - but that is easily
remedied.

While 'cleaning up' the layout given as a
suggestion last month, all wiring
between switches, potentiometers etc.
was included on the board. This means
that these components can be wired
direct to the board; the drawing given as
figure 5 in the original article can be
ignored.

One final point: it is advisable to
increase the value of P4 to 47 k (50 k),
as shown in the parts list below. M

R1 - 3k9

R2 * 47 k

R3,R8 * 1 k

R4= 56 k

R5 - 100 k

R6 = 2k2

R7 = 220 O

R9 = 100 k

PI ■ 100 k log

P2 = 470 12 (500 SI) preset

P3 - 4k7 (5 k) preset

P4 = 47 k (50 kl preset

P5 = lOklin

Capacitors:

C1,C7= 100 n
C2.C3 ■ 2p2/10 V
C4 = 1 fi (not elco!)

C5= 15 n
C6= 1 ji/IOV

Semiconductors:

IC1 = XR2206CP
T1 = BC109C, BC549C,
or equ .

D1 = DUS

Switches:

SI .S2.S3.S5 = singlepole
S4 = single-pole, make

Elektor, April and May 1978: Vocoders.
Elektor, December 1979: Vocoders
today.

Elektor, December 1979: Toppreamp.

new bus
board for
mkToproeessors

In most microprocessor systems, the
various subsections are interconnected
via a bus board. The Elektor SC/MP
system is no exception; its bus board
(p.c. board no. EPS 9857) is designed
to take up to three plug-in cards.

The trouble with microprocessor systems
is that they grow ... If several memory
cards are to be used in the SC/MP
system, when including NIBL-E for
example, even two bus boards may not
provide enough space. For this reason, a
new bus board has been designed. As
can be seen, it is the same size as the old
one - but it will fit five cards instead of
three. The two additional connectors
are mid-way between the three original
connector positions. This makes for
easier up-dating of an existing SC/MP
system, certainly from the mechanical
point of view.

At the same time, the wiring between
adjacent bus boards has been simplified.
At the left-hand end of the board,
contact rows a and c have been trans-
posed (note that this means that no
connector should be mounted at this
point!) so that the interconnection
between two bus boards can now be
done with wire links, with very little
danger of short circuits between the a
and c connections. The same applies
when connecting the bus board to the
supply: here, too, the criss-crossing of
wire links is avoided.

The new bus board is supplied under
no. EPS 80024. M

1-32 - elektor January 1980

modern methods of voltage regulatior

modem methods of
wltage

how to use switching
voltage regulators

For optimum performance, it is
desirable that good voltage
regulators are used in modern
electronic systems. Usually, these
regulators are of the conventional
type based on the series pass
transistor. These present few
problems either in design or
manufacture; however, they are
not noted for their efficiency.
This has led to an increasing trend
to the use of switch-mode power
supplies which are far more
efficient. In this article we take a
look at the advantages and
disadvantages of switching
regulators from a practical point
of view.

In conventional power supply circuits
the series pass transistor is operated in a
linear mode at a point someway
between the two extremes, cutoff and
saturation. The transistor acts as a
variable resistor and will dissipate
relatively large amounts of power due to
the voltage dropped across it. Power
dissipation will increase proportionally
with increase in load current or input/
output voltage differential. This is the
major disadvantage of conventional
voltage regulators.

In contrast, the transistor in a switching
power supply circuit is operated only at
cutoff or saturation, it is either full on
or fully off. These are the optimum
working conditions for a transistor. As
we all know, efficiency is the ratio
between output and input power. In
general, a voltage regulator based on the
series pass transistor will seldom reach
an efficiency of 50%, whereas switching
regulators can reach a figure of 75%
(and sometimes even higher). This may
not seem revolutionary, but a closer
look will reveal its true value.

Take a power supply unit designed for a
10 Watt (5 V, 2 A) output. With an
efficiency of 50% the input power
requirement is 20 Watts — twice that of
the output. The difference between the
two, 10W, is dissipated by the circuit.
For the same power unit, but using a

switching regulator with an efficiency of
75%, the dissipation works out to be
3.3 W, an improvement of 300%! This
enables the use of a much smaller
heatsink and a transformer of about two
thirds the size of that in a conventional
system.

The other side of the coin

Compared with conventional regulators,
it must be admitted that the beauties of
the switching system are marred by a
few blemishes. The output ripple is
considerably higher (maybe some tens
of millivolts) and their response to load
surges is more sluggish. The circuitry
can produce hissing or whistling sounds
and, moreover, are prone to the radiation
of RF interference due to the fast
switching process. This, of course, may
have adverse effects on other parts of
the system. The efficiency of switch-
mode power supplies relies to a large
extent on the switching speed and,
therefore, 'fast' transistors (and diodes
for that matter) are necessary. The
selection of these devices has to be
made very carefully with due regard to
parameters. Suitable types are manufac-
tured for this specific purpose but
they are expensive. One other cause of
concern, especially to amateurs, is the
need for an inductor - and a special one

Photograph 1. A slightly unorthodox
presentation of a circuit! Still, it should be

clear enough. The circuit will supply 5 A at

Figure 1. Block schematic diagram for a switching regulator. S represents an electronic switch,
such as a fast switching transistor. Inductor L stores energy as long as S is closed, to release it

when S opens.

How do switching regulators
work?

There are two basic types of switching
power supply currently in use. A block
schematic diagram of the simpler
version is shown in figure 1. The regu-
lator control circuit differs from normal
in that the solid state switch, S, either
switches output power on or off without
consuming energy. The circuit also
features a 'swinging' inductor and a
diode whose function will become clear
further on.

The principle illustrated in figure 2
would approach the ideal. Here, the
mains voltage is directly rectified and
applied to a switching circuit fitted with
a regulator system so as to produce an
alternating current of controlled voltage
at a frequency much higher than the
mains. This regulated voltage is reduced
to the required output level and then
rectified and smoothed. The transformer
in this case is a ferroxcube power
transformer designed to operate at high
chopping frequencies, about 25 kHz for
example, and is much smaller than a
conventional mains transformer with a
similar power rating.

It should be noted at this point that in
practice switching regulators do not use
a symmetrical squarewave, but rather a
controlled waveform normally set at a
6 : 1 duty cycle — this being found
most efficient.

Free extras

Switching systems are capable of
providing more than just stepped down
positive voltages. They can also produce
negative outputs, but a real gain over
conventional systems is their ability to
provide higher output voltages than the
input! This offers the possibility of
deriving a number of positive and
negative outputs from a single source by
using different regulator circuits.
Figures 3, 4 and 5 illustrate the
principle and show that the components
used are identical, only the configuration
differs.

Circuit description
Figures 3a and 3b show the operation of
a step-down voltage regulator. The
unregulated voltage is applied across the
input terminals Uj n and the stabilised
output is produced across the load R(_.
With switch S open, no input current
can flow and the full input voltage
appears across the switch contacts. With
S closed however (figure 3a), the full
input is applied across diode D (non-
conductive) and also appears at the
'high' end of inductor L. Although
initially the full voltage appears across
the inductor, the current through it will

■n methods ot voltage regulator

only increase exponentially to charge
capacitor C2 and raise the output
voltage. As soon as the output voltage
across R[_ and C2 rises to a predeter-
mined level, S is made to re-open and
block the input current (figure 3b). The
current through L will not instantly
drop to zero, since the stored magnetic
energy objects to being left in the
inductor; as shown by the arrows in
figure 3b. thecurrent nowflows through
D (conductive in this direction) continu-
ing to charge C2 and feed R|_-
When the current supplied by L drops
below the current taken by the load, C2
makes up the difference and, by losing
its charge, lowers the output voltage. As
soon as the output drops below its
predetermined limit, the switch will
close and the first cycle will be repeated.
To assist those readers who want more
precise information, the graphs of
figure 3c sketch the various current and
voltage waveforms in the circuit.

A number of manufacturers are now
producing integrated circuits designed
specifically for switch-mode power
supplies.

A typical example is represented in
photograph 1. This 5 volt 5 amp regu-
lator features a Fairchild SH 1605
integrated circuit and only five other
components. A few performance details
are worth noting;

• Input: between 12 and 18 V;

• Maximum output: 5 V, 5 A;

• Minimum output current: 1 A;

• Ripple: 100 mV;

• Efficiency: 70%

Polarity reversal

By a slight change of the configuration
(including reversing the diode), the
regulator circuit can be made to reverse
the polarity of the input voltage.
Figure 4 shows how this is done using a
Texas TL 497 1C.

The operating principle should now be
familiar: magnetic energy is gathered in
inductor L during the period that S is
closed. Once S opens, L generates a
current that charges C2 to a negative
voltage via the diode D. When C2 is
charged to the required amount S closes
and off we go again.

Voltage step-up

The circuits described so far step the
input voltage down or reverse its
polarity. However, as mentioned earlier,
switching regulators are also capable of
.stepping the voltage up. Figure 5
illustrates the principle of operation.
With switch S closed, current can only
flow through the inductor L. When S
opens however, a large voltage is induced
in L which charges up capacitor C2 via
the diode D. Once the voltage across C2
is sufficiently high, the switch is made
to close again.

A practical circuit using the Fairchild
fX A 78S40 is also shown in figure 5. This

4

Figure 4. Reverse-polarity outputs at voltages higher or lower than the input are obtainable with
this configuration.

5

supplies regul.

lutputs at voltages higher than the rectified

jltage regu

elektor january 1980 - 1-35

component, like the previously
mentioned Texas 1C, can be used in all
three basic configurations, namely, step-
down, reversal and step-up.

From a 10 V input, this circuit can
supply 160mA at 25 V with an ef-
ficiency of 79%. The pA 78S40 permits
currents as high as 1.5 A to be handled.
The corresponding output current is
half this amount i.e. 750 mA, It must be
noted however, that the internal diode
(D1) cannot carry more than 300 mA.
The performance can be extended, of
course, by fitting an external diode and
switching transistor.

More technical data

Only a few applications of the integrated
circuits have been discussed. The list of
literature at the end of this article refers
to manufacturers' data sheets, for
readers who wish to design their own
equipment.

A few hints

The construction of a switching regu-
lator is similar, to a certain degree, to
the construction of high frequency
circuits. The steep switching edges and
the consequent high frequency contents
demand considerable care. It is impera-
tive that those conductors which carry
switched currents are as short as
possible. The circuitry must be grounded
at one single point only.

The output capacitor principally deter-
mines the output ripple and tantalum
types are recommended as being
preferable for the high frequencies
involved. As an alternative to a single
large capacitor, the total capacitance
required can be made up from ordinary
electrolytic types connected in parallel.
Some further improvement can be
gained by using capacitors intended for
working voltages of twice the output
voltage.

Unfortunately the majority of common
rectifier diodes are unsuitable. This
same rule applies to the switching
transistor: if not fast enough, the
semiconductors will dissipate excessive
heat, reducing regulator efficiency.

Inductor construction

Most amateur constructors prefer to
steer clear of the problems associated
with coil design and construction.
Although ready-made inductors are now
available, it is sometimes unavoidable
that the hobbyist himself has to calculate
and construct the inductors required.
Fortunately, this is not too difficult;
even so, some suggestions may be

The inductors in question are usually
wound on ferroxcube cores, such as
Siemens N27 or Philips 3C8, which suit
the purpose owing to their low losses at
the high frequencies involved, some 20
to 50 kHz.

6

Figure 6. This graph specifies the safe amount of transferable power as a function of the
(ferroxcube) core size of an inductor.

To simplify calculation, only two design
parameters will be treated, namely, the
permitted core field strength and the
inductance required. The graph of
figure 6 specifies the safe amount of
transferable power as a function of the
inductor core size. For field strengths
beyond a safe value the inductance
decreases with consequent excessive
current through the inductor. This
'unsafe' current will not only rapidly
destroy the switching transistor and the
rectifier diode, but also cause the
output voltage to rise dangerously. For
these reasons, it is important to use
cores of generous section. The graph
indicates that a pot core of 30 mm
diameter can handle as much as 30 W.
Having decided upon the core section,
the desired inductance can now be
obtained. To this end, the A|_ inductance
parameter must be found. This is a
function of the type of magnetic
material, the core section and the
magnetic gap, and is given by the
manufacturer. In most cases it is possible
to settle on parameters that permit a
convenient number of turns, such as 50
or 100, to be wound. A small number
may seem to be more convenient, but
should be discouraged since it leads to
higher inductor losses. Optimum ef-
ficiency is obtained by using the thickest
wire consistent with completely filling
the core.

The resultant inductance is equal to the
A[_ parameter figure expressed in
nanohenries multiplied by the square of
the number of turns. For a core with an>
A(_ of 400 nH and a desired inductance
of 300 pH this works out at

References

Philips: Data Handbook 'soft ferrites'.
Siemens: Datenbuch 'Ferrite'.

Fairchild: Data Sheet SH 1605.

Data Sheet pA 78S40.
Application note SH 1605.
Application note pA 78S40.
Texas Instruments: Application Report
'designing switching voltage regulators
with TL 497'. H

requency ('sampling rate') must be at
wice the highest signal frequency
z in an FM -stereo system),
sr, the circuit becomes less effective as

y 1980 - 1-39

ik output ci

can be as high as 650 jiA

500 it A, and operating

bandwidth is 9 MHz at a bias current of 1 mA.
Noise voltage is typically only 8 nV/v/Hz at

1 kHz, and total harmonic distortion is

typically only 0.4%.

The unit is light and robust and can be fc
stacked and locked with all compartr
held securely in place. The Partfolio 2
available ex-stock from Toolrange and i
example of the large range of elect
production and assembly aids featured i
new Toolrange catalogue.

Toolrange Ltd..

Upton Road.

Reading RG3 4JA.

Tel.: (0734) 29446 or 22245

generators, triangle/sine-wave convertors,

comparators, and audio pre-amplifiers.

The devices are supplied in 16-lead dual-in-
line plastic packages and are available with
operating temperature ranges of — 55° C to
+125°C (CA3280AG) and 0°C to +70°C
(CA3280).

RCA Limited /Solid Si
Sunbury-on - Thames,
Middlesex , England.
Tel.: Sunbury-on-Tham

Dual variable operational
amplifiers

A new family of dual variable operation
amplifiers, the CA3280G Series from RCA
Solid State, combines two operational trans-
conductance amplifiers in a single package
and offers significant performance advantages
in parameters such as output current, ampli-
fier matching, input offset voltage, band-
width, slew rate, noise and reliability. The
two amplifiers are independent, differential-

oscillators assure a stable reference signal and

accurate timing. Typical accuracy is i 0.01%

± 1 count (! 0.005% ± 1 count for the phase-
locked loop unit).

The range of complementary digital velocity
transducer includes opto and magnetic pick-
off types and infra-red units. Many are avail-
able in bolt-t

Orbit Controls Limited,
Lansdown Industrial Estate.
Gloucester Road.

Cheltanham, Gloucs GL51 8PL.
Tel.: 0242 26608

Two bubble memory development
kits

To enable engineers to learn how to use
magnetic bubble memories, GEC Semiconduc-
tors have just announced two Intel kits. The
first is Bubble Memory Prototype Kit type
BPK-71 and the second is type IMB-100
1 Megabit Bubble Memory Development

BPK-71 is the simplest kit. It basically com-
prises the Intel type 7110 1 Megabit bubble
memory with the standard drive semiconduc-
tors, a current pulse generator and a for-
matter/sense amplifier. All that is needed to
make this a complete basic memory system
j-based controller and a power

; supplier

trailer. The

ncludes appli

e necessary mpi

lplete documen-
and a complete

the magnetic bubble
memory system to interface directly to a
Multibus and 8080/8085 pPs.

The second 'kit' is a full development board
(IMB-100), containing not only the basic
memory and its drive and formatting/sense
circuits, but also a complete controller to
provide the necessary I/O faci

d of tx

Speed measurement system

Orbit Controls is marketing a complete speed
measuring system. It comprises 11 types of
tachometer and over 30 different kinds of
digital signal transducer.

The tachometers include both 4- and 6-digit

separate standard components.
The IMB-100 is complete (e>
power supply) on a printed circt

100 MHz oscilloscope

A new high-performance oscilloscope from
Gould Instruments Division, the Gould
Advance OS3600, is the company’s first
oscilloscope to break the 100 MHz bandwidth
barrier. Featuring a new compact case design,

exacting working environments, particularly
in fieldservice applications. The new instru-
ment is also available with an optional digital
measuring unit, the DM3010, which makes it
a complete measuring system for a wide
range of analogue and digital applications.
The OS3600 is a dual-channel instrument
with a maximum sensitivity of 2 mV/cm.
The full 100 MHz bandwidth is available
from 5 mV/cm, and the 2 mV/cm sensitivity
is maintained up to 85 MHz. A high-writing-
speed cathode-ray tube with 16 kV overall
accelerating potential gives bright, clear traces
at sweep speeds down to 5 ns/cm, and signals

trigger hold-off control allows the trigger to
be disabled by up to one trace length to
facilitate triggering of a complex waveform
that has no separate sync, or trigger pulse.
In addition to the normal range of vertical
display modes, a pushbutton-actuated trigger-
view facility is incorporated. This allows
continuous display of the trigger signal
simultaneously with one or both of the input
channels, and with an external trigger signal
it effectively provides a third channel with a
sensitivity of either 50 mV/cm or 0.5 V/cm.
Other standard features on the oscilloscope
include a calibrated delay vernier control,
‘uncal’ warning lights on all variable gain/sweep
controls, a trace location facility, an internal
illuminated graticule, versatile X-Y display

tial amplitude measurements.

The optional DM3010 digital measuring unit
consists of a 3'/j-digit multimeter which can
be used completely independently of the
oscilloscope for measuring d.c. voltage,
current and resistance. It can also be used in
conjunction with the OS3600 for accurate
measurements of signal amplitude, time
interval and frequency. Timing relationships

The Gould Advance OS3600 carries the
Gould 2-year guarantee.

Could Instruments Division,

Roebuck Road, Hainauit. Essex.

Telephone: 01-500 WOO

Push-button thermocouple
selector unit

The Model ’3508’ is a new push-button
operated thermocouple selector unit from
Comark housed in a miniature DIN-size panel
mounting case. Its use increases the flexibility
of the company's digital thermometers by
enabling one thermometer to be switched,
quickly and reliably, between up to seven
thermocouple sensors.

Two-wire inputs are connected to terminal
strips on the rear of the unit. These terminals
have been carefully positioned to minimise
errors which can be caused by draughts on the

designed to be compatible with Type K
(Ni-Cr/Ni-AI) thermocouples. It is supplied

Vacuum fluorescent display digital
frequency meter

lenuy meiei ......

A simple clip-on case is available, it reqi
v LCD frequency meter offering the to convert the unit for bench top use.
ages of the wide enviromentai capabili- ava j| a ble is a small, double-decked’ l
a VF (vacuum fluorescent! display is case to take the selector unit and a th<

elektor january 1980 - 1-41

The unit employs the very latest dual gate
MOSFETs, and provides an overall gain of
40 dB with a 2-3 dB noise figure. A pin diode
AGC attenuator is used in an internal AGC

IF derived AGC is available via the control
gates of the FETs.

A fully buffered local oscillator output is
provided, together with an IF preamp using
an FET. 6 varactor tuned stages provide the
exceptional RF selectivity necessary to make
best use of the.high gain.

Brentwood, Essex.

Telephone: 1 0277 1 227050.

(1396 M)

Fast A/D converters

A new 16 and 14 bit A/D converter series
from DMC provides an exceptional corn-

stability. First designed for the toughest
analytical instrumentation requirements,
these converters now make 16 and 14 bit
performance practical for a wide variety of
applications.

Model 2816 converts 16 bits in 100 micro-
seconds, and Model 2814 converts 14 bits in
50 microseconds. Maximum linearity error
is ± 0.0015% of full scale for the 2816, and
t 0.003% of full scale for the 2814. Both

watt, and PSRR is 0.002%/%.
convenience. There are four full-scale standard

are provided for gain and offset. They can be
easily recalibrated in the field, or set to cancel

(1383 M)

Offer valid until 31 st January 1980.

COMPLETE CONSTRUCTIONAL DETAILS OF THE ELEKTOR
FORMANT SYNTHESISER

The book the synthesiser world has been waiting for!

ELEKTOR now release the FORMANT BOOK with a FREE CASSETTE.

ORDER DIRECT FROM ELEKTOR, 10 LONGPORT CANTERBURY, KENT CT1 1PE.