flatten wm ir IS\/inn rnom uuifh an onnalicor Audralia S1.50* Austria S. 36 Denmark Kr. 10 Germany DM. 4.20 New Zealand S1.50 Sweden Kr. 14 • rstom mended Belgium F. 59 France F.8 Netherlands DFL. 3 50 Norway Kr 10 Switzerland F.4.40 An attractive mains alarm dock with radio switching function and battery back up! Complete kit with case only £15.92 (ind. VAT & p& p) MA1023 module only £8.42 (ind. VAT). All mail to:- RO. Box 3, Rayleigh, Essex SS6 8LR. Telephone: Southend (0702) 554155. Shop: 284 London Road, Westdiff-on-Sea, Essex. (Closed on Monday). Telephone: Southend (0702) 554000. ELECTRONIC SUPPLIES LTD September 1979 volume 5 number 9 selektor A display of 16 lines of 64 random characters may look impressive, but in practice the corre- sponding memory capacity is somewhat restricting. Even a simple BASIC program will require more space. For this reason, the page extention for Elekterminal should prove a useful addition. using an equaliser Although there are mar system and/or listening an extremely useful tr Unfortunately, however : task, namely monitor output page extension for Elekterminal With the aid of the extension board descr capacity of the Elekterminal can be exp (each of 16 lines x 64 characters). one-nil for audio The advent of digital audio has prises: digital designers discovers the very limit of their capabilitie formance standards commonly s equipment; analog designers, on prised to discover that digital equip In this article, both of these 'sui can digital audio work so well, a Use of a parametric equaliser allows the fre- quency response of a domestic hi-fi setup to be tailored to a degree previously only attainable get in recording studios. One section in particular parametric equaliser should prove of interest A combination or state-vari! to a great many readers: to ”^ ll de^ib»d'in > th!r the parametric tone con- advantages over the more cc trols, with adjustable turnover frequencies. a pplikator rial analog 'Flatten your living room audio analyser with an equaliser'. Great, Wi, ^ ut an a . cc r u . ,a '' but how? The bulb Of rore harm than gi this issue is dedicated to ^CdjJ^haiffand/oT descriptions of the pensable piece of eq necessary hardware, and how to use it in practice. T «p : on ejektor Self-oscillating PWM amplifier. market advertisers index UK-30 f Into the bio-electronic age The fight for dominance in information technology is little appreciated by politicians or understood by the man in the street. But Europe should have no illusions about what is at stake, says Douglas Stevenson, vice president ITT and group executive, components and semiconductors. The age of biocom- ponents. where man can operate machine by thought alone, is very near . . . Fifty years ago was not a good time for forecasters. In early 1929 most of them were still optimistic about the New York stock market. By October the crash had come. The world was not to emerge from the Depression until World War II. Had I been making a forecast in 1929 about the development of electronic components, I too would have been wrong. Most of the basic circuit el- ements like resistors, capacitors, induc- tors, the electron tube, the cathode ray tube were known. In no way, though, would I have forecast the development of semiconductor technology. Yet the transistor was less than 20 years away! Since then the pace of development has speeded up. Things change so rapidly that the period we can forecast with any degree of certainty gets shorter. Nevertheless, certain trends can be projected and others predicted. For example, the number of people employ- ed in the electronic components industry will continue to shrink. At the same time, the fewer people will be devel- oping and producing a greater variety and a greater volume of components. It is important to distinguish between volume and value. In real terms the total industry will grow in value over the next decade by no more than five per cent per annum. In physical terms though the number of functions that will be per- formed will increase in the order of 20 to 30 per cent a year. Electronics will play a much greater part in our lives. Towards a leisured society Look at domestic applications. Most labour-saving devices in the home are based on a 19th century development, the electric motor. It has been the major technical factor in removing the domestic servant from our society in such great numbers. Today, households that would never contemplate the employment of a servant have several labour-saving devices, which are no more than con- cealed fractional horsepower motors: vacuum cleaners, washing machines, driers, polishers, mixers, fans, pumps, lawn mowers, power tools. Essentially these all lighten physical effort. The impact of electronics in the home has so far largely been in making life more pleasurable. It has been an entertainment medium — appearing in the form of radio, television, hi-fi sets, recorders and, more recently, video A recent development, the pocket calculator, is a pointer of things to come. The enormous growth area is in com- puting: all the processes involved in the storage, handling, transmission and display of information. The number and power of these processes will increase in the home, and more so, in industry and commerce. Electronics will make our lives easier by taking over the job of remembering many simple but necessary things, doing what has to be done at the right time. An example not far away is the pre- programming of family television viewing. It is possible to produce an edition of the Radio Times with bar- encoded programme references. A week beforehand, say, the viewer can select the programmes he wants to see and, with a light pen, scan the encoded refer- ences. These will be stored by the TV set. which will switch on and off auto- matically when the time comes. That is one simple application that within the next decade we shall come to accept as the normal way of organising part of our relaxion. Our present methods will seem quaint. Another application already growing is in security systems. These range from individual programmed locks to com- plete surveillance, checking and alarm systems. elektor September 1979 - 9-01 The future though is not so much in individual applications as in integrated systems. There will be home automation systems compact enough to go under the stairs or take up little room in a cupboard. They will do everything from controlling lighting, central heating and other appliances like cookers and washing machines, to providing meter readings for electricity, gas and water. We can already see in Prestel the home linked to the outside world by a combi- nation of the telephone and the TV set. Without stepping outside the house one can consult a computer-based data bank for up-to-the minute facts and figures on all sorts of services. Linking a home automation system with the telephone and the TV set, we shall have the essential elements of recording, control, transmission and display of information. Many of our activities that now involve going out in all weathers, finding a place to park, battling with the crowds, even staying away from home, will in future be accomplished from our own armchairs. These systems will be based upon the very low-cost computing power that the microprocessor offers. This will give tremendous impetus to the development of automatic systems. The technology is available now to take us through the next 10 to 15 years. The question is one of application. An energy -dependent world Developments like this are not simply a move towards a more leisured or lazier society, depending upon your point of view. They are an economic necessity. Every day it becomes clearer that the turning point of the 1970s was the sharp increase in oil prices at the end of 1973. That brought home to us the value of energy, the fact that the world's resources were finite and had to be conserved. People had to adjust to the fact that the high growth rates of the 1960s were no longer possible. Electronics has two contributions to make. Apart from saving power, elec- tronics also make possible a completely new approach to a problem. There is a world of difference between physical communications and telecommuni- cations. It is much easier and cheaper to communicate information than to trans- port people, be they suburban com- muters or supersonic day-trippers to New York. As the cost of labour rises and, in real terms, telecommunications charges fall, so it will be cheaper to have home meter readings impulsed over a line rather than taken by a human reader. The postman could disappear in favour of a home facsimile. The energy factor cannot be under- estimated, At the end of the century there will be an energy gap. which could lead to international and social insta- bility, and a resulting cataclismic nuclear war would not be impossible. In time, the gap has to be filled by safe fusion energy. There might be a period between the two, 10 years or so, when the future of the world hangs in the balance. The new industrial revolution What then are the relative futures for discrete components and integrated circuits? So much has been said recently about integrated circuits, micropro- cessors in particular, that discrete components would seem to have a distinctly limited future. In real terms they will continue to grow up until about 1985. Decline — but not in power elements — should then take place. It will be a slow process. By the turn of the century I foresee demand for discrete components in physical terms being about 75 per cent of what it is today but, although volume will be down, value will be up. The components industry will continue to develop along lines of greater integration in a broad sense of the term. We have seen com- ponents become circuits, become a system, become a system that is pro- grammable. We have to think more and more in terms of sub-systems. The major component manufacturers have already entered the sub-system fields and even gone into complete systems. Examples are the complete driver units for ground-to-air missiles, which are complex quasi-systems in their own right. This trend will continue wherever the basic technology is inter- dependent with the function of the total system. You cannot separate the Any component that provides an inter- face with a human being, or acts as a power unit, has an indefinite future. Into these catagories come items like push button switches, displays, power units, motors — be they linear or rotating. Human beings are not going to diminish in proportion to microelec- tronics. They have to receive and communicate information. Similarly, to interact with the real world, miniature devices will still have to have their powers amplified and directed. On the other hand discrete passive components will decline. Included in these are the discrete resistor, inductors and the low capacity capacitor. Many of the functions performed by these passive devices can now be simulated cheaply by active elements in a semi- conductor. To survive in circumstances of rapidly developing technology, changing product mixes ; shifting price structures. | manufoi ?' . ■ are going to have to get their forecasts right, if not the first time then very quickly the second. The major process of survival of the fittest — and in the fight no manufacturer can assume he is a natural survivor - should have taken place by the early 1 990s. The Japanese will make every attempt to get the same control of the industrial and professional sectors of electronics that they have achieved in consumer electronics worldwide. That will be the major political factor in the industry over the next 10 to 15 years. It is different in kind from a commercial battle over the manufacture of items like motorcycles and cars or the con- struction of supertankers. It is nothing less than a fight for dominance in the whole area of information technology, which is the key to everything else. The aptember Japanese understand very well the interdependent triangle of the future, survival and technology/political power - and will act accordingly. The Western World has to have no illusions about what is at stake. This, I believe, is understood at indus- trial senior management level but not fully appreciated at top political level and hardly at all by the man in the street. Manufacturing capability in Isi and the ability to apply low-cost com- puting power means nothing less than a new industrial revolution. It is the absolute cutting edge of technology in the world today, whether it's going to end up with the control of chemical plants or sophisticated toys. On a practical working level, the industry will concentrate into massive units developing and manufacturing com- ponents. In 50 years' time, some 30 per cent of these units will be produced in Japan, some 40 per cent in the US, and selected areas of Western Europe will account for another 30 per cent. The secondary technologies will in- creasingly go off-shore. The totally integrated specialist In the fight for survival, the vulnerable companies will be medium-sized: those that have neither the resources and the mass markets of the large nor the skill and flexibility of the specialist manufacturer. There will be no place for, say the specialist manufacturer of a high-quality microwave or optical device turning over in current values up to $20 million a year. To survive he will have to have an edge with his techno- logy and do superbly well at it. If I were looking for a secure long-term pension, I would not invest in a manu- facturer turning over less than $ 200 million in a product spread. The future profile of the distribution of company sizes will be double- humped, with some level and very uneven ground in the $ 20 — $ 200 million area. There, it will be very difficult to support the R + D, the capital invest- ment, the marketing. As a simple example, a set of tools just to make a colour TV tube, which can be regarded as a medium-technology product, currently costs $6 million. Twenty years ago it was possible to survive by making 50,000 tubes a year. Today a break-even figure is about two million. For any hope of industrial survival, we shall have to maintain technology in depth. That means deciding which technologies we have to be in. These must be the primary technologies. We have to maintain at least parity with our competitors in these. In other technolo- gies, those of secondary importance, it will be necessary to maintain a capability to bring into being, if need be, a reserve of industrial muscle. The EEC might declare to the Japanese that it is not going into the production of certain devices, but that it will maintain the capability so that Europe is not held to ransom. A possible limitation on our ability to do so will be the scarcity of truly creative physicists and engineers. We may well lack manpower of the right calibre and in the right numbers. Just as there will be totally integrated systems, so there will be totally integrated engineers and physicists. By the turn of the century, individuals will need to possess integrated disciplines to be able to design systems. The equipment will demand a human being who correlates 100 per cent with it. In the production of electronic com- ponents we shall see the elimination of the man on the shop floor — except for maintenance purposes. Within 20 years, no unskilled people will be used in the electronics industry. With totally con- trolled environments there will not even be a need for people to sweep the floor. On the other hand, there will be a heavy capital investment in machinery, which will be making products with a short life cycle. Fault diagnosis will be done by computers. Again, systems will be integrated and of such a complexity that only a few large organisations will be able to afford them. A profound change in work Output will be in such volumes that it will have to have assured markets. Producers will lock into their customers. One will adopt the other. Small com- panies will have to interface with their customers on a continuous basis. Once again the word is integration. Technological developments of this kind and magnitude are going to mean profound changes in society. The pattern of work will change. A great reduction of working hours is unlikely. We shall not see the 30 hour week in the next five decades. A component we do not yet have but would dearly love to develop is one that can convert sunlight, not into power as a solar cell does, but into chemical energy as do organic living species. This involves producing artificial membranes in the laboratory con- taining compounds which perform specific or even analogue functions. In ITT work is already being done on membranes that can separate negative and positive charges. Thus, by distin- guishing ions, these membranes could have very practical applications as storage elements or in pollution control. They could carry out simple tasks like sensing and filtering anything from a toxic atmosphere to a very low concen- tration of impurities. They could do this with an efficiency and accuracy that is beyond the scope of current physical methods. Superclean environ- ments are possible. Developments of this kind are only the start of translating biological functions into other useful energies or actions. This enormous area will be the next great stage in the evolution of com- ponents. The sort of thing I have in mind is photosynthesis on a large scale, the equivalent of a plant taking in sunlight and moisture — and growing. Another example of the efficient storage and transmission of energy. Electronics will move into bio-engin- eering, bio-physics and bio-chemistry. We accept as an everyday fact that we can synthesise the human voice. So why not food for a hungry world? Going even further, why not connect a human being directly to a computing system? I do not believe it is beyond the bounds of possibility that the output of a human brain can be directly fed into a computer. What an amplification of mental power! And without going through any software, A considerable amount of mathematical analysis has already been done on the brain. The missing link is the bio-component or subsystem. This would take electronics into neuro- logy. Such an advance could speed up developments in an undreamed of way. At present we are obliged to use soft- ware, a stage that may occupy many man-years in translating a sequence of precise, detailed instructions acceptable to an unthinking machine. There is a shortage of software people. Hence the implementation of projects gets delayed. It is an enormous problem. If we could have a direct human connection to the computer how much simpler life would be. We already have artificial limbs and fingers actuated by brain signals. With an organic interface a person can place his fingers on a sensor and pass 'thought' signals to instruct equipment. By the end of this century, we shall see the first bio-electronic components and subsystems performing, at the very least, basic functions like separation and storage. Direct connection of man and machine belongs to the 21st century. 9-04 - ele The great advantage of an equaliser is that, unlike conventional bass and treble tone controls, which can provide only a fairly limited amount of boost or cut at the extremes of the audio spectrum, it is possible to iron out (equalise) peaks or dips in a response over the entire range of audio frequencies. Not only that, but with a parametric equaliser, the centre frequency, Q and gain of the equaliser filters can all be tailored to exactly compensate for non-linearities in the response of any given system. Although the use of equalisers was originally limited to professional sound recording studios, their undoubted benefits have led to an increasing number of amateur applications: dedicated hi-fi enthusiasts, having lavished considerable attention and expense on cartridges, pick-up arms, turntables, amplifiers and loudspeakers, are now resorting to equalisers to 'upgrade' the last link in the audio using an equaliser chain, namely the listening room. Unfortunately, however, many amateurs fail to make the most of the facilities offered by a sophisticated parametric equaliser, and simply end up using it as a sort of 'super-duper' tone control, twiddling the knobs to get a bit more bass here, less treble there and so on. This article is therefore intended to provide a few insights on how to achieve effective room equalisation, whether it be for domestic or PA-system appli- cations. Although there are many different types of equaliser, they all perform the same basic task, namely the correction of deficiencies in the fre- quency response of one's speaker system and/or listening environment. As such they represent an extremely useful tool in the quest for 'perfect' hi-fi. Unfortunately, however, equalisers are all to often misused, and in extreme cases actually do more harm than good. The following article takes a look at the various types of application for which equalisers are most suited, and also explains how to get the best out of this versatile instrument. Equalising your living room In recent years the subject of room equalisation has become something of a fad. Various audio design consultants and well-known manufacturers of audio equipment have conducted extensive research into the response of domestic listening environments. Bruel and Kjaer, for example, offer a comprehensive measurement and equalisation system for listening rooms, whilst Philips loud- speakers are specially designed to compensate for the deficiencies of the 'average living room'. The subject of room equalisation, with particular reference to the effect of the placement of loudspeakers, has been discussed in a spate of recent articles, and numerous hobbyist magazines have produced designs for (graphic) equalisers. There is no doubt that people are now generally aware of the effect of the shape and contents of the listening room on the reproduction of the audio signal. That the room has considerable effect is hardly surprising, especially when one considers how much care and attention is paid to the internal construction of loudspeakers (bracing ribs, damping materials, air-tight seals etc.): in a sense, with different loudspeaker placings, in the room's frequency response, the listening room is simply a giant swop the furniture around etc. Although Assuming, for example, that the room loudspeaker cabinet, in which the whether the living room will remain in question has the response shown in listener sits. However, as a rule little or liveable-in is another question! figure 2a. Using an equaliser the response nothing is done to improve the response A simpler solution to the problem of of the audio system can be tailored to j of the room. Of course it is possible to 'upgrading' your living room is to look like that shown in figure 2b, i.e. take such steps as to change the curtains, employ an equaliser, which will the inverse of the room's response, with I fit wall-to-wall carpeting, experiment compensate for the inherent deficiencies peaks at 1600 Hz and 4 kHz, dips at 50 f (HZ) 9936 2 Figure 2. An example of how, in principle, it is possible to obtain a uniform frequency response with the aid of an equaliser. The irregular response of figure (a) is smoothed out by setting up the inverse response (shown in figure (bl) on the equaliser filters. The result (figure (c)), in theory at least, is the desired perfect reproduction. linssisi Bg HK ' iiisifia "ItlSB Si _ iitiwmi 1 .iiiiUil !S M, !S! I'anOHM ra^gnsss iiiKwnS !iiS' BUS H r September 1979 — 9-17 .tro • lb #*oo 1C 2a j^f" f st 1 fehi A monitor output is useful for con- necting an equaliser, as suggested else- where in this issue. Fortunately, it is a fairly simple matter to add this facility to an existing amplifier. In most cases, only minor surgery is required: the signal path must be cut at a suitable point in the (pre-)amplifier. The top of the volume control is usually as good a place as any (figure la). The two loose ends, A and B, can be con- nected to a DIN socket as shown in figure 1b: pins 1 and 4 are used for recording (preamplifier output), pins 3 and 5 for playback (via the volume control) and pin 2 is connected to supply common. For a mono connec- tion, pins 1, 2 and 3 are normally used; pin 5 is left floating, and pin 4 may or may not be connect to pin 1 . The point at which the signal path is interrupted should have a reasonable nominal signal level (100 mV ... 1 V); furthermore, no DC should be present at this point. It is usually a good idea to add a 'monitor' switch, as shown in monitor output 2b 2c figure 1c, so that the original connec- tions can be restored if no equaliser or tape recorder is connected. Connections to the equaliser The equaliser can be connected as shown in figure 2a, 2b or 2c. In fig- ure 2a. the monitor connection is used. In some cases, a series resistor or voltage divider may be present in the 'A' con- nection (preamplifier output); if so, this should be removed. With switch S2 in the 'monitor' position, the equaliser is in circuit; in the other position ('source') the equaliser is bypassed. The disadvantage of this circuit is that the monitor connection on the amplifier is always in use: it is no longer available for connecting a tape recorder. The solution is shown in figure 2b: add a further monitor connection at the input of the equaliser. The original monitor switch, S2, is always set to the 'monitor' position and S3 is used as the monitor switch for the tape recorder. The equaliser is switched in or out of circuit by means of S4. Finally, some commercial amplifiers (particularly those intended for PA work) have a connection at the back marked 'PRE OUT/MAIN IN'. The equaliser can be included at this point, as shown in figure 2c. M ; - r ‘ . . . i >£>#J ■ ■ N’SftTft.. * J>.?Jfii!i'?->fl(U8C>C.>£>M"S8<85?3>*>*!.«?3?n;s*?C55';4 /3?ye«aS3^5»<«8i.>D>MM-iY>«i>t)M,«,!-s « .« n ;. s; .’?S8IU*-.S’*t8««VMvO>BIS<5)l;*t>»>M t C .?*»0«‘A£>n is*«s E?.7J0e8#K).>8«'tA- *< ; i'-s’OA I->4»P# 8 « <- t F»«i •• : extension fin* elekterminal A memory capacity of 16 page lines is in practice somewhat restricting. Even a simple BASIC program will generally ' require additional lines. For this reason expansion of the video memory is highly desirable. To increase the number of pages in the ; VDU's memory it is first necessary to provide a control circuit which will select the correct page, bearing in mind that the 16 lines displayed on the screen may be composed of sections of two success- ive pages. To this end a page counter is required, which selects the desired page by enabling the appropriate memory 1C. * The basic principle is illustrated in the block diagram of figure 1. Pages 1, 2 and 3 are accommodated on the exten- sion board, whilst page 0 is housed on the Elekterminal board. The page counter is in turn controlled by the CRTC of the Elekterminal and by the up and down keys of the ASCII key- ■ To be able to manipulate several memory pages satisfactorily the following 1 functions are necessary: - the page counter must be capable of counting up and down. - the memory must 'wrap round', i.e. I upon reaching the end of the last page, the start of the first page must reappear on the screen. - conversely, when 'counting down', ! the last page must follow the first. I - it should be possible to reproduce sections of two successive pages on the screen. The above facilities can be summarised • by representing the memory as a drum, on which the four pages are spread out. i The drum can be revolved in either direction, and any 16 successive lines can be displayed on the screen. With the aid of the extension board described here, the memory g counter capacity of the Elekterminal can be expanded to 4 pages (each of _. 8 . , , , , , . The operation of the page counter can 16 lines x 64 characters). Interconnecting the two boards is not a best be explained with reference to the problem, since they can be mated quite simply by means of connectors. CRTC in the Elekterminal circuit. The latter contains a page-end comparator which provides two output signals, RP and RS. The RS output is used to indicate the transition somewhere in mid-screen from one page to another. If a complete page is on the screen, the RS output is high. If however sections of two pages are on the screen, then the page at the bottom of the screen is taken as the 'actual page'. During this portion of the page the RS output is high, whilst during the portion of the previous page it is low. For example, if lines 7 ... 16 of page 2 and lines 1 ... 6 of page 3 are on the screen, then the RS output is low for the first 10 lines and high for the last 6 lines. The RP output provides a '0' pulse when a page boundary is exceeded at the bottom of the screen. This pulse is only generated if pressing the LF (line feed) or ESC (escape) key will result in the transition to the next page. Together, the RS and RP signals are used to control the page counter. page' aptember 1979 — 9-19 Circuit As can be seen from figure 2, the circuit of the page counter is quite straightfor- ward, and consists of an up-down | counter (IC1), a 4-bit full adder (IC2), and a 2-to-4 line decoder (IC3). The three additional pages of memory are formed by 18 RAM’s, type 2102A4 I (figure 3). It is also possible to use low power memories for this application (type 2102AL4), which would result in a saving of roughly 30% in current con- sumption. The extension board also includes an anti-bounce circuit (round 1 N3 . . . N6) for the page-up and page- down keys on the ASCII keyboard, which could not be used until now. ; Their purpose is to enable the user to ' 'turn over' a complete page of memory at one go, i.e. scroll a full 16 lines up or down, regardless of whether it is one complete page or formed by sections of two successive pages. When the RP output of the CRTC goes low, or the page-up key is pressed, the up-down counter is incremented by 1; pressing the page-down key causes the counter to decrement by 1. The full adder then determines the binary sum of the counter contents and the RS signal. Depending upon the result, the decoder takes the corresponding output low, thereby enabling the appropriate | memory 1C. When a complete page is on screen the RS output is high, with the result that the page numbers are all increased by 1 . The page numbering recognised by the page counter is shown in the block diagram of figure 1. As already mentioned, page 0 is situated on the Elekterminal board. If sections of two successive pages are on-screen, the RS output will be low for the first page, I and high for the second, so that the counter will automatically 'turn the page' at the correct point. For a descrip- tion of the operation of the page memories the reader is referred to the article on the Elekterminal (Elektor44, December 1978). Printed circuit board The printed circuit board for the extension to page memory (see figure 4) is provided with two connectors thereby facilitating interconnection with the terminal board. The 26-way connector should be soldered to the underside of the extension board, so that it mates with the connector socket on the terminal board. A number of connec- I tions however are not made via this 1 connector. These are BO . . . B4, B6 and the connections to the page-up and page-down keys. Provision is made for an 8-way connector, the pins of which are connected to the corresponding pins of the second connector on the terminal board. Of course the connector is not essential, it is equally possible to make these connections simply using ribbon DATA Figure 1. Block diagrem of the extension to page memory. Page 0 is accommodated on the Elekterminal board. N1.N2 “ IC4 = 1/2 74LS00 Figure 2. Circuit diagram of the page counter and anti-bounce logic. The numbering of the input and output connections corresponds to that used on the Elekterminal board. For the connections to the page-up and page-down keys there are two possi- bilities: either the key contacts can be connected directly to the extension board, or they can be routed via the extension board. If connectors are being used, then the latter option is the simplest. A small modification to the terminal board is also required before the memory extension is complete, namely the wire link between CE of IC3 and ground (see figure 5) should be removed. Scrolling Once the memory is provided with extra pages, scrolling the text up a line at a time will normally be done with the aid of the ESC key. If the LF key is used, the text will scroll up, but the following line will be blanked, i.e. the line will appear vacant whilst the contents of the line are also erased from page memory. As mentioned, with the aid of the page- up and page-down keys the text can be scrolled in either direction a page at a time. Upon reaching the end of page memory (64 lines), the page counter wraps round to the start of the first page. Power supply If normal memories are used, the current consumption of the extension circuit is roughly 600 mA. By employing low power memories this figure can be reduced to approximately 400 mA. It may prove necessary in some cases to uprate the Elekterminal power supply. Readers are referred to the article on the SC/MP power supply contained in Elektor 36, March 1978. M Digital systems have one major advantage over their analog counterparts: they can tolerate extremely high interference levels without loss of information. Rapid advances in digital technology in recent years is forcing designers in such traditionally analog areas as tape recording, long-line transmission and reverberation to take a long, hard look at their digital competitors. The advent of digital audio has produced quite a few surprises on both sides of the fence: digital designers discovered that only by pushing to the very limit of their capabilities could they meet the performance standards commonly set by conventional analog equipment; analog designers, on the other hand, were surprised to discover that digital equipment could sound so good. In this article, both of these 'surprises' are examined. How can digital audio work so well, and why is it so difficult to get it to work in practice? storage. This is the main reason why digital audio is so interesting! The main questions regarding digital audio will by now be obvious: how can digital technology be used for audio applications; how good can the quality be, in theory and practice; and what is it going to cost? The answers depend to a large extent on one essential unit: the analog-to-digital converter. Analog-to-digital conversion A digital audio system contains five distinct sections: an analog input circuit, an analog-to-digital converter, digital processing and/or storage units, a digital to-analog converter and an analog output circuit (see figure 1). No matter what techniques are employed in the two conversion sections, their basic function is the same: 'translating' an analog (e.g. audio) signal into an equiv- alent digital signal and vice versa. The 'equivalent digital signal' consists of a rapid succession of binary numbers (or 'words' as they are commonly called — for no apparent reason); each ‘word’ represents one particular voltage level at one particular moment in time. 'One voltage level', 'one moment in time' . . . virtually all the major differ- ences between analog and digital audio There are fundamental differences be- tween digital and analog systems. A very basic analog circuit (such as a single- transistor emitter follower) can easily handle a signal (voltage) that varies continuously, taking on any value between some maximum and some minimum. It will introduce very little distortion (less than 0.1%) and a small amount of noise will be added. However, it is virtually impossible to eliminate the added noise and, as more and more of these stages are connected in series, the signal quality will progressively worsen. A digital system, on the other hand, would require something like a 12- or even 16-bit databus to pass the same information. However, the quality of the signal can then remain the same no matter how many stages are to be connected in series. A digital system must be quite sophisticated if it is to achieve the same quality as an analog system, but it then has the advantage that no further reduction in quality need result from signal processing and digital audio: the whats, the whys and the ho«/c one-nil for audio 1979 -9-23 In s digital audio system, the analog input stage (A) is followed by an analog-to-digital converter (A -*DI. The signal can now be digitally (D) processed, transmitted or stored. A digital-to-analog (D -»A) converter and analog output stage complete the chain. stem from these two. Let us first consider voltage levels. An analog signal varies between some maximum and some minimum level, and can take any value between these two extremes. In theory, therefore, an infinite number of different voltage levels are poss- ible: 0.12345 V is slightly less than 0.12346 V, and 0.123455 V is midway between these two levels ... In practice, however, there is a limit to the accuracy with which an analog voltage can usefully be defined. This limit is a result of an unavoidable analog phenomenon: noise. Assume, for instance, that the analog noise level is in the order of 0.0001 V (i.e. 0.1 mV). The difference between the three voltages given above is then 'masked' by the noise: a 'true' signal level of 0.1234000 ... V could be shifted to any level between, say. 0.1233 V and 0.1235 V - depending on the level of the noise signal at the particular moment in time that we are interested in. For the same reason, an output signal of exactly 0.1234 V can be obtained for any input level in the range from 0.1233 V to 0.1235 V. One output level 'represents' a range of possible input levels. A similar effect exists in digital systems - but for an entirely different reason. As stated earlier, each 'word' in a digital system represents one particular voltage level. In a given system, the number of possible 'words' is limited: using 12 bits, say, the binary numbers range from 000000000 000 to 111 111 111 111. In this case, 2 12 or 4096 different num- bers are possible. Therefore, only 4096 different voltage levels can be rep- resented — out of the infinite number of possible levels in any analog system! The only solution is to divide the analog signal range into the same number of smaller regions as there are digital 'words’. For instance, if analog voltages between -2 and +2 V are to be pro- cessed in a 12-bit digital audio system, the range could be divided into steps of approximately 1 mV. Any input voltage between, say, 1 .022 V and 1 .023 V would then be represented by the binary number 001 111 111 110. This process is called quantization, and the inaccuracy that it involves (a given analog signal level can only be rep- resented to within, say, ± 0.5 mV) is known as 'quantization error'. It is also referred to as 'quantization noise', since the effect is similar in many ways to analog noise. However, in some cases it may sound much worse . . . The second phrase to be discussed is 'one particular moment in time'. An analog signal varies continuously: if it is 1.000 ... V at one particular moment, it may be found to have dropped or increased significantly a fraction of a second later. Fortunately, however, it can be shown that if the signal is 'sampled' at a sufficiently high rate, then no information will be lost. In other words, if the signal level is measured at sufficiently short intervals it is possible to reconstruct the original signal exactly from these measured Theoretically, the 'sampling frequency' must be at least twice the highest frequency present in the signal that is to be sampled. For instance, if a system is intended to pass audio signals over the full range from 20 Hz to 20 kHz, the sampling frequency must be at least 40 kHz. In practice a higher sampling frequency is normally required, to avoid all sorts of nasty effects — as will be discussed further on. A block diagram The basic principles of a digital audio system, as described above, can now be summarized in a block diagram (figure 2). The incoming (analog) audio signal must first be passed through a low-pass filter. to remove any signal components at frequencies higher than half the sampling frequency. The next step is to sample the analog signal: the signal level is measured (and 'stored') at, say 25 microsecond intervals (corresponding to a sampling frequency of 40 kHz). Each sampled voltage level is then converted into a corresponding digital 'word'. The result, so far, is that the analog input signal has been converted into a rapid succession of binary numbers. Ignoring practical problems, which will be discussed later, the only theoretical sources of poorer signal quality have now been passed: the low-pass filtering at the input (limiting the band-width of the signal) and the conversion process with its associated quantization error. A digital signal is now available. It has the major advantage that it is extremely tolerant of abuse: it really takes some doing to maltreat this signal to the point that the individual binary numbers are no longer recognisable. The 'rapid succession of binary numbers' can be delayed, transmitted over long lines, stored on tape, etc . . . and in most cases the output will still contain sufficient information to recreate a 'clean' digital signal that is identical to the original input. Passing this signal through a digital-to-analog converter and an output low-pass filter produces the analog output signal. It will be obvious from the above that the analog output signal can never be identical to the original input signal. Quite apart from practical problems, the quantization process will always get in the way — dividing the analog signal range into a limited number of smaller ranges, and collapsing each of these into a representative 'centre voltage'. Quantization noise If the analog input is a high-level speech or music signal, the audible effect of 2 Block diagram of a complete digital audio system. To make use of the 'perfect' signal handling capabilities of the 'digital system' proper (for signal delay, transmission, storage or other manipulation), the other five Mocks must be added. Regrettably, since they do introduce distortion, noise, and other 'nastiness'. quantization will be very similar to white noise. The apparent signal-to- noise ratio is determined by the number of 'quantization intervals' into which the analog signal range is divided - and, therefore, by the number of bits used in the system, this is illustrated in figure 3. In figure 3a, the output from a 4-bit system (16 levels) is shown. This signal is equivalent to a mixture of the intended (sine-wave) output and an error signal, as can be seen; by way of comparison, figure 3b gives the result of mixing the same sine-wave with a noise signal. For each additional bit used in the sys- tem the number of available quantization intervals doubles, so the amplitude of the error signal is halved — effectively, the 'signal-to-error ratio' is improved by 6 dB. It is therefore reasonable to assume' that the signal-to-noise ratio in a digital system will be equal to 6dB multiplied by the number of bits — e.g. 72 dB for a 12-bit system. Considering the fact that 72 dB is quite good, as signal-to-noise ratios go, one might assume that a 12-bit system is good enough for most applications. If better performance is required, one could always add a few more bits — say, a total of 16 bits would give 96 dB signal-to-noise. Regrettably, life is rarely so simple ... In the first place, extra bits are expensive. it can be proved mathematically. This will be obvious if we take a closer look at the 12-bit system, as an example. 12-bits correspond to some 4,000 levels, whereby the firs? (or 'Most Significant') bit defines whether the required level is in the range 0 . . . 2047 of 2048 . . . 4096. The last (or 'Least Significant') bit, on the other hand, corresponds to one level step - from 1 234 to 1 235, for instance. This means that the level step corre- sponding to the first bit is some 2,000 times larger than that for the last bit. If the latter is to have any significance, the step for the first bit must be accurate to within 1/2000, or one-twentieth of one percent. Using 1% component toler- ances? Forget it! To make matters worse, this type of highly accurate level detection must be carried out at high speed: the complete analog-to-digital or digital-to-analog conversion must be completed within the sampling period - i.e. within 20 microseconds or so. For a 16-bit system, conversion accuracy to within approximately 15 parts-per- million is required . . . and 50,000 times per second, at that! It will be obvious that, at this rate, we are rapidly approaching the limit of present-day technology. To make matters worse, more bits are required in practice for a given signal-to- noise ratio than the 6 db-per-bit rule would imply. Speaking very broadly, one additional bit is required in a playback-only system (using pre- recorded tapes or records) and two additional bits are preferable in a full record-and-playback system. These bits are needed to counteract all sorts of nasty effects associated with the quantization process. Quantization nastiness When a normal audio signal at a suitable level is fed through a digital audio system, the quantization noise will usually be equivalent to white noise, and the signal-to-noise ratio will be 6 dB per bit. However, there are some very important exceptions to this rule, and in practice digital audio systems can sound much worse. Quantization distortion. As an example, assume that a low-level sinewave is applied to a digital audio system; the peak level is slightly less than one quantization interval (figure 4). Since the signal only crosses one quantization level, the output will be one of two possible digital 'words'. This is equiv- alent to a squarewave output: the system is operating as a hard limiter. In this case, the quantization error is equivalent to distortion — there is no noise in the analog sense! The audible result can be similar to crossover distortion in a power amplifier. Granulation noise and birdies. In the example given above, the quantization process introduced distortion. Similarly, if the input signal exceeds the maximum level for which the digital system was designed, 'hard clipping' will occur: all eloktor September 1979 - 9-25 levels above the maximum are coded and reproduced as equal to maximum level. Once again, the result is severe distortion: in other words, higher harmonics are added to the signal. As long as these harmonics remain within the permissible frequency range of the system (i.e. less than half the sampling frequency), the result will simply be a distorted output. However, when harmonics are generated above this frequency, things will really go wrong. The problem is that, effectively, these high frequencies are also sampled, producing sum and difference fre- quencies that can 'fold down' to within the audible range. As an example, assume that a 9.5 kHz sinewave is applied to a digital audio system that uses a 50 kHz sampling frequency. If distortion occurs as a result of the quantization process, harmonics can be produced at 19 kHz, 28.5 kHz . , . 47,5 kHz, 57 kHz . . . etc. As a result, 2.5 kHz and/or 7 kHz components may be produced. These will remain present in the analog output signal after the second low-pass filter. This type of error signal is neither noise nor distortion in the normal analog sense, since the new signal components are discrete frequencies but they are not harmonically related to the original signal. For this reason, they are far more irritating than either noise or distortion. This effect is sometimes referred to as 'granulation noise'; it sounds something like two pieces of sand-paper being rubbed together. In some cases, the beat notes may drift rapidly through the frequency range, producing an effect like birds singing. Modulation noise. The effects described so far are inherent to digital audio systems — even in theory. In practical systems, inperfections in the actual electronics are a further source of error. It it outside the scope of this article to discuss these in detail — interested readers are referred to an extremely good discussion by Mr. Blesser in the Journal of the Audio Engineering Society (see literature). Suffice it to say that, in general, the effect of these errors is that the noise output will vary with the analog signal, producing modulation noise. Severe errors could, of course, produce all kinds of other effects, but these are unlikely to occur in practice. Dither noise The noise and distortion products discussed so far have one thing in common: they are all more irritating and sound more unpleasant than white noise. Subjective tests show that this additional 'irritation' is equivalent to 6 ... 12 dB less signal-to-noise ratio. In other words, a 12-bit digital system with a measured SN-ratio of 72 dB will 'sound' approximately as good as a straightforward analog system with a signal-to-noise ratio of only 60 ... 66 dB . One way to cure this problem would be to add a few more bits - reducing the noise signal to the point where it is inaudible. However, additional bits are expensive. An alternative solution is to add a small amount of white noise to the analog input signal. The peak-to-peak value of this so-called 'dither' signal is approxi- mately equal to one quantization interval. Without going into (mathemat- ical) detail, it can be stated that this will effectively eliminate the 'quantization nastiness', and result in a deterioration of the signal-to-noise ratio of only 2 . . . 4 dB. The same 12-bit system mentioned above would then have an effective SN-ratio of 68 ... 70 dB. A good rule-of-thumb in practice is to assume that one bit is required to counteract the irritating effects of quantization noise. For a 16-bit system, for example, the SN-ratio will be at least 15 x 6 = 90 dB, and it may be one or two dB better. Peak overload prevention I f an audio system - any audio system - is overloaded, the output will be dis- torted. In a digital audio system, how- ever, the results can be disastrous. As mentioned earlier, if the input signal exceeds the maximum level for which the system was designed, 'hard clipping' will occur as a result of the quantization process. The resultant harmonics are effectively sampled, producing new frequencies within the audio band. To avoid this, the signal must be limited before the input low-pass filter. In a playback-only system (such as the new 4 Philips 'compact disc'), the peak program level can be monitored before the recording is made, so that limiting becomes merely a question of correct level-setting. If the system is to be suitable for recording 'raw' program material, however, the only safe solution is to add a hard limiter before the low-pass filter. The clipping level for this limiter will have to be set at approximately 3 dB below the nominal 100% level of the digital system, to ensure that the peak signal level will remain within the permissible limit even after low-pass filtering. Another way of looking at this is to say that the digital system must have at least 3 dB leeway above the nominal full-drive level; this costs one additional bit (since half -bits don't exist). The rule-of-thumb given earlier can now be extended as follows. If the 'dynamic range' of a digital audio system is defined as the number of dBs between the peak input level and the effective noise level, this dynamic range will be approximately equal to the number-of-bits-minus-one times 6 dB for a playback-only system and the number-of-bits-minus-two times 6 dB for a system that must also be suitable for recording. In the former case, the performance can be improved by 1 or 2 dB by careful design; in the latter case, up to 4 or 5 dB improvement is possible. This means that if a digital audio recorder is advertised as 'using a 16-bit system' and having a 'dynamic range of 86 dB', these claims are quite probable. On the other hand, if 96 dB is claimed for a 16-bit recorder, the designers must be extremely clever — or else the advertising copy-writer has slipped up. . . What of the future? Digital audio is here to stay. The twin advantages of guaranteed high perform- ance and reliability are too good to miss. Solving the practical problems discussed above is a question of time, and asdigital technology advances and prices come down it is to be expected that digital equipment will filter down the audio market until even the cheapest audio equipment goes digital. It is not difficult to envisage a point in the not-too-distant future when even the trusty LP disc is replaced by a PLOM (Play Only Memory) on a single (silicon?) chip. Meanwhile, those of our readers who are interested in an extremely full and detailed discussion of the theoretical and practical aspects of digital audio are referred to: Literature: 'Digitization of Audio: A Comprehensive Examination of Theory, Implementation, and Current Practice', Barry A. Blesser, Journal of the Audio Engineering Society, October 1978, Volume 26, Number 10, Pages 739. .. 771. M ic equalise The article on using an equaliser, also contained in this issue, gives a detailed discussion of the problems posed by deficiencies in the frequency response of loudspeakers and of the listening environment. It explains that the solution of these problems is to use an equaliser to adjust the overall frequency response of the hi-fi chain/listening environment. Use of an equaliser will therefore not be discussed in detail in this article. Before proceeding with a discussion of the parametric equaliser it is perhaps a good idea to discuss why it is superior to the more common 'graphic' equaliser. A 'graphic' equaliser such as the Elektor Equaliser consists of a number of band selective filters with fixed centre fre- quencies spaced at equal inter- logarithmic frequency scale, usually at octave inter- vals. though more expensive units may boast third-octave filters. Each of these filters s equipped with a gain control so that it can apply boost or cut to the band of frequencies over which it is active. The rm 'graphic' arose from the common Figure 2 shows how the characteristics of a parametric filter section may be varied. Figure 2a shows variation of the gain, figure 2b shows adjustment of the bandwidth, while figure 2c shows adjustment of the centre frequency. Figure 3 illustrates the adjustments possible with the parametric tone controls. Figure 3a shows how variable boost and cut may be applied to the extremes of the audio spectrum, as with normal tone controls, while figure 3b illustrates the unique feature of the parametric tone controls, namely the adjustable turnover frequencies of the bass and treble controls. Having briefly discussed the differences between parametric and graphic equal- isers, the advantages of a parametric equaliser can now be illustrated. In a nutshell, the purpose of an equaliser is to make the frequency response of an audio reproduction chain flat by providing gain where there are dips in the response and attenuation where there are peaks. Figure 4a shows the response of a typical reproduction chain, as might be measured using an audio analyser. This has a number of obvious deficiencies. The 'grass' on the trace is due to a large number of sharp (high Q( resonances, which can be as P5 i t i A combination of stated variable filters and a highly specialised Baxandall tone control network is used in the 'parametric' equaliser described in this article, which offers considerable advantages over the more common 'graphic' equaliser. Use of a parametric equaliser allows the frequency response of a domestic hi-fi setup to be tailored to a degree previously only attainable in recording studios. Such is the versatility of a parametric equaliser that even sceptics who turn up their noses at audio equalisers may be forced to revise their opinions. slider poten- tiometers in such equalisers, whose slider position is er- oneously supposed by some to represent the frequency response of the system. However, the term 'graphic' will be used to dis- tinguish between this type of 'equaliser and the parametric equaliser. The only variables in a graphic equaliser are the gains of the ndividual filter sections, since the centre frequency and Q (which determines the bandwidth) of each filter are fixed. A parametric equaliser has fewer filter sections than a graphic equaliser, but all the para- meters of the filter are adjustable, e.g. gain, bandwidth and centre frequency. A block diagram of the Elektor para- metric equaliser is shown in figure 1. This consists basically of just three parametric filter sections - band selective filters whose gain, centre fre- quency and Q are all adjustable. De- ficiencies at the ends of the audio spec- trum are catered for by a parametric Baxandall-type tone control to provide bass and treble adjustment. These controls operate in a similar manner to the parametric filter sections, but employ lowpass and highpass filters rather than band selective filters. much as 20 dB deep. Fortunately these peaks and troughs are inaudible due to their very sharpness, since they each occupy a bandwidth of only a few Hz. This is perhaps just as well since it would be impossible to cancel out each of these resonances. If this 'grass' is ignored then the response becomes something like that shown in figure 4b, in which the major deviations from a flat response are more readily apparent. It is evident that the response falls off sharply below 50 Hz and above 10 kHz, that a large peak exists at about 750 Hz and a trough at about 6 kHz. In addition there is a slight 'ripple' in ‘ the response due to a number of peaks ! and troughs a few dB deep. If one accepts the fact that deviations of a few dB can be ignored (and that in any case they will be very difficult to eliminate) then the response curve can be simplified to that of figure 4c, which shows only the principal deviations from a flat response. These are the deficiencies that must be removed by an equaliser. Parametric or graphic? It is fairly obvious that to remove a peak or trough from the frequency response the correction applied must be the exact inverse of the deficiency, i.e. the boost or cut applied must be the same as the depth of the trough or height of the peak, it must be applied at exactly the right frequency, and the Q of the correction network must be the same as that of the peak or trough. It is apparent that these criteria can hardly ever be fulfilled by a graphic equaliser. Firstly, it is unlikely that the centre frequency of a peak or trough would coincide with the centre fre- quency of one of the equaliser filters. Secondly, since a graphic equaliser has filters with a fixed Q the shape of the filter response cannot be tailored to fit the curve of the peak or trough. In fact the only parameter that can be varied in a graphic equaliser is the degree of boost or cut. With a parametric equal- iser on the other hand, the gain, centre frequency and Q of a filter section may be varied so that it is almost an exact fit for the peak or trough which it is to eliminate. At the extremes of the spectrum Baxandall tone controls with variable gain and turnover frequency can be used to compensate for the 'droop' which occurs. Like the graphic equaliser, a parametric equaliser may have any number of filter sections. The filter sections are necessarily rather more complex than those of a graphic equaliser; however, since each filter section is considerably more versatile it is possible to achieve satisfactory results with fewer filter sections, so that the cost is comparable with that of a graphic equaliser. For normal domestic use an equaliser consisting of three parametric filter sections plus Baxandall tone controls should be quite adequate. Parametric filter section The block diagram of a parametric filter section is given in figure 5. The heart of the filter is a selective network, which will be described in detail later, whose centre frequency and bandwidth (Q) can be independently varied. The gain of the filter can be varied by a ganged potentiometer, PI. The selective network is a state-variable filter or two-integrator loop, which readers of the 'Formant' synthesiser articles will recognise as being essentially similar to the Formant VCF. However. in this circuit the centre frequency of the filter is manually controlled by a two-gang potentiometer Rj nt , whose two sections vary the time constants of the integrator stages. The Q of the filter, and hence the bandwidth, is varied by altering the values of Rq. Complete filter circuit Figure 7 shows the complete circuit of a parametric filter section. The state- variable filter around A1 to A4 is immediately recpgnisable, as is the variable gain amplifier, IC1. The Q determining resistors and potentiometers Rq become R6, R7 and P2, whilst the centre frequency is set by P3. This arrangement differs somewhat from that shown in figure 6. However, if R j n j were a potentiometer connected as shown in figure 6 then it would have to have an inconveniently large value if the desired tuning range were to be covered. The arrangement of figure 7 is electrically equivalent and allows the effective value of Rj n t to be 9-30 - eleklor September 1979 varied from 10 k with P3 at maximum to about 2.65 M with P3 at minimum. This allows the centre frequency of the filter to be varied between about 40 Hz and 10 kHz. The Q of the filter may be varied between about 0.45 and 5 using P2, while the gain can be ad- justed by PI between ±15dB, which should be more than adequate for room equalisation purposes. If desired the tuning range of the filter may be varied by changing the value of Rjnt. using the equation of figure 6 to calculate the required maximum and minimum values. Different components may then be substituted for P3, R12, R13, R15 and R16. The minimum value of Rj n t (P3 at maximum) is equal to R13 (R16), whilst the maximum value of Rjnt (P3 at minimum) is equal to P3a + R12 R13, Figure 6. Circuit of the state variable filte similarly for P3b, R 1 5 and R 1 6. The Q adjustment range may also be varied by altering the values of R8, R9, R10, R 1 1 <= R) and R6/P2a, R7/P2b (= Rq), using the second equation given in figure 6. However this informa- tion is included only for the benefit of experimenters, and the average constructor is advised to stick to the component values given. Tone controls The circuit of the parametric Baxandall bass and treble controls is shown in figure 8. This employs the same principles used in the parametric filter section. However, instead of using a band selective filter network the bass control uses a lowpass network connec- ted between two buffers A1 and A2, whilst the treble control uses a highpass o o «HH3 8 q Sfii.HIrt network connected between A3 and A4. The breakpoints of these filters can be varied, between 50 Hz and 350 Hz for the bass control using P3. and between 2 kHz and 13 kHz for the treble control using P4. The maximum gain of both controls can be varied between ± 15 dB using PI and P2. Construction To make the equaliser more versatile it was decided to use a modular form of construction so that as many filter sections as required could be included. This also means that the sophisticated i tone control section can be used as a unit in its own right by those readers who do not want an equaliser but would like a versatile tone control Each filter section is therefore built on an individual printed circuit board, the track pattern and component layout of which is given in figure 9, whilst a separate board is used for the tone controls, the layout of which is given in figure 10. The boards are so designed that, when they are stacked side by side, the output of one board aligns with the input of the next. The connection points for the potentiometers are all labelled with letters, which correspond to those printed in the circuit diagrams of figures 7 and 8. The interconnection of three filter sections and a tone control section to form one channel of a complete equaliser is shown in figure 11. If a stereo version is required then this arrangement must, of course, be duplicated. To avoid cluttering the dia- gram the potentiometer connections are shown to only one filter section and the tone control section. However, connections to the other three filter sections are identical. Since the inputs and outputs of each section have the same DC potential (zero volts) the input coupling capacitor Cl and resistor R1 are required only on the board connected to the input. On every other board R1 can be omitted and Cl be replaced by a wire link. Since the zero volt rails of each board are inter- connected via signal earth the '0' connection of every board except the tone control should be left unconnected, otherwise earth loops may occur. Only the '0' connection on the tone control board should be connected to the 0 V terminal of the power supply. For the power supply the use of a pair of the commonly available 1C voltage regulators is suggested. Alternatively, if the equaliser is to be incorporated into an existing system with a ± 15 V supply then it may be possible to derive the supply to the equaliser from this. The choice of a suitable housing for the equaliser is left to the taste of the individual reader. One point, how- ever, is worth noting. Adjustment of the equaliser is fairly time-consuming, but once the controls are set they shquld not require readjustment unless there are any changes in the reproduction chain or listening environment. It is thus a good idea to make the controls tamper-proof, for example by fitting a lockable cover plate in front of them, or by fitting spindle locks to the individual potentiometers. Alternatively the knobs could be dispensed with altogether, the ends of the spindles slotted to accept a screwdriver and the potentiometers recessed behind holes in the front panel. Bibliography: 1. The Elektor Equaliser, ElektorNo. 33 January 1978. 2. Kieis, D. Reduction of acoustic feedback in sound systems applica- tions; paper at the 44th AES conve- tion, Rotterdam, 1973. H Heavy stuff: audio at 200 watts One of RCA's specialities is high-power transistors. Three recent additions to the range are the BD 550, BD 550A and BD 550B. These three heavyweights are intended for use in quasi-complementary audio output stages, and a few of RCA's designs are described here. The most remarkable characteristic of the three transistors is their high collector-emitter breakdown voltage - especially the BD550B, with its VqeO of 250 V 1 The main specifi- To start the ball rolling, figure la gives the main characteristics are given in table 2. The design is not particularly revolutionary, but it serves to illustrate the principle. The main problem when designing high-power amplifiers is to find an output device that will withstand the high voltages and currents required; it must also have a sufficiently high slew rate to handle the highest audio frequencies at full drive. The BD 550A is a good start, but even it would fall short in an amplifier of this type. parallel in each half of the output stage - making four in all for one (mono) power amplifier. One driver is provided for each pair of output devices. The circuit is arranged so that T8...T10 operate as one very-heavy- duty NPN transistor; similarly. Til . . T13 simulate an almost complementary PNP tran- The quasi-complementary output stage is preceded by two long-tail pairs: T1/T2 and T4/T5; internally, the whole amplifier is DC- coupled. T3 and T7 are both used as current sources. Negative feedback (about 30 dB) is applied to the base of T2 via R6. The quiesc- ent current is set by T6; it is adjusted by VCBO v CEO VcerI r be ■ VEBO D 550 BD 550A BD 550B BD550A BD 5! >CER RgE ■ 100 r 'EBO VCEO V CER VcE'I’OV VcE “ 175 V Vce " 250 V Vce * 95 V Vce * ’50 V vce * 200 v VEB-5V 0.2 A 0.2 A; RfjE 3 '00 fl l c ■ 0.2 A; V C E = 10 V A; V C E * 4 v A; V C E - 4 V A; l B ” 0.5 A A; l B - 0.25 A l C - 4 A; V C E ■ 4 V 1C * 2 A; V C E ' 4 v 110 - 130 - 5 typ. 0.75 1.76 250 - 275 - 5 typ. 1 2 Without an accurate picture of the frequency response of the sound reproduction system, the use of an equaliser can do more harm than good. For this reason an audio spectrum analyser, which can pinpoint the deficiencies in a particular audio chain and/or listening environment, is a virtually indispensable piece of equipment for the equaliser-user. Attempting to set up a room acousti- cally by twiddling the controls on an equaliser and 'playing it by ear' is an almost certain recipe for heated tempers and high blood pressure, such is the difficulty of the task. To obtain any real benefit from an equaliser it is essential that the user knows exactly what changes he wants to implement in the frequency response of the audio system in question. It therefore follows that a reliable audio spectrum analyser is required to provide the acoustic infor- mation which is a necessary preliminary to effective equalisation. An audio analyser system basically consists of three sections: a test-signal source (pink noise generator), a micro- phone to monitor the output of the audio system under test, and a suitable means of analysing and displaying the energy level of the incoming signal. Broadly speaking, audio analysers fall into one of two types, depending upon whether the analysis is real-time or not. '■ Real-time analyser A real-time analyser is the most sophisti- ' cated, but also the most expensive way of obtaining a detailed picture of the spectrum of an audio signal. The operation of real-time analysers can be explained with reference to the block diagram of figure 1 . A broadband test signal is fed to the audio system under test. Normally the test signal consists of pink noise, which has a uniform energy level over the entire spectrum. The output of the audio system is picked up by a measurement microphone and fed to a bank of octave or third- octave filters, which split the input signal into a corresponding number of adjacent frequency bands. The output voltage of each filter is then rectified Jio-analyser iber 1979 - 9-39 and displayed. Various types of display are possible — a moving-coil meter, an 1 oscilloscope, or, as in the commercially available spectrum analyser shown in figure 2, a matrix of LEDs. The advantage of a real-time analyser is that it enables the average energy level of the entire spectrum to be determined at a glance. However, in view of the large number of displays and filter sections which are required, real-time analysers are not cheap. The above-mentioned pocket analyser of figure 2, together with a suitable noise generator, costs in the region of £ 600 — and that is only a fraction of what some of its 'larger brothers' can cost! Since however, the primary application of the analyser is to monitor the response of an audio system to a constant test signal (the output of the pink noise generator, which has a uniform spectral intensity) real-time analysis is something of a superfluous luxury. A much cheaper, but none the less satisfactory arrangement is to have a single tuneable filter, which can be swept up and down the frequency spectrum as desired. This is in fact the solution adopted in the Elektor audio analyser. EH - 0 - 0 - T© 0-0T® 45 M Figure 1. Block diagram ol a real-time tpeetrum analyser. The Elektor audio analyser I The block diagram of the Elektor, non real-time analyser is shown in figure 3. ' As can be seen, the basic principle of spectrum analysis remains the same, the i only difference being that a single filter . and display are employed, resulting in a considerable saving in cost. As far as the placing of the filter is concerned, three possible configurations come into consideration. In figure 3a the variable 2 t Figure 2. Photograph of a commercially f available hand-held real-time analyser, I incorporating a LED-matrix display. filter is situated between the pink noise generator and the input to the audio system, whilst in 3b it is fed from the output of the microphone. In figure 3c two filters are employed in an effort to obtain the best of both worlds. Although in theory there should be no difference between these three arrangements, things are not so simple in practice. With the configuration shown in figure 3a, all manner of interference and stray noise can reach the microphone and adversely effect the measurement. With the arrangement of figure 3b, this problem is effectively obviated, since only interference which lies within the passband of the filter can reach the microphone. A disadvantage of this set-up, however, is that only a very small portion of the pink noise spectrum is used, whilst the audio system in question is of course required to reproduce signals over the entire range of audio frequencies. The arrangement of figure 3c thus represents the ideal solution, however in view of the increased cost and complexity of two tracking variable filters, it was decided that, for this type of application, one of the simpler circuits (figures 3a and b) would prove sufficient. The basic requirements for an analyser of the above type are therefore: — a pink-noise generator — a bandpass filter with stepwise or continuously variable centre fre- quency — a suitable microphone with preampli- — a rectifier circuit — a display circuit As far as the choice of microphone is concerned, it is clear that, unless it itself has a fairly flat response, one cannot hope to obtain an accurate picture of the response of the audio system/ listening room under test. For this reason it is important to invest in a reasonably good quality microphone capsule and preamp. As a display circuit, a multimeter is as good as any, and has the advantage of being cheap and commonly available. The remaining circuits, which form the heart of the analyser - and the substance of the rest of this article — are shown in figures 4a, 4b and 4c. Noise generator As can be seen from the circuit diagram of the noise generator shown in figure 4a, it in fact consists of a pseudo-random binary sequence generator, which has a longer than normal cycle time. This ensures that the noise has a high spectral density and that it is not characterised by the annoying 'breathing' effect obtained with short cycle times. The length of the shift register (IC1 ... IC4) is 31 bits, and since the frequency of the clock generator (N5 . . . N7, Cl, C2, R3, R4) is roughly 500 kHz, the full cycle time is approximately an hour and a quarter! EXOR-feedback is provided by N1 . . . N4. The circuit however has no anti-latch up gating. Instead there are two pushbutton switches; the START button ensures a logic 1 at the data input Q 0 of the shift register (pin 7 of IC1), thereby starting the clock cycle. The cycle is inhibited by pressing the third octave filter described in the R40 and R41 are added. Table 1 iists STOP button, S2. In this way it is article on the CMOS noise generator in the various resistance values required to possible to (temporarily) disconnect the Elektor 33 (January 1978). The output give the I SO standard centre frequencies, noise source without switching off the level of the filter can be varied by means When calibrating a parametric equaliser, supply voltage - a useful if not down- of potentiometer PI, whilst the centre a filter bandwidth of less than 1/3 of an right indispensable feature. The frequency can be varied between octave is required. By altering the value (pseudo-) white noise output of the approximately 40 Hz and 16 kHz by of R16 to 220 and replacing R1 7 by shift register is fed to the pink-noise means of the stereo potentiometer a wire link a bandwidth of approximately filter formed by R5 . . . R11, P2a/P2b. If stepwise control of the 1/12 of an octave can be obtained. C5...C11, before being amplified in centre frequency of the filter is desired, the circuit round A1. P2a/P2b can be replaced by a pair of attenuator networks and a twin-ganged Rectifier Circuit switch. The necessary modifications are It is of utmost importance that the Bandpass filter detailed in figure 5. Resistors R20 and amplitude of the test signal be measured This section of the circuit (shown in R22 are replaced by a wire link, the accurately. If a pink noise test signal is figure 4b) is virtually identical to the values of R21 and R23 are altered, and used in conjunction with filters which Figure 4c. The rectifie 9-42 - ele have a constant octave or 1/3 octave bandwidth (i.e. filters with a constant Q) Table one should really measure the RMS value of the noise — not an easy matter. Fortunately, however, a reasonably simple alternative exists - namely to measure the average of the modulus value, i.e. the average of the full-wave rectified noise signal. This is obtained by feeding the output of the peak rectifier to a lowpass filter. The rectifier circuit is built round IC8. The input level control is followed by an amplifier, A5. The actual (full-wave) rectification is performed by A6, A7, R27 ... 31, D1 and D2. The output of A7, which always presents a low impedance, is connected via R32toC16. Because this capacitor has the same charge and discharge time, the voltage on the capacitor will equal the average value of the full-wave rectified noise voltage. The time that this voltage remains stored on the capacitor is determined by the RC time constant. R32-C16, or, if S3 is depressed, R32/R33-C16. Depressing S3 causes C16 to charge and discharge much more rapidly, so that the capacitor voltage will follow rapid variations in the noise voltage. Thus S3 is intended to provide a rapid overall view of the variations in noise level for different centre fre- quencies of the filter. For accurate measurements, the longer time constant of R32.C16 should be used. After being amplified in A8, the voltage on C16 is displayed on the multimeter. An offset control is provided (P4, R34 . . . R36) to enable the meter to be calibrated accurately (zero deflection under quiescent conditions). Construction A printed circuit board, which is shown in figure 6, has been designed to accom- modate the circuit of figures 4a, b and c. 31.5 1/1 31.5 1/3 40 1/3 50 1/3 63 1/1 63 1/3 80 1/3 100 1/3 125 1/1 125 1/3 160 1/3 200 1/3 250 1/1 250 1/3 315 1/3 400 1/3 500 1/1 500 1/3 630 1/3 800 1/3 1000 1/1 1000 1/3 1250 1/3 1600 1/3 2000 1/1 2000 1/3 2500 1/3 3150 1/3 4000 1/1 4000 1/3 5000 1/3 6300 1/3 8000 1/1 8000 1/3 10.000 1/3 12.500 1/3 16.000 1/1 16.000 1/3 202 + 2S22 202 + 202 506 407 +202 407 +309 407 +309 10 O + 102 10 0 + 309 12 O + 506 12 0 + 506 22 0 27 0 + 108 33 O + 202 33 O + 202 22 O + 22 O 560 68 O + 303 68 O + 303 82 O + 802 1000 +180 100 O +47 O 100 O +47 O 120 0 + 68 0 220 0 + 27 O 270 O + 47 O 270 O + 47 O 390 O + 18 O 470 O + 68 O 680 O + 47 O 680 O + 47 O 820 0 +150 0 Ik + 390 O 1 k8 + 330 O 1 k8 + 330 O 3k3 + 390 O 5k6 + 1 k 39 k +1k2 39 k + 1k2 18k w 68 k 8k2 68 k 8k2 68 k 8k 2 68 k 8k2 68 k 8k2 68 k 8k2 18k w 68 k 8k2 68 k 8k2 68 k 8k2 18k w 68 k 8k2 68 k 8k 2 68 k 8k2 18k w 68 k 8k2 68 k 8k2 68 k 8k2 18k w 68 k 8k2 68 k 8k2 68 k 8k2 18k w 68 k 8k2 68 k 8k 2 68 k 8k2 18k w 68 k 8k 2 68 k 8k2 68 k 8k2 18k w 68 k 8k2 68 k 8k2 68 k 8k2 18k w 68 k 8k2 Remarks: column 1 : centre frequency in Hz column 3: value of resistor to be connected between the junction of resistors R40 and R21 and ground and between the junction of R41 and R23 and ground, rounded up to values from the E 1 2 series. column4: value of R16 column 5: value of R1 7 (w » wire link) 5 The design of the board is such that either of the configurations shown in figures 3a and 3b can be adopted. The construction of the standard version | circuit should present no special problems. The wiring for the poten- tiometers and switches should be kept as short as possible. The connections for ■ these components are arranged at one end of the board. Problems of a practical nature do arise, however, if one desires a number of switched filter frequencies, since one then requires a switch with a corresponding number of ways. Since switches with a large number of ways are both expensive and difficult to obtain, an alternative solution is simply to use the desired number of double-pole single-throw switches. This of course involves operating two switches each time one wants to alter the centre frequency of the filter. In addition to the switch(es), the choice of fixed filter frequencies involves the following alterations on the board (see 9-44 - ele > r septe sr 1979 figure 5): R21 and R23 become 4k7 R20 and R22 are replaced by a wire link a 4k7 resistor (R40) is soldered between the 'top' two tags of P2a a 4k7 resistor (R41) is soldered between the 'bottom' two tags of P2b The resistor pairs forming the switched attenuator network are mounted exter- nally on the switch(es). Suitable values are given in the table. With a continuously variable filter frequency it is useful to equip P2a/b with a pointer and scale. The scale can of course be calibrated in frequencies, but it is not strictly necessary. What matters is that one has a series of reference points - peak or dip at such and such a filter setting, etc. If, however an absolute frequency scale is desired, this can be obtained by using a tone generator and noting the frequency when the output voltage at point C is at a maximum, when feeding a pure sine- wave into point B. Using the analyser The multimeter (10 to 12 V full-scale deflection) which is used to display the amplitude of the noise signal is connected to the output (point E) of the rectifier circuit. In the absence of an AC drive voltage (i.e. point D disconnected or else P3 turned right down) the DC voltage at this point should be set by means of P4 to exactly 0 (m)V. The correct setting for P4 is obtained by repeatedly switching down the voltage range of the multimeter and checking the reading by reversing the polarity of the probes. It should be borne in mind that, because of the long time constant of R34 and C16, it will take some time for adjustments to P4 to have any effect. The long discharge time of the storage capacitor in the rectifier circuit together with the natural inertia of the meter ballistics ensure that the needle responds only very slowly to changes in the level of the filter output. Thus when sweeping the filter up and down the audio spectrum, care should be taken to vary the filter frequency gradually, lest peaks or dips in the response are camouflaged by the slow response of the circuit. If the analyser is used to measure a system with a completely flat response, the mean meter deflection (i.e. the mean between the maximum positive and negative deflections) should be independent of variations in the filter frequency. An audio system with a completely flat response would be pretty hard to find, however, something which does have a more or less flat response is a wire link! — by joining points A and B and C and D in this way (i.e. connecting the output of the noise generator to the bandpass filter and the output of the filter to the rectifier circuit) it is possible to test the operation of the audio analyser, and in particular, of the pink noise and bandpass filters. Variations of up to ± 2 dB (0.8 ... 1 .25) in the mean meter reading are accept- able. To prevent the rectifier circuit from being overloaded, the mean meter reading can be adjusted to occur at around 3 ... 4 V. Finally a word of warning: care should be taken to ensure that the noise signal does not overload one's audio equip- audio-analyser Figure 7. A prototype of the audio analyser. ment. The risk of this happening is somewhat greater than in the case of a sine or squarewave input signal, since the distortion caused by overloading will be that much less noticeable (but none the less disastrous!). Tweeters in particular are susceptible to damage by being overloaded with high level noise signals. Constructing the audio analyser is one thing, using it is another. The reader is therefore referred to the article on 'Using an equaliser', which deals with the subject of using the equaliser/ana- lyser combination to measure and then correct a room's response. Literature: 1. 'Digital noise generator', Elektor 21 , January 1977 2. 'CMOS Noise generator', Elektor 33, January 1978 H L TAP 1979 - 9-45 Over the years there have been numerous circuits designed to protect one's car from the attentions of thieves. Many of the designs have aimed at foiling the person who succeeds in bridging the ignition contacts or who has a false key. In such cases the usual idea is to employ a second switch in the lead to the ignition coil, which is hidden or camouflaged from the thief. In principle this approach is quite attractive, how- ever it does have a couple of drawbacks. Firstly, the switch must of course be well hidden, and yet within reasonably easy reach of the driver — two seemingly conflicting requirements. Secondly, considerable vibration. Many small relays used in cars are provided with flat contact 'tongues', which are ideal for this type of application. By employing a slight trick, it is possible to ensure that when the car goes into the garage for repairs or servicing, there is a simple way of keeping its 'secret' well hidden. If point 1 of the circuit is connected to one of the 'forks' of the contact tongue, then before taking the car into the garage, one simply connects point 2 of the circuit to the other 'fork', so that the car then starts normally. It will be apparent that, with only minor modifi- cations to the relay connections, the TAP thieves on the head Touch activated anti-theft device for cars E. Schorer once the ignition is switched off and the ignition key removed, the second, concealed switch must also be in the off position, otherwise the anti-theft circuit is pointless. However it is all too easy to forget to operate the concealed switch when leaving one's car in a hurry. The circuit shown in the accompanying diagram represents an attempt to get round both these problems. To start the engine the ignition switch, SI, is first closed. This however fails to energise the ignition coil, since the contacts of the relay, re/a, which is inserted in the ignition lead, remain open. If however the touch contacts are bridged with the finger, a small base current will flow through T1, turning on this transistor and the Darlington pair, T2 and T3. As a result, the relay, re, pulls in, and once the contact re/b is closed, the relay will remain in that state. The engine can now be started normally. When the ignition switch is opened, the relay will automatically drop out, thus 're-arming' the anti-theft facility. The circuit itself is quite straightforward. The RC network, R3, C2, which is included in the supply line of T1, and the stability capacitor Cl, shield the circuit from the effects of any voltage transients generated by for example the wiper or heater motor, which may already be in operation before the relay is pulled in. This prevents the relay being actuated spuriously. As far as the design of the touch switch is concerned, it is left up to the indi- vidual to choose the optimal form of camouflage. A suitably reliable and robust type of relay should be used, since it will obviously be subject to circuit can be used as a touch switch with many applications in the car (e.g. windscreen washers, wipers etc.). The circuit can easily be mounted on a small board, roughly 1,5x4 cm. It is recommended that both sides of the circuit be covered with a layer of protective lacquer. M to R a . The positive feedback, which in figure 2 was realised via R c and Rfl, may at first sight not be apparent in the circuit of figure 3, however it is present. Due to the I delay introduced by the CMOS gates, the circuit in fact oscillates in the same way as a conventional CMOS oscillator. The duty- cycle of the output waveform is adjusted to 50% by means of P2 (with the inputs shorted), A breadboarded version of the above circuit worked satisfactorily without a loudspeaker. A distortion of 2% was measured with an audio output signal of 6 V pp . However, once the circuit was connected to a loudspeaker, the distortion rose to a completely unaccept- able 40%. Current sources circuit can be expected if R a and Rb in figure 2 are replaced by controlled current sources (see figure 4). Capacitor C is then charged and discharged by currents which can be regarded as remaining constant for the duration of each switching cycle. In the long term, the current ij n is directly proportional to the input voltage Uj n . The output current, i„, is proportional to the asymmetrical squarewave output voltage, u 0 . When u 0 is Bubble etcher for P.C.B. production. punched tape or M. SWIFT-SASCO ily its full range of EPROMs to cus- Powerhouse Microprocessors Ltd.. 2 Gresham Road. 5 7 Alexandra Road. Brentwood. Essex. Heme / Hempstead. Telephone: 102271 227050, Herts. HP25BS. Tel: 10442148422 (12 elektor September 1979 • UK17 CFMQ serii excellent ski bra ted. Wahl International Ltd.. BEAM Communications Ltd.. 117 Piccadilly, London W1 V 9FJ, Telephone: 01-491 3502 ^ 225 M , stopband below -70 dB. Priced at less than £ ' types. TOKO CFM fi ransformer Pocket-sized digital thermometer The new Wahl Digital RTD- Platinum Heat Prober ther- mometer is a pocket-sized high Ambit International, 2 Gresham Road, Brentwood. Essex, Telephone: 10277) 227050 Mechanical IF filters TOKO introduce the w smallest mechanical IF filtf frequencies in the range 4 480 kHz. The basic mechanical elem market features a selection of 15 inter- changeable RTD-platinum probes or laboratory application over temperature ranges of -50°C to 500 C. Large, easy to read LED display reads to 0.1°C. Accuracy of ± 0.5% 1 1 digit. Repeatability of ± 0.2%. Built-in rechargeable bat- ELEKTOR PUBLISHERS LTD., ELEKTOR HOUSE, 10 LONGPORT. CANTERBURY. CT1 2BR. Surname Street/Ave./Blvd. County /province/state Country ELEKTOR BOOK UK30 - elektor September 1979 Fotolak POSITIVE LIGHT SENSITIVE AEROSOL LACQUER G.F. MILWARD ELECTRONIC COMPONENTS LTD 369 Alum Rock Road. Birmingham B8 3DR. Telephone: 021-327-2339 Elektor book service ADVERTISERS INDEX BOOK 75 AJD Supplies UK11 Ambit International UK16 Audio Electronics UK12 Catronics UK28 Classified UK29 Chromasonics UK26 Codespeed UK26 Contour Electronics (Comtech) UK9 Cossor Electronics UK14 David George Sales UK14 De Boer Elektronika UK7 Elacom UK16 Elektor UK14, 15, 17, 18, 22 23, 24, 25, 26, 28, 30 Ferranti UK25 Fraser-Manning UK13 G.F. Milward UK30 G.M.T. Electronics UK19,20,21 Greenbank Electronics UK16 Greenweld Electronics UK12 Keytronics UK25 Maplin UK2 Marshall's UK10 Monolith Electronics UK27 Phonosonics UK27 Ramar Electronics UK27 T. Powell UK32 Technomatic UK31 Vero Electronics UK18 Watford Electronics UK8, 9 'DATA' he prole DIGIBOOK This bn Elektor digital electr Elektor's tyt backed DIG IBOOK ' £ DIGIBOOK price £4.50 (DIGIBOOK PLUS PCB) post & pack 25p extra USA & Canada S9.50 books please use the Elektor reader