m m t U.K. 50 p. U.S.A./CAN. $1.50 unm *9 equaliser uaa 180 ppm ■■ ■ function generator noise generator Austria Belgium Denmark France Germany Nethelands Norway Swed.*n Switzerland S 33 f 58 DM 3 80 DFL. 3 /5 K r 9 Kr. 9 mcl. moms E-4 — elektor january 1978 decoder elektor 33 Volume 4 Number 1 Editor : W. van der Horst Deputy editor : P. Holmes Technical editors : J. Barendrecht G.H.K. Dam E. Krempelsauer G.H. Nachbar A. Nachtmann K. S.M. Walraven Subscriptions : Mrs. A. van Meyel International head offices: Elektuur B.V. P.0, Box 75 Beek (L), Netherlands Telex: 56617 Elekt NL U.K. editorial offices, administration and advertising: Elektor Publishers Ltd., Elektor House, 10 Longport Street, Canterbury CT1 1PE, Kent, U.K. Tel.: Canterbury (0227) -54430. Telex: 965504. Please make all cheques payable to Elektor Publishers Ltd. at the above address. Bank; 1, Midland Bank Ltd*, Canterbury, A/C no, 11014587 Sorting code 40-16-11, Giro no. 3154254, 2. U.S.A. only; Bank of America, c/o World Way Postal Center, P + 0* Box 80689, Los Angeles, Cat 90080, A/C no* 12350-04207, Assistant Manager and Advertising ; R.G* Knapp Editorial : T, Emmens ELEKTOR IS PUBLISHED MONTHLY on the third Friday of each month* 1. U.K, and all countries except the U.S.A. end Canada: Cover price £ 0,50* Number 39/40 (July /August), is a double issue, 'Summer Circuits', price £ 1. — . Single copies (incl. back issues) are available by post from our Canterbury office, at £ Q + 60 | surface mail) or £ 0.95 (air mailt. Subscriptions for 1978, January to December inch, £6.75 (surface mail) or £ 12.00 (airmail). 2. For the U.S.A, and Canada; Cover price S 1 .50. Number 39/40 (July/August), is a double issue, 'Summer Circuits', price S 3. — , Single copies (incl. back issues) $ 1.50 (surface mail) or S 2.25 (air mailt. Subscriptions for 1978, January to December incl., S 13. — (surface mail) or $ 27. — - (air mailt. All prices include post & packing. CHANGE OF ADDRESS. Please allow at least six weeks for change of address. Include your old address, enclosing, if possible, an address label from a recent issue. LETTERS SHOULD BE ADDRESSED TO the department concerned; TQ = Technical Queries; ADV = Advertisements; SUB - Subscriptions, ADM - Administration; ED = Editorial (articles submitted for publication etc.); EPS ~ Elektor printed circuit board service. For technical queries, please enclose a stamped, addressed envelope or a self-addressed envelope plus an IRC. THE CIRCUITS PUBLISHED ARE FOR domestic use only. The sub- mission of designs or articles to Elektor implies permission to the publishers to alter and translate the text and design, and to use the contents in other Elektor publications and activities. The publishers cannot guarantee to return any material submitted to them. All drawings, photographs, printed circuit boards and articles published in Elektor are copyright and may not be reproduced or imitated in whole or part without prior written permission of the publishers, PATENT PROTECTION MAY EXIST in respect of circuits, devices, components etc. described in this magazine. The publishers do not accept responsibility for failing to identify such patent or other protection. National ADVERTISING RATES for the English-language edition of Elektor and/or international advertising rates for advertising at the same time in the English, Dutch and German issues are available on request. DISTRIBUTION in U.K.; Spotlight Magazine Distributors Ltd., Spotlight House 1, Bentwell Road, Holloway, London N7 7AX, DISTRIBUTION in CANADA; Gordon and Gotch (Can.) Ltd., 55 York Street, Toronto, Ontario, M5J 1S4. Copyright © 1978 Elektor publishers Ltd - Canterbury. Printed in the Netherlands. decoder What is a TUN? What »s 10 n? What is the EPS service? What is the TQ service? What is a missing link? Semiconductor types Very often, a large number of equivalent semiconductors exist with different type numbers. For this reason, 'abbreviated' type numbers are used in Elektor wherever possible: * '741 1 stand for ^A741 , LM741, MC641, MIC741, RM741, SN72741, etc. • TUP' or TUN' (Transistor, Universal, PNP or NPN respect- ively) stand for any low fre- quency silicon transistor that meets the following specifi- cations: UCE0, max 20V 1C, max 100 mA hfe, min 100 Plot, max 100 mW fT, min 100 MHz Some 'TUN's are: BC1Q7, BC10S and BC109 families; 2N3856A, 2N3859, 2N3860, 2N3904, 2N3947, 2N41 24. Some 'TUP's are: BC1 77 and BC1 78 families; BC1 79 family with the possible exeption of BCt59and BC179; 2N241 2, 2N3251 , 2N3906, 2N4126, 2N4291. • 'DUS' or 'DUG' uJiode Univer- sal, Silicon or Germanium respectively) stands for any diode that meets the following specifications: DUS [DUG UR, max IF, max |R, max Plot, max Cp, max h 25V^ 100mA IjuA 250mW 5pF 20V 35mA 100 jiA 250mW 1 0pF Some 'DUS's are: BA1 27, BA21 7 r SA218, BA221, BA222, BA317, BA318, BAX13, BAY61 , 1N914, 1N4148. Some 'DUG'S are: OA85 r OA91 , OA95, AA116, • 'BC1 Q?B', 'BC237B', 'BC547B' all refer to the same 'family' of almost identical better-quality silicon transistors. In general, any other member of the same family can be used instead. BC107 (-8, -9) families: BC107 (-8, -9), BC147 (-8,-9), BC207 (-8, -9), BC237 (-3,-9), BC317 (-8, -9), BC347 {-8 r -9), BC547 (-8, -9), BC171 (-2, -3), BC182 (-3, 4), BC382 (-3,4), BC437 (-8, -9), BC414 BC177 (-8, 9) families: BC177 (-8, -9), BC157 (^8, -9), BC204 (-5,-6), BC307 (-8,-9), BC320 (-1 , -2), BC350 M, -2), BC557 (-8, -9), BC251 1-2, -3), BC212 (-3, 4), BC512 (-3, 4), BC261 (-2, -3), BC4 1 6 . Resistor and capacitor values When giving component values, decimal points and large numbers of zeros are avoided wherever possible. The decimal point is usually replaced by one of the following abbreviations: P (pico-) = 10^ n (nano-) = 10* v (micro ) - 10"* m (mi 111-) = 10~ 3 k (kilo-) = 10 3 M (mega-) - 10 6 G (giga) = ID* A few examples: Resistance value 2k7: 2700 H. Resistance value 470: 470 H. Capacitance value 4p7: 4.7 pF, or 0.000 000 000004 7 F . . . Capacitance value IGn; this is the international way of writing 10,000 pF or ,01 juF, since 1 n is 10~ V farads or 1000 pF, Resistors are * Watt 5% carbon types, unless otherwise specified, The DC working voltage of capacitors other than electro- lytics) is normally assumed to be at least 60 V. As □ rule of thumb, a safe value is usually approxi- mately twice the DC supply voltage. Test voltages The DC test voltages shown are measured with a 20 kST/V instru- ment, unless otherwise specified* U, not V The international tetter symbol 'U r for voltage is often used instead of the ambiguous J V'. 'V' is normally reserved for Volts'. For instance: U b - to V, not V b = 10 V. Mains voltages No mains (power line) voltages are listed in Elektor circuits, it is assumed that our readers know what voltage is standard in their part of the world! Readers in countries that use 60 Hz should note that Elektor circuits are designed for 50 Hz operation. This will not normally be a problem; however, in cases where the meins frequency is used for synchronisation some modifi- cation may be required* Technical services to readers • EPS service Many Elektor articles include a lay-out fora printed circuit board. Some — but not all — of these boards are avail- able ready-etched and predrilled. The 'EPS print service list' in the current issue always gives a com- plete list of available boards. • Technical queries. Members of the technical staff are available to answer technical queries (relating to articles published in Elektor) by telephone on Mondays from 1 4.00 to 1 6.30. Letters with technical queries should be addressed to: Dept. TQ. Please enclose a stamped, self addressed envelope; readers outside U K. please enclose an I RC instead of stamps. • Missing link. Any important modif ications to, additions to r improvements on or corrections in Elektor circuits are generally listed under the heading 'Missing Link' at the earliest opportunity. contents elektor january 1978 — E-5 :::::: ::: !>:i ■ :& Si-: > 3 #:': :": v - ■■■■■ ■■ :::: hi; :::V s®, i hi;: " T -V * * !i 1 IBi* :• ,*.*:*%' | * i m I ttf'H I hi® % i’ i#*v : ‘ Wf. - -!J: ■ : -ip ?|PlPip "T' i&p.: 1 | j - l ■ x:::. -m :i:: : - The elektor equaliser is a so-called octave equaliser, whereby the gain within each octave can be individually varied, in PA systems, for instance, this unit can be used to obtain a flat frequency response by presetting the individual gain controls. Alternatively, the unit can be used as a powerful weapon with which to modify a system's frequency response as required, and for this application it is recommended to use rotary potentiometers instead of presets. -i:::::; ... :... .W* ".h” ..... . . .... ^ V ■ t ; ■. ::: ?::: , ... xu. Mlh :;h; . ■::: - : :#i: W&: -.¥$#: j'j+H .. .. . : :::::: - € :: 4 fl ■ -■I'lir ■■•in ...:. JtW^v -xiyX;: v::*:: :::: ^ : • ■ :: 1: ;;p m §1 V! XX ... ... ... . ipP ; i w ■ism!?.:- • ::>> • - " " : • •>! ......... -x-Swwfi: ::: ...... ....... •f®"' M W 28p Posor Neu 5V 12V 10V 18V As above Tandem 82p sing-es with D.P. switch 65p Vertical Si Horizontal LM3Q9K 1.45p LWI31 7 3.35p Pihef Presets 1 00 to 10 mEg 12p 14 and % watt 5% carbon film resistors El 2 2p LEDS Red Green Yellow D LL. Sockets 8 pin 12p14pin 14p1Gpin 16p .125" I5p 27p 72p For T.T.L, See previous Advertisement ^ 16 P 30p ALL COMPONENTS BRANDED AND NEW, LET US QUOTE' FOR YOUR PROJECT REQUIREMENTS, JUST SEND S.A.E. WITH YOUR LIST - NO MINIMUM ORDER CHARGE. TEST EQUIPmEIIT AM pr iL«i me lir dft V A r | ■ AMP I L F- 1 h Kh DIGITAL MULTI- METERS new s i r-j*: : i air POM 3!i WlCK£ I !JMM £3195 CJP 45p 5N rfin? 3 N n SNJKMiEiEIM E'i rtnsi 4 fihmE £2.0( IHA 0 AO S -Ann 1 5 rjh.rni Ci.4( IfiASlrt t Mini 4 nhms C1.4C TiSAS?& 2 wad B nkms ... . . L1.2E AM35P 35 wad S fErn:. . . CS.Et CLOCKS ia !R dal -si WWS3l4 EJ.'ZA HR 4^5 diflili Lf Dfojin.aplHT display .. £fl.&5 MMH16 ,i. !•••• in iq! , |- i v. LEO iJisfi j>' . • £2.95 DlCil I A I CHIPS 'v.Hl iIhI.-I MCI 4553 0-999 deci mal cnuntfir £-q. 7 fl ICLTIPT 3K. Jig.r fcai mul tinrccc* hnsic lL£f)l ..... FIRM ICLJ! 06 3 k- rligr {fignsl myliirn^tHr bask ILCOI £T5,5jJ TV GAMES fwnh rlsrni A Y-3-8MQ 6 jjarpes.'scCTrng/HiLHi-rE ,, ...LI 99 FI f K I OK p[; PANEL F? OR IJ-ISP! AYS V( £4.M} !>B ?0l lun^riLifl i u be 1 1 BO ?Q0 V’> £1.67 Tunas; 4 ^ h iji-gii MEX uvr pj.. fl.flR Z VI OBO n Tim ,c a I rulrf 1 1 SO - 200 VI. ri .57 CRYSTALS - MH, HC 6 . F.J lOO PtH,- 1 1 MHr llh . ... , f2 7R 2 MH/. 4 MHa Hr:£l,J +> Raff r can uni ,>jiri 14 W, fcH^IFI |]a, £3 tid 1 0. 7 W ) \J i ii non l 1 > s i.h! IE Filter 25 kNj hanrltk."ilEh £7.- Lar^K range o' LCDs ? SMP , r n ri* ddE) iiyt • <1 ■ - lui i r tersl KEYBOARD 19 SLVi-i:*in! . gni rl pi a«ed wi() r :| .- 1. 1 it 1 e a] X 7,1 CMS 7 "ni f 1.50 REED REV-ITCHES A ma ;.-iq r- n-udeg 1 !)i?n CMS - Of ' 1 fJ £ 2 50 JOYSTICK CONTROL HiijI’ gus.i;’, TV Sbjn;~ - on r 2 u> IrtteiJ 4 k 100 k Lin vanned FM Tuner Tqnrsr Lm 1 C3.SD 1UF :E5,s 13 35 MiEll fflf iri^h ll’M.I SINCLAIR DM2 W'5'iiJ ! in n-vijK 1 her r 11 1 ip' 1 E52.GK? C R 75|i AC/OC vd tv & LLir> 'i-Mi 1 inVyO .1 A n|ji ■ui-i.il l.il 'V i'ilj Reiis-rariim I ln :70 M .ieliji,k.v' e»5E )EB,4S'eirtTaJ 1 'jf. I Jig y PilUL riMt 1 bH !-. IK( ,P ||||I Fegr^r rq AC/UC ^ rx s r r. ijc.' i.unei»r. . 1 1 . 1 1 - . . r.mtpt k-ih TMK 509 30 k per voli uy.Ub LT701/TI I kAroM k*iei 1 vly C6.95. 11 I 2 70 k.'uc: 1 bld.LKi. fIJ b k Pfr uijli ... £7.95 LJ^EKI s ' r. ■' 1 1 ’.■! . i J ' : n t : I V rm .■ .■!. ■ up in 3C 1 P.l t 1 1 £16.95 TflBI RfJ k . L*r>t ■ • ,ji! 10 riinni!] AC vdHi, I 2 i.inoi- Dl , •% I |E ranges QC rgiri'm 111 V' 1 ■ A In 111 1 1 ■■ i >| n 1 i .ii ii^--- mi ki^nn . m 1 R Mei, £22.56 6 BDH k.'Wlir C^7.2S HO tlli.lLi .Uvd-TFi 1 nn L i'v.J I 7J raryj,-. iflin lrBr.sij.1oEk7iEcM.fT £.12,59 G*naral Mixi.63 ICE ■ e; i q ■ ■ -ii in|sc:or wide- rnnon ourpa! f7 5 fl 1C F lOp.' LSI F'NPi'NPN i >ii lEOi 1 1 I.km ICONIobia.'Lieia £71 .5d - R A IT F AMI fc KPEIK r SUPPLIED B.jildinq vgur r.- i- speaker ivitcrT'f' 1, E ' :j m r h * i y h T ^1 iK.ik h i h h iicl ki l\ Also mHLmpliiiikfH, and T-.Kfri Irpm Siock, SEE UHlri Inqi ik i.r.'i - 1 ■ i E3.95 iJUl tUVlVVMhC KU.,LUNUUIM 1 bN 01-724-3564. OPEN 9-6, MOIM-SAT. TEST EQU fPMENT • SPEAKERS •COMPONENTS MICROPHONES# MODULES • fC'S • DISPLAYS CALL IN AND SEE FOR YOURSELF. £1.65 Fin in £4 l-u £ 1.67 £1.65 £1.67 WEL LFR 6Z00D PK Dual nr.ii fin k ■ ■. ■ K lH % U i : 55n. 2 ! \3D&5A 1 1?fJ un 1 J aiT^I Fui'v ouii'anrRRd $ lor £1. Id'Or . £3. - t DO '(Jr £77 50 l ODD hit f 750. - ALSO IN STOCK 1 CM square s linn relit 'A ‘.-r> - !j VIA ii l(n E2.W3 CtMHjr/imi Mu; . capsules hi 1 1’ V I p e-amp . FI KD 600mV. ql^rs -m ail. LOR . Cl z5 LEI I 7g..L°1 1 VI r.1 1 , 1 1 n r r | AM.'FM i r i rj d l. I K I rpcq and IF strip £7.3fl tJ.Urs Rich ptnt 1 b'ue 1 red .... i"J printer! -ti.ii k lr. F.2.RD 1 1| i * -r-. : Transgocf-rs f*7.5Q , ;. pap seiektor elektor January 1978 — 1-01 Artificial pancreas? Although roughly 25% of the popu- lation are susceptible to diabetes, only 3% actually contract this much feared illness; of these diabetics approximately 30% (he, roughly 1% of the total popu- lation) are dependent upon insulin (diabetes m el lit us), whilst the remaining 70% can be treated by means of a care- fully regulated diet and, in some cases, by pills which reduce the level of sugar in the blood . The illness is caused by a failure of the pancreas to produce sufficient insulin, a substance which is of crucial import- ance for carbohydrate metabolism. Lack of insulin diminishes the ability of the muscles and other tissues to utilise sugar for the purposes of nutrition, so that the sugar simply builds up in the blood- stream until it is excreted in the urine. What actually precipitates the onset of the disease is not yet fully understood, although it is known that the cells in the pancreas called the islets of Langerhans cease to function. Since insulin is no longer provided by the pancreas, it therefore becomes necessary for the diabetic to administer the insulin to himself by means of injections. It is possible to recognise the diabetic by the set times during the day at which he eats, and also by the small packet of sugar or glucose sweets which he always carries. The reason for this latter pre- caution is that in addition to suffering from too much sugar in his blood, be. from a shortage of insulin, the second dangerous state for a diabetic is to have too much insulin - a condition which, unless remedied by an intake of sugar, can quickly lead to the patient falling into a coma. The discovery in 1922 by the Canadians Banting and Bert that insulin could be isolated and extracted from the pancreas of pigs and cattle meant that diabetes was no longer a fatal illness. Since then types of insulin have been developed w r hich have reduced the number of injections needed from between three to six a day to just one a day. However, scientists have continued to seek a system which would automatically monitor the blood-sugar level of the diabetic and regulate the amount of insulin being administered; in other words, produce an 'artificial pane re ash At present the concentration of sugar in the blood of a diabetic is ascertained by means of blood samples which change the colour of certain chemicals, the particular colour indicating the amount of sugar being carried by the blood- stream to the muscles and other tissues. Since it is rather unpleasant for the patient to have to take repeated samples of blood, the quantity of sugar present in the urine can also serve as a suitable measure. Thus until now the diabetic has attempted to balance the relative amounts of sugar and insulin by means of injections and a strictly controlled diet, whilst regularly checking his blood - sugar level by means of the above men- tioned blood and urine tests. However, these methods as well as being somewhat awkward and unpleasant for the patient are also fairly approximate, as is evinced by the imbalance found in the metabolism of large numbers of dia- betics. Fortunately, research into the treatment of the diabetic is continuing, and recently several different teams of researchers have come up with develop- ments which, taken together, could prove a first step on the road to an artificial pancreas. Automatic insulin drip At Siemens Erlangen (Germany) a re- search team working under Dr. Manfred Franetzky have developed a miniature ‘pump*, which allows very small amounts of highly concentrated insulin to be fed continuously into the blood- stream of the diabetic (see figures 2a to 2c). The pump is scarcely larger than a matchbox, and weighs approximately 120 grammes when filled with sufficient insulin for nine months. The amount of fluid which is released can be varied between 0,1 fi\ and 10 {j\ per hour. The insulin requirements of the diabetic vary throughout the day, showing strong peaks around 9.00 and 14.00. It appears from experiments which have been carried out that, although the needs of the patient vary considerably in the course of the day, each diabetic nonetheless has his own particular 'programme 1 of requirements. As Jong as an implantable sensor which w r ou!d monitor the changes in the diabetic's sugar level remains unavailable, the possibility of a pre-programmed insulin dosage (using e.g. a clock 1C) is being investigated. Figure 1. Dr, Kaiser demonstrates the use of the laser absorption spectroscope, which can be used to measure the levels of various organic compounds tin the bloodstream, for example polypeptides urea, cholesterin and, of course, glucose. Although having to 'kiss' the test prism may seem a strange procedure, it is considerably less unpleasant than having a blood sample taken, and the measurement is 1000 times more accurate. The measurement could also be made many times a day. Figure 2a, The insulin dosage pump from Siemens, This is able to deliver extremely accurately controlled amounts of insulin, and has a reservoir containing up to nine months' supp ly. • III- I: : |||fi |j|| V- W: T: ' " ...... *!: : v : ■ ; is > •• ;; : | - msm selekto 1-Q2 — elektor January 1978 Figure 2b, The insulin pump and control unit. Since a diabetic's insulin requirements vary throughout the day, the possibility of a clock- controlled dosage programme is being considered. Figure 2c, Rear view of the control unit. Figure 3. Cutaway diagram of the glucose electrode developed by Laine, Schultz, Thomas, Sayler and Bergmann, Figure 4, Absorption spectra of various compounds found in the bloodstream. polypropylene tapfc membrane lead wire anode silver wire cathode perspex case signal wires glucose oxidase matrix signal wires 98 ?Q 2 Blood-sugar monitor A great deal of excitement was gener- ated in the field of diabetic research last year by two breakthroughs in the moni- toring of blood-sugar levels. A Japanese research group announced the develop- ment of an 'enzyme electrode 7 for direct monitoring of sucrose levels in the bloodstream and an American group announced the development of a 'glu- cose electrode'. This latter is of particular interest, since it is fairly compact (see figure 3), produces an output voltage proportional to the glucose level in the bloodstream, and at first sight would appear to be ideal for implantation. Unfortunately, the electrode is unsuitable for long-term use, since, when placed in the blood- stream, it quickly becomes choked with tissue shed from the blood-vessel walls, and also encourages the formation of blood clots. Laser absorption spectroscopy Another interesting approach to monitoring the level of glucose (and other compounds) in the blood has been developed by Dr r Kaiser of the Max Planck Institute in Munich, Laser absorption spectroscopy utilises the principle that compounds selectively absorb light of a particular wavelength. It is therefore possible to identify and quantify specific compounds, such as glucose, in the blood, bv measuring the absorption of light of the appropriate wavelength. Figure 4 shows the absorption spectra for various compounds in the blood. The absorption wavelength ol glucose is around 10 microns. This wavelength can conveniently be provided by a carbon- dioxide laser, which covers the range 9,15 to 10.9 microns. The interesting thing about this technique is that it does not require a blood sample to be taken. The absorption measurement can be- taken through the skin at any point where the blood flow' is near the surface and fairly rapid. The lips are particularly suitable and to take a measurement the subject merely presses his- lips against l m mw ww li m m3 m w -f m * xi: F-ffi x-Sj P ii:;:;!: III ■ |if! ■i- 7 >:■ -■ Pil pi 4-M. ^ ;P :il l-llpj : : ;i;i: ■: : x :i K«.x|:i ■ !& # ;: ^ " >: 4 ¥ i--*« " . :x #7 0 4 x ^ * ' ■ f 1 1 1 II S - i, -5: -y - x: * . -i * x ' . aflMji' ri ii P r — « -i|x " i®?' ' ... .K ii? - i.i m m m si - I ■ Si? .4 **•#.«%£ :|§ :'§> fl M ■ ***•#: ... :Si . m •:?; # ^ 7^ ^ |j|' M f| #• • ... 4#.*+ : ii*- ^ ^ "V ’ * k 4. -Sf ' :i;. 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' ..-X4 |. x 4 Px ■ :..y \,y.. tig.'.*:*'- : : : !;. ::: .. XX, •# Sr • 4- X. xx: :7i : iiix-xp .... x xi: i.i : : :: i:iii ix : •# • ‘ : '' C . ;4:f v ! •’: ' ••• ' ' . . " ■ « ' :.. rM’ ■ ■■ - : - ■■ x ... jv. . /Sjv •■■■■■ ’ i. -.v.-.-j- . .■.v.'-.v.vX^ m- -m -X •»«- : . ■ :::: XvX :¥iv: : :x:.v. . ■:¥:¥ ' ' vcoYi-i-iSi-'-" ” • - ■ w •• - ,... .... ,w;y-. ■ i ..... W V" . : • • ' .. >•: x- ^ ... .. vx>* ‘.x* wm . X selektor elektor january 1978 — 1-03 the test prism of the spectroscope. Dr, Kaiser is shown demonstrating this technique in figure 1 . Figure 5 shows the arrangement of the spectroscope. Infra-red light from the laser is split, by a sc mi -silvered mirror, into two beams which are passed through two prisms, a reference prism and the test prism. The beams are then recombined and focused onto a photo- detector. The two beams are alternately interrupted by a mechanical chopper, which allows comparison of the light passing through the two prisms. When the lips are pressed against the test prism, absorption of the laser light by glucose in the bloodstream occurs. Comparison of the test and reference light beams by the potosensor allows the glucose level to be measured. Apart from its use in the treatment of diabetes, the laser absorption spectro- scope allows many other compounds in the bloodstream to be measured simply by choosing the appropriate laser wave- length, for example polypeptides, eholestcrin and urea. The onset of dis- eases characterised by the appearance, in the bloodstream, of specific compounds could thus be detected at an early stage by a routine test. The necessity for an expensive and time- consuming chemical analysis of a blood sample would be obviated. The laser absorption spectroscope can also differentiate between glucose and ethanol, which was previously difficult because of their very similar absorption wavelengths (see figures 6a and 6b). Ho w ever Dr. Kaiser's method effectively gives a much more 'expanded' wave- length scale (figure 6c) which allows distinction between the two com- pounds. The laser absorption spectro- scope may therefore also find application in drunken driving cases, as an alternative to the controversial blood sample and the possibly unreliable urine test. Artificial pancreas The ultimate result of all this research must be to produce an 'artificial pancreas' w r hich can be implanted in the body. This would free the insulin- dependent diabetic from the worry of administering his own insulin injections and would also tailor insulin dosage much more closely to the body’s needs. The laser absorption spectroscope and the insulin pump are both valuable steps in this direction. Speculating on future developments, the next possibility might be an implantable insulin pump with several years' supply of insulin, fitted with a microprocessor based control unit. This would initially be programmed to deliver insulin in accordance with calculations of the user’s insulin demand based on blood- sugar measurements taken over a period of time. After implantation, the programme could be updated in accordance with blood -sugar measurements obtained daily using a laser absorption spectroscope, the data being communicated to the implanted pump unit by radio or induction loop transm it ter. A reliable power source for the unit would be a major problem, but modern developments in long-life chemical batteries and implantable nuclear power sources may provide a solution. Alternatively, it might be possible to equip the pump unit with rechargeable batteries, which could be charged each night using a coil placed against the chest wall to induce current in a pickup coil inside the implanted unit. The ultimate ideal would be to have an implanted blood -sugar monitor, so that this information couid be ted continu- ously to the pump control unit. However, at this stage it is not clear how this w r ould be accomplished, since the glucose electrode sensor lacks long term reliability, and the laser absorption spectroscope is both complex and bulky. Acknowledgements It is an established fact that the number of scientific fields in which electronics is providing valuable assistance is continually growing. This article is intended to introduce our readers to an area of research which may stimulate the interest and perhaps even the ingenuity of the electronics engineer. For their cooperation in providing information for this article we wish to thank: Dr, Ing. Manfred Franetzky, Siemens AG., Frlangen Prof* Dr + K.D. Hepp, Med. University, Munich Dr. Nils Kaiser, Max-Planck Institute, Munich (Garching) 4 Wave length (microns) 1-04 — elektor January 1978 & Figure 5. Arrangement of the laser absorption spectroscope developed by Dr t Kaiser. Figure 6. The laser absorption spectroscope allows extremely fine wavelength discrimi- nation to be achieved, which can, for example allow glucose to be distinguished from ethanol, even though their absorption wave- lengths are similar. selektor References : ‘Continuous Extracorporal Monitoring of Animat Blood Using the Glucose Electrode’, Laine, Schultz, Thomas, Say ter and Bergmann, Los A ngeles Diabetes, Febr 1 9 76, vol 25, no. 2 pp. 81-89 ‘Laser Absorption Spectroscopy with an A TR -Prism \ Nils Kaiser, Congres Diabe to logic in Geneva Sept. 1977 Enzyme Electrode for Sucrose Ikuo Satoh, Isao Karube and Suichi Suzuki, Biotechnology and Bioengineering, vol. Ill, 1976, pp. 269-272. M Tuning Electro- . Rower stabilisation facility C0-> laser mech, Q-switch CMOS noise generator The generation of noise using digital techniques was discussed extensively in the January 1977 article, and for a complete theoretical treatment of the subject readers are referred to this issue, lhe National Semiconductor digital noise generator IC is a complete pseudo- random binary sequence generator on a single chip. It contains a clock pulse generator, 17-bit shift register, ex clusive- OR feedback and anti-latch-up gating. The pseudo-random noise output of the MM 5837 has a cycle time of 131,071 dock periods. The clock frequency of the MM 5837 is not externally programmable, but may lie between 55 kHz and 1 19 kHz at a supply voltage of 14 to 15 V. Ihis means that the pseudo-random sequence cycle lime may lie between 2,4 s and 1,1 s. and the 3 dB point of Lhe power density spectrum may lie between 24 kHz and 5b kHz. The clock fre- quency is, moreover, extremely depen- dent. on supply voltage. However, with a 15 V supply, the clock frequency tolerance means that the spectral density of the noise output will be between 1.2 and 2.4 lines per Hertz, which is certainly high enough tor the noise spectrum to be considered as continuous for all practical purposes. Noise generator circuit Figure 1 shows the complete circuit of a noise generator. This consists of the MM 5837 noise generator 1C, a 3 dB/ octave *pink noise 5 filter, and a bandpass filter with fixed, third-octave bandwidth and adjustable centre frequency. The TIL version of the noise generator did not include these two filters, bul readers who already have the T 1 L cir- cuit can add the filters simply by m it ting 1CT and feeding the output f the TTL circuit into point Ah Tor leaders building a noise generator from scratch 1 however, it is rec- m mended that the MM 583 7 IC be used. Pink noise filter As mentioned in the January 1977 Just twelve months ago, issue 21 contained an article on the digital synthesis of noise using a pseudo- random binary sequence generator. The circuit used 10 TTL ICs and soaked up some 260 mA from the power supply. Now, inevitably, there is a single IC that does the same job, consumes only one-fourteenth of the power, gives three times the output signal and occupies about one-twentieth of the volume. article, if white noise is fed to a band- pass filter with a constant Q-f actor (constant octave bandwidth) then as the centre frequency of the filter is in- creased the RMS output voltage will increase at 3 dB,' octave. It, as is often the case, third -octave filters are used to make selective frequency response measurements on a system with a white noise input, this 3 dB/octave rise can be a nuisance, since a system with a Hal. frequency response would apparently have a +3 dB/octave slope when measured in this way. The solution is to use noise whose output amplitude falls at 3 dB/octave, to compensate for this effect — so- called ‘pink noiseb This is achieved by feeding the (approximately) white noise output of the MM 5837 to a passive, low pass filter with a slope of —3 dB/ octave. Since ■normal 5 fillers usually have slopes in multiples of 6 dB/octave, a 3 dB/octave filter must be approxi- mated in 'staircase fashion using a series of 6 dB/octave filters with dif- ferent centre frequencies, R1 to RIO and C2 to C\2 in figure 1. Of course, the mathematical analysis of the filter is rather more complex than this simple explanation would indicate! The output of the 3 dB/octave filter is buffered by an emitter follower 1 1 , the output of which is fed to an oper- ational amplifier with a gain of 10, IC2. The pink noise output A2, from the wiper of PI , has sufficient amplitude to drive most audio equipment. Third octave filter Feeding pink noise through a filter having a bandwidth of one-third ol an octave gives an output signal known, not surprisingly, as third-octave noise. An ideal third-octave response is shown in figure 2a. Within the passband there is no attenuation of the signal, and outside the passband there is infinite attenuation. In practice, of course, this passband is impossible to achieve, filters with infinite slopes simply do . not exist. A good approximation to an ideal bandpass response is illustrated in figure 2b. This is achieved by using two selective filters with slightly different 1,06 — elektor January 1978 CMOS noise generator 3 m > < ^ _ tr a 3k 3 H 00 60 itofn ■1 7Q a 2 >■ ■ o 330il DO ' UJ * TJ n i ■ | * _ in. (_] 1 I T 1UK » 3}W C^T3 ^ _E) Cfj + + N> -J < < cr> tji Parts list Resistors: R 1 , R 1 5 , R 1 8 , R 1 9 , R 21 r R23 “10 k R2, R 1 1 = 6k8 R3 h R 1 2 - 4k7 R4 = 3k3 R5 = 2k2 R6 = 1 k5 R7 = 1 k R8 = 680 H R9 = 470 il R10 - 330 n ft 1 3 r R 1 4 - 47 k R16 = 68 k R17 = Sk2 R20,R22 - 39 SI PI = potentiometer 47 k (50 k) log P2a/P2b - stereo potentiometer 10 k log Capacitors: Cl* - 100 ju/25 V C2 = 330 n C3 - 220 n C4 ~ 1 50 n C5,C1 3,C14 r C1 7 r C1 8. Cl 9 = 100 n C6 - 68 n C7 = 47 n C8 - 33 n C9 - 22 n CIO = 15n Cl 1 = 2 m2/63 V C12 * 10 n Cl 5,C1 6 = 1n5 I- _ J L0 PJ -xj — %_ f Q IH V Figure 1. Circuit of the new pseudorandom noise generator. IC1 replaces the 10 TTL ICs of the January 1977 circuit! However readers already possessing the TTL version can add the 3 dB/octave and third-octave filters by omitting I C 1 and connecting the output of the TTL version to point A1, Figure 2a. Ideal third -octave bandpass re- sponse Figure 2b. A good approximation to an ideal response can be obtained using two cascaded selective filters with staggered centre fre- quencies. However, it is not easy to build a filter with continuously variable centre fre- quency using this technique. Figure 2c. The third-octave filter used in the noise generator is a single selective filter with a Q of 4.3 2 h which is realised using a state- variable filter configuration. Figure 3. Printed circuit board and com- ponent layout for the noise generator { EPS 9859). Sem i conductors . T1 = BC 547B.BC 107B, BC 147B or equivalent IC1 * = MM 5837 (National Semiconductor) IC2 - 741 (MINI DIP: EC3a + JC3b + ICS c + IC3d - - X ft 421 2 CP (E XA R ) quad op-amp * Is omitted if the ttl circuit in E lektor 21 , January J 77 is used. centre frequency. At the band edges their combined slopes arc quite large, whilst within the passband the response is relatively flat. Unfortunately, the filter in the noise generator must have a variable centre frequency, which is difficult to achieve with staggered selective filters they cannot be made to track with sufficient accuracy. A compromise solution is therefore adopted in the form of a single selective filter with a Q-f actor of 4.32 (see figure 2c). The filter circuit used is the so-called state-variable filter. The pink noise signal from t.hc wiper of PI is fed via a voltage-follower buffer IC3a to the input of lC3b. The state -variable filter is built around 1C' 3c and IC3d; by means of a ganged potentiometer P2, the lime constants of the integrators and hence the centre frequency of the filtei may- be varied. The Q-f actor of the filter is determined by feedback to IC3b and is independent of the filter centre fre- quency. This type of filter is extremely CMOS noise generator elektor January 197S — 1-07 2ESHIN y SI 41 stable over its entire frequency range, which may be adjusted from 40 Hz to 10 kHz approximately, by P2, Hie third-octave noise output is available at point A3, the output of 1C 3c. Construction The noise generator and its associated filters are accommodated on a very compact printed circuit board, the track pattern and component layout of which are given in figure 3. If the filter circuits are to be used with the existing TTL noise generator then 1C! and C l can be omitted, and input A! is connected direct to the output of N30 on the TTL noise generator board. If the white noise output ot the TTL r. :se generator is not required, which frequently il is not, then several niodi- fiw jt.ons can be made with advantage, firstly . R5, K6 , C2 and C3 of the TTL circuit may be omitted, and the clock frequency can be lowered to about 87 kHz by increasing Cl to 27 m This has the effect of increasing the spectral density to about 12 lines per Hertz, which makes the output an even better approximation to a continuous noise spectrum. A white noise output from the MM 5837 is not provided, for the simple reason that this 1C cannot provide white noise over the entire audio spectrum, since the clock fre- quency is too low. At first sight this may seem a little odd. The basic en- ter ion is that the clock frequency should be 2.2 times greater than the highest required noise frequency. With a worst case clock frequency of 55 kHz it would appear that the MM 5837 satisfies this condition for a noise bandwidth from 0 to 20 kHz, However the noise must also be truly Gaussian, and this criterion is influenced by the _ noise waveform. With the rectangular pulse waveform produced by the pseudo-radom binary sequence gener- ator the clock frequency must be at least 20 times the highest desired noise frequency, a condition which the MM 5837 cannot satisfy for the entire audio spectrum. The MM 5837 will produce Gaussian noise 'only up to a few kHz. M Stop Press Further development work on the noise generator has revealed that the 3 dBf octave filter can be considerably simplified by optimising the circuit parameters* The modifications also double the noise output signal from the filter. The following component changes are necessarv : Component New value R 1 - 6k8 R2 = 3 k = 1k8+ 1k2 R3 - 1 k R 4 = 300 a = i so n + 1 20 n 02 -1m C3 - 270 n G4 = 94 n = 2 x 47 n Cl 2 = 33 n R5 to R10 end C5 to Cl 0 are omitted. Any assessment of the ‘musicality 5 of an audio system must be purely subjective, since quantitative measurements do not always correlate with subjective experi- ence, Users of hi-fi equipment (naturally enough) often set the amplifier tone controls to give a sound that is most pleasing to their cars, which may not agree with other people's ideas of the best sound. The same is true when an equaliser is used. Although in theory the intention is to produce a flat frequency response, most users will adjust the response by ear, and even if measuring equipment were used to set up a flat response this would not necessarily give the most pleasing sound. Nevertheless, an equaliser can provide a degree of control over frequency response that is impossible with conven- tional tone controls, as will become apparent. Normal tone controls, such as the Baxandall type, consist of a combi- nation of high and low pass filters of the form show r n in figure la. These operate at the low frequency (bass) and high frequency (treble) ends of the audio spectrum, and have little effect in the middle of the audio range, by varying the impedances Z L and Z 2 the filters may be made to either boost or cut. the bass and treble relative to the mid-range. The controls may be of the fixed slope type with variable break frequency as in figure lb or. more commonly, of the variable slope type with fixed break frequency, as shown in figure lc. Although these circuits offer an inex- pensive form of tone control, they have several disadvantages. Firstly, they oper- ate only at the extremes of the audio spectrum, and secondly, since each con- trol is effective over a wide frequency range, boosting or cutting of narrow bands of frequencies is not possible. The alternative Figure 2 illustrates the principle of an equaliser in which the audio spectrum is divided into a number of sub-spectra. The signals within any sub -spectrum may be boosted or cuL by up to A dB. f’i and fj represent the band edges of any sub -spectrum, and the centre fre- quency f 0 is the geometric mean of these two frequencies. The bandwidth B An equaliser consists of a number of tone controls, each covering a specific portion of the audio spectrum. The object is to use these controls to obtain a flat frequency response of the audio reproducing chain. This article discusses the theory and practice of an equaliser using eight controls. Inductors are used in the filters, however these are not wound components but are simulated electronically* is the difference between f a and , On the logarithmic frequency scale shown in figure 2 each sub-spectrum occupies an equal bandwidth of one- third of an octave, i.e. for each sub- spectrum fi = 0.8906 fo, f 2 = 1 .22.5 f 0 and B = 0.23 ] 8 f 0 . A compromise The foregoing represents an almost ideal type of filter offering precise control over the entire audio spectrum, but in practice it would be difficult and expensive to realise. To begin with, no less than thirty controls per channel would be required. Even then, some compromise would be required: filters with infinitely sharp cutoff at the band edges are impossible to realise even in theory — let alone in practice! Filters with very sharp cutoffs are practicable, but require a large number of com- ponents. Some sort of compromise is therefore necessary in a practical design where cost and complexity must be considered. The first step is to reduce the number of frequency bands (and hence the number of controls) to something more practical than thirty. Figure 3 shows ihe frequency bands covered by the filters used in the Elektor equaliser. The audio spectrum is split up into eight one- octave bands from 44.6 Hz to 11.3 kHz, with no filrers being provided above or below ? this frequency. At first sight it would seem that no control is being exerted over an important part of the audio spectrum. However, since the filters do not have the ideal steep-sided passband characteristic the control response does extend below 44.6 IIz and above 11.3 kHz. In addition, if bass or treble boost or cut is required at the extreme ends ot the spectrum it can be applied by a good Baxandall type of tone control before switching in the equaliser. The second compromise is the use of resonant circuits in the filters. Figures 4 and 5 show the response of two filters set for various degrees of boost and cut, with the ideal rectangular response superimposed The difference between the two is that the filter of figure 4 has a higher Q than that of figure 5, It is evident that, towards the edges of the elector equaliser elektor January 1978 — 1-09 Figure la. The Baxandall tone control net- work consists of an amplifier with frequency- dependent impedances in the input and feed- back loops. When these are balanced the fre- quency response is flat, but when they are unbalanced by altering one of the tone con- trol potentiometers, boost or cut occurs. Figures 1b and 1c. Baxandall tone control networks can either have variable turnover frequency and fixed slope (figure 1b), or variable slope and fixed turnover frequency; but in either event they control only the extremes (bass and treble ends) of the spec- trum. Figure 2. Frequency response of an 'ideal' equaliser, which divides up the audio spec- trum into 30 third-octave sub-spectra. The gain of the system within each sub-spectrum can be varied between +A dB and — AdB, Figure 3. As figure 2, but now with a coarser division into 8 one-octave sub-spectra. The concept of the Elektor equaliser is based on this division. 9-032 1b *lc 3? MB) 0832 it CN ia t- ;n w i cn o me tD CO LO O O Q If} CM {Q O LC ?- T- T- CN fN cn o o on o co C SJ in lo lo CNtD UD J£ .M -l£ ^ ^ ^ O CM CO Q 5832 2 9332 3 t“10 - elektor january 1978 elektor equaliser passband, the response of the high-Q filter is several dR down on l he ideal rectangular response, whereas that of the low-Q filter is not. On the other hand, the response of the high-Q filter is also a long way down where it over- laps into the passband of adjacent fil- ters, whereas the response of the low-Q filter is only a few dR down, and will thus interact more with adjacent fil- ters. This is illustrated in figures 6 and 7, which respectively show two adjacent high-Q and two adjacent low-Q filters with various combinations of boost and cut. The individual responses of the two filters at full boost and full cut are shown, together with the combined response of the filters for all possible combinations of boost and cut., as fol- lows: Curve Mum her Description filter f 0 i , full boost filter f 02 ? full boost filter f oi > full cut filter fo 2 , full cut combined response, lull boost combined response, full cut combined response, foi boost, f 02 cut combined response, for cut, fo 2 boost. fa f s 983: 4 The question arises as to which type of filter response should be chosen for the equaliser, low-Q, high-Q, or a compro- mise between the two? In practice it appears that an equaliser built using lower Q filters is judged to be more ‘musical* by a majority of listeners. Filter principles Having decided on the type of response required from the filter, the next step is to decide how to achieve it. Figure 8a shows the basic circuit of one filter section, L and C form a senes resonant elektor equaliser elektor January 1978 — 1-11 circuit* With the slider of R1 in the extreme left-hand position the op-amp functions as a voltage follower. Assuming the op-amp has sufficient open-loop gain t h c e f foot o f R 1 h e t w een the two inputs is negligible since very little voltage appears across it. The input resistor R and the resonant circuit form a bandstop filter. Within its passband the impedance of the resonant circuit is low, so the signal appearing at the non- inverting input of the op-amp, and hence at the output, will be severely at.t enuated. With the slider of R1 in the extreme right hand position the op-amp func- tions as a non-inverting amplifier with a negative feedback network comprising feedback resistor R and the resonant circuit. The impedance of the series resonant circuit within its passband is low, so the gain of the amplifier is high and the signal is boosted. Outside the passband the impedance of the LC circuit is high, so whatever the selling of R1 the op-amp simply functions as a voltage follower and the signal is neither amplified nor attenuated. With Rl in its m id -posit ion the attenu- ation introduced by the bandstop filter action is exactly cancelled at all fre- quencies by the bandpass filler action, since the resonant circuit has an equal effect on both the signal path and the feedback path. The gain with Rl in the mid-position is thus OdB al alt fre- quencies. Having explained the basic principle of the filter the filler parameters may now be calculated. Potentiometer Rl and the series resonant circuit may be rep- resented by a star network of three impedances Z\ to Z3 (figure 8b). To simplify the analysis the input and out- put circuits can be separated by trans- forming this circuit into its equivalent delta circuit as shown in figure 8c. This allows the following equations to be derived : Figures 4 and 5* Since the ideal rectangular passband is impossible to realise, an approxi- mation must be made using resonant circuits. These can either be high^Q, as in figure 4 r or low-Q, as in figure 5. Figures 6 and 7. Showing how adjacent filters interact. The bigh-G filters (figure 61 have a much smaller 3 dB bandwidth and so interact less with the passbands of adjacent filters than do the low-Q filters (figure?}. Although it would thus appear that high-Q filters give better control with less interaction, low-Q filters are preferred from a musical point of view. 1-12 — elektor january 1978 elektor equalise (1 - x 1 } 2 + (1 - X 2 ) 2 + K? x 2 KS x 2 with x - — T to ' o = _L /P Q R e V C ’ c$R, + R e + 0K and K .2 = ocjSR i + R e + oR where a and j3 are functions of the slider position of Rl as shown in figure Ha; a - \ — j3. The gain at the centre frequency (f= I'o, x = 1 ) is K 1 /K 2 which is dependent on the slider position of Rl. The gain at fo may be varied between R + R e r — _ and .... . R e R + Re’ he* between phis and minus AdB in figures 4 and 5. Once suitable values for A and the 3 dB points of the filter response have been chosen the required Q can be derived. For the practical cir- cuit maximum boost and cut of 12 dB were chosen, and the --3 dB points were made to coincide with the ‘ideal' band edges, as shown in figure 5. Having decided this the relative values of R and K e can be determined and the Q calcu- Figure 8a. The basic fitter section of the equaliser, which will provide boost or cut depending on the setting of the pot. Perform- ing a star-delta transformation on the circuit {figures 8b and 8c) makes analysis easier. Figure9. Basic circuit of a simulated induc- tor. Figure 10. This circuit is equivalent to the cir- cuit of figure 9. The synthesised inductor has a number of parasitic elements associated with it. However, if k is 1 the shunt resistance becomes infinite, R e can be used to determine the Q of the resonant circuit, and the effect of Cg can be made negligible by making R g very large. Figure 11. The effect of C g and R g can be eliminated entirely by using a buffer stage, but this is not realty necessary in this appli- cation . Figure 12, Complete circuit of the equaliser. lated. For a maximum gain of 1 2 d B and — 3 dB points as shown in figure 5 a Q of 1.5118 is necessary. Inductors — wound or simulated? It is worth considering at this point it the inductors used in the series resonant circuits should be conventional wound components. A few quick calculations using ihe fore- going equations show that, for the low frequency filter sections inductances greater than l Henry are necessary if practicable values for R and R c are to be maintained. Such values can only be achieved conventionally by using large ferrite pot cores, which are bulky and expensive, to say nothing of the tedious p r o cess o t winding the coils. It is thus cheaper and more convenient to synthesise the required inductors electronically. Figure 4 shows the cir- cuit of an electronic inductor eon- sistmg of a voltage follower, two resistors and a capacitor. Figure 10 shows the equivalent circuit of this arrangement, with the synthesised inductance shown at the bottom right of the diagram El can be seen ihai two parasitic networks appear in parallel with the inductor. The first of these i> a resistor Re Mere of the op-amp. If k is made equal to \ (which it is. since the op-amp is con- nected as a collage follower) then the value of tins resistor becomes infinite, and it cannot affect I he inductor. The second parasitic network consists ol Rg and Cg 7 and the effect of this can be reduced by making large elektor January 1978 — 1-13 elektor equaliser 2x BC bb7B I ■ V3r OO-lh- 2x BC&B7B ^3 PC549C BCB49C SC 549 C 20mA ICIa + 1C 1 b + Idc+lCId IC1 = XR4212CP IC2a- IC2b+ IC2C+ IC2d = IC2 = XR4212CP “ © I 100m -r © ] lOOn T compared to R e . The circuit then effec- tively becomes an inductor in series with a resistor R e , which determines the Q when the inductor is connected into a series resonant circuit. 1 1 would be possible to eliminate the shunting effect of Rg and Og entirely, by inter- posing a butter, as shown in figure 1 i , hut for the purposes of the equaliser this is unnecessary. Performance of synthesised inductors Conventional inductors produce negli- gible noise, and the distortion that they introduce is due mainly to saturation of the core material, which is negligible at moderate signal levels, Flee Ironic induc- tors, on the other hand, are subject to certain limitations. Noise and distortion are introduced by the operational ampli- : :er. and the voltage that can appear across the inductor is limited by the , pping level of the op-amp. Harmonic distortion can be reduced by ensuring that the voltage follower has good linearity, i.e. the gain remains . -list ant at unity for all output levels. T h is me a n > c hoosi ng a n o p-a m p with a a output resistance, which is further reduced by t h e 1 0 0'3 n eg a I i v e feedback _■ is applied. As most op-amps oper- ... w a l h a c lass A - B output stage, er< > ss- ici distortion can be a problem, so it ■ essential to operate the op-amp into r'y high impedance load so that it n class A . A ■ refinance the voltages across the . t and across the (synthesised) induct ■: in the resonant circuit are in antiphase and are both equal to Q times the voltage applied across the circuit, so it is essential that these voltages do no l exceed the ouipul range of the op-amp. It is thus necessary to compromise between operating at high signal levels to obtain a good signal-to- noise ratio, and operating at low signal levels to avoid clipping. Fortunately, the Q of Ihe circuits used in the equaliser is low , so this is not too much ol a problem. With the foregoing criteria in mind the inductance values and senes capacitors of the resonant circuits can be calcu- lated using the following equations: f 2 _ lo - 2 __ 1-' ; 0 - R e C 47t"lc; L - R e RgCg (k - 1) from which follow: 1 „ Q ( - : C it — 2jrf 0 QRe ’ % 2jri 0 Rg In the circuit described here we will use: R g - 82 k; R e - 1 k;Q= 2. Furthermore, 1 01 “ 63 11/ 1 02 = 125 H/. f 0 3 = 250 H/ fo 4 = 500 Hz I os ^ 1 kH / fo6 = 2 kHz to? = 4 kHz fog = 8 kHz The correct values for C and Cg can now be calculated. The circuit Figure 12 shows the complete circuit of the equaliser. For acceptable audio per- formance the op-amp associated with each filter section according to figure 8a must be a high-quality, low-distortion, low-noise amplifier, which means that discrete components are used in prefer- ence to monolithic its. It would be p ro liibit i vel y expensive lo p rov i d e a separate amplifier for each filter section (quite apart from the problem in main- taining a satisfactory S/IN ratio), so several filter sections share one ampli- fier. This could cause problems due lo interactions between individual filter characteristics if filter sections with adjacent passbands were connected to the same amplifier, so filter sections f oi > fm, I os, fo 7 share the amplifier comprising T2> 13 and T4, while filter sections foe > to* , fos , foa share ihe amplifier consisting of T5, T6 and T7 . (Note that this interaction between individual filter characteristics is an entirely separate effect from the com- bined filter response of two overlapping passbands mentioned earlier.) The input signal is fed first to an emit- ter-follower T 1 , then through the tw r o cascaded filter amplifiers T2 to T4 and T5 to T7. The resemblance between these amplifiers and the theoretical op- amp shown in figure 8 may not be immediately apparent, but it becomes more so once it is realised that the base of T2 functions as the non-inverting input and its emitter as the inverting input., 13 and T4 arc connected as a Darlington pair with a very high gain. The same comments apply to T5 ... T7. Kach resonant circuit consists of a simu- lated inductor built around one-quarter of a quad op -amp, together with a series capacitor. For example the 63 Hz resonant circuit consists of a simulated 1-14 — etektor january 1978 elektar equalise Figure 13. Modifications to the basic filter section of the equaliser, if 'Baxandaif type bass and treble controls are required. Figure 14. Frequency response of the bass and treble controls. Frgure 15. Circuit modifications required for adding bass and treble controls. With the values shown, the turnover frequencies (f; and f 3 in figure 14} are 500 Hz and 4 kHz, respectively; the maximum cut or boost is 12 dB + Figures 16 and 17. Printed circuit board and component layout for the equaliser (EPS 9832). 44 [dig inductor comprising ICla, R15 (- R c ), R23 (= R g ), Cl 3 and Cl 4 (= C g ), and a series capacitor C11/C12 (- C}. In most cases the inductor capacitor and series capacitor are made up from two parallel capacitors to obtain the exact value, but this is not necessary in the case of C25/C26 and C39/C40, so C26 and C40 are omitted. Adding tone controls Although the Fdektor equaliser is quite adequate as it stands for incorporation into an existing hi-fi set up (with bass and treble controls), some additional control is required at the top and bot- tom end of the audio spectrum for use in systems without existing tone con- trols, A simple modification allows a ‘Baxandair type of tone control to be added, which will give boost and cut at the very extremes of the audio spec- trum, This facility can be extremely useful, for example to provide some bass lift to compensate for the lack of bass output from bookshelf speakers. Figure 13 shows the modifications to the basic filter section of the equaliser. Comparing this with figure 8 a it can be seen that the series-resonant circuit has been replaced by an LR circuit for the bass control and a CR circuit for the treble control These circuits still give boost or cut depending on the position of or but since a resonant circuit is not used a high and low filter response is obtained, rather than the band filter response obtained with the resonant cir- cuit. The frequency response of the bass and treble controls is shown in figure 14. The turnover frequencies fi to f 4 are determined by the values of L, C, R e and R, in accordance with the equations shown in figure 14. The practical circuit, shown in figure 15, is designed for turnover frequencies f 2 = 500 Hz and i ’3 = 4 kHz. and a maxi- mum boost and cut of 12 dB, which is the same as ihe rest of the equaliser cir- cuit, The treble control is added simply by including an extra potent iometer,P^, between points G and H on the equal- iser board, together with resistor and capacitor C. The bass control, however, requires a simulated inductor, and to obtain this it is necessary to omit the lowest equaliser control, Pi, and to convert the circuit around 1C la for use as the bass control inductor. In other words, PI is now replaced by the 5 k bass control, Cl 1 and Cl 3 are amitted, Cl 2 is replaced by a wire link and C14 becomes 22 n. Construction A printed circuit board and component layout for the equaliser are given in figures 16 and 17. and construction should present no problems. The equal- iser can be built into its own case, or may be incorporated into other equip- ment such as an amplifier system. For stereo operation two printed circuit boards are, of course, required. If the circuit is simply to be used as a room equaliser in one listening room then potentiometers PI to P8 can be presets mounted direct on the printed circuit board, which are adjusted once and then left. If 1 he circuit is to be used in place of a tone control system or to produce special effects then (screened!) connections must be brought out to potentiometers mounted on the front panel. The potentiometers may be either rotary or slider types, but we feel that (he sliders often used in such equal- isers give no real advantage over rotary types, since the ‘graphic’ response which they apparently display is a myth. Furthermore sliders are more expensive, and frequently less reliable than rotary types. With all the potentiometers in their centre posi lions, the equaliser has a gain of unity, and so may be connected into any audio chain without affecting the overall gain. However lor the best compromise between noise and over- load margin (v L max 2 V) it is best to connect the equaliser between the pre- amp and control/ main amplifier where signal levels are ol ihe order of a few hundred millivolts. The tape socket ot an existing amplifier may be used for this purpose. H elektor equaliser elektor january 1978 — 1-15 Parts list C26 is omitted C27 = 560 n C28 = 82 n C29 = 27 n C31 = 1 20 n C32,C35 = 39 n C33 = 6nS C37 = 1 n5 C38,C41 = 470 p C39 = 10 n C40 is omitted C42 - 1 5 p C43 r C44 ,C4 5 ,C46 , C47, C48 = 100 n MKM G49 r C50 - 10 ju/25 V Semico nductors: T1T2J5 - BC549C, BC 109C or equivalent T3,T4 h T 6,T7 = BC 557 B , BC 1 77B or equivalent 4C1JC2 = XR 421 2CP (Exar) Resistors' R1 = 68 k R2,R6,R1 1,R1 5,R16 r R17, R 1 8,R 1 9,R20 r R21 ,R22 = 1 k R3 - 6k8 R4 r R7,R8,R9,R1 2,R1 3 = 3k9 R5.R10- 12 k R14 = 220 k R23 r R24 r R25 r R2G, R 27,R28 r R29,R30 - 82 k PI ... P8 = (preset) potentiometer 10 k lin (see text) rv mi "■ Ml WQ^4 " - t w h|rC 1-16 — eteklor january 1978 TAP-tip electronic open fir? Many circuits for TAPs (Touch Acti- vated Programme switches] have previously been published in Elektor, However, ail of these required the use of two pairs of touch contacts, one to set the TAP to the 'on' position and one to reset it to the 'off 1 position. The novel feature of this circuit is that it requires only one touch contact. Touching the contact once sets the 1 AP; touching it a second time resets the TAP. Nl and N2 form a flip-flop (bistable multivibrator!. Assume that initially the output of N2 is low. The inputs of Nl are also pulled low via R2, so the output of Nl is high. The inputs of N2 are thus high, which satisfies the criterion for the output to be low, which was the original assumption. Ci is charged to logic high through R3 from the high output of Nl. If the touch contacts are now bridged by a finger, the logic high on Cl will be applied to the inputs of Nl through Rt and the skin resist- ance, The output of N 1 will go low, so the output of N2 will go high, holding the inputs of Nl high even if the finger is removed. The TAP Is now set. Once the finger is removed, C I will discharge through R3 into the low out- put of NL If the touch contacts are subsequently bridged, the inputs of Nl will be pulled low by Cl (since it Is now discharged). The output of Nl will thus go high and the output of N2 low. which will hold the inputs of Nl low even after the finger has been removed. The TAP is now reset to its original state. Cl will charge to logic high through R3 from the output of Nl, ready for the contact to be touched again. The only constraint on the operation of the circuit is that the interval between successive operations of the switch must be at least half a second to allow Cl time to charge and discharge. M There is no doubt that most people find the sight of an open fire pleasing. There seems to be something soothing in watching the flames flicker and play about the coals. On the other hand, coal fires are difficult to light, slow in producing beat, and also extremely messy to clean out. For this reason many people prefer the convenience and speed of an electric fire, and reluc- tantly relinquish the pleasures of a hearth fire. Manufacturers of electric fires have realised this fact, and at- tempted to entice the consumer to buy electric by fitting the front of their fires with a coal- or log effect. Unfor- tunately, the lamps which are used to illuminate these fronts sometimes provide only a constant light, thereby considerably diminishing the realism of the effect. However, using only a hand- ful of components from the junk- box, it is possible to construct a small circuit to restore the 1 flicker' in your fire. The way the circuit functions is quite simple. When power is applied capacitor Cl is charged via the lamp, resistor R2 and diode 1)1. After several half cycles of the mains supply, the voltage across this capacitor exceeds the trigger volt- age of diac Di 1 . This diac in turn triggers thryristor Thl, with the result that capacitor C2 charges up rapidly through this thyristor and 1)3, How- ever, when the mains voltage next crosses zero, this thyristor turns off. Capacitor C3, which is part of the trigger circuit of Tri 1 , is now charged rapidly by capacitor C2 via resistor R3. This l DC bias" on C3 decreases as capacitor C2 discharges. This in turn results in a gradual change in the triggering angle of the triac, causing the lamp La to flicker. Once Cl has again reached the trigger voltage of the diac, the entire cycle repeats itself. As far as component values are con- cerned, care should be taken to ensure that the maximum current taken by the triac is at least twice the maximum current drawn by the lamp La. For a normal sized fire a 4 A type should prove sufficient. The triac must also be able to withstand the peak mains volt- age i.e. approx. 400 V, A 400 V { 1 A) thyristor should also prove suitable. Dl Many electric fires are equipped with a special coal- or log-effect to simulate the appearance of an open fire* This effect is sometimes spoilt, however, by the fact that the lamp provides a constant rather than a flickering tight. The circuit described here is intended to remedy that defect. S. Kaul may be any commonly available 600 V rectifier diode. During construction it should be remembered that the full mains voltage may appear across any point in the cir- cuit. For this reason it should be well insulated. H peak programme meter ©Sektor January 1978 — 1-17 peak programme meter A meter for measuring audio signal levels must satisfy several criteria. Firstly, the A.C, signal must be rectified before it can be displayed on a moving coil- or other D.C meter, since large signal peaks can equipment, the meter must be capable of rapid response to signal peaks. In addition, since signal peaks may last too short a time for the meter to he read, the meter must store peaks long enough for the user to read the meter. Finally, since the human ear has a logar- ithmic response, the meter response should also be logarithmic. Block diagram Figure 1 shows a block diagram of the peak programme meter drive circuit, it consists of two stages, a peak rectifier i A) and a logarithmic amplifier (B), with a sensitivity control, PI, between them. The rectifier charges a capacitor to the neak value of the A.C. input signal and the logarithmic amplifier gives an out- put voltage proportional to the logar- ithm of the D.C. voltage on the capaci- tor. 1'his output can be used to drive a moving coil or other meter, which can scaled linearly in dB, Complete circuit The left channel of the PPM drive jircnit is shown in figure 2. The peak rectifier, built around A 1 , rectifies negative half-cycles of the input wave- form. The signal is applied., via Cl arid - 1. to the inverting input of AL The right channel circuit is identical, but components are identified by an apostrophe C). I ndtT quiescent conditions A1 is oper- ating open -loop, since D2 is not forward ■ ased and there is thus no negative :: iback via R4 When the input volt- goes negative the output of A1 vAings rapidly positive until limiting iccurs. D2 conducts and ("2 charges c: d'.\ through D2 and R5. Equilib- reached when the positive volt- on C2 equals the negative input :_ge. when feedback via R4 will reduced the voltage at the inverting nput of A1 to almost /.ero. In the case _f an A.C, input signal, C2 will, of course, charge to a positive voltage equal to the peak negative input voltage. On positive half-cycles of the input signal, the output of A1 swings negative and D2 is reverse-biased. Since there is no negative feedback to the inverting input, 01 is included to limit the maxi- mum positive excursion at this point to 0.6 V, as otherwise the common- mode range of A1 could be exceeded. Since C2 cannot discharge through D2, its only discharge paths are through 1 and R4, which means that the discharge time constant of the peak rectifier is just less than one second. Logarithmic amplifier Extremely accurate logarithmic ampli- fiers can be made by exploiting the exponential collector versus base- emitter voltage characteristic of a transistor. However, this type of logar- ithmic amplifier is unnecessarily com- plex for use in a simpie peak meter circuit, so the approach adopted is to make a "piece wise-linear’ approximation to a logarithmic curve. The principal characteristic of a logar- ithmic amplifier is that the output volt- age increases arithmetically in response to geometric increases in input voltage. To take a simple example, if a 10 mV input gives an output of IV, then ten times this fie. 3 00 mV) would give an output of 2 V and 1 V would give an output of 3 V, etc. An approximation to this type of curve can be achieved by progressively reducing the gain of an op-amp as the input voltage to the amplifier is increased. In figure 2, A 2 has a gain of about t SO at low signal levels. However, once the output level reaches around 4.6 V, D3 conducts, increasing the amount of negative feedback and reducing the gain. At an output voltage of approxi- mately 5.6 V, 1)4 conducts, and at an output voltage of about 8 V, D5 conducts. The gain of A2 is thus pro- gressively reduced as the input signal increases. Of course, the diodes do not conduct abruptly at a particular voltage their dynamic resistance reduces gradually as the voltage increases. This means that the piecewise curve does not have a series of sharp break points, but Secondly, overload Using only one 1C and a few discrete components it is possible to construct a peak programme meter drive circuit that will provide a logarithmically-scaled indication of the peak A.C, input level. The circuit can be used with either LED or moving coil voltmeters to make a compact, two-channel PPM, 1-18 — elektor January 1978 peak programme mete r 1 Table 1, Principal specifications of the PPM drive circuit. Maximum sensitivity : nominal output 10 V DC for 1 50 mV RMS input. Maximum input level: 5 V RMS, Input impedance: ^ 43 k Supply voltage: 12 18 V 118 V absolute maxi- mum!} Current consumption: 30 mA (1 S mA per channel) is relatively smooth, as shown in figure 3. Although this method of producing an approximation to a logarithmic curve is simple and cheap, it does have one or two minor drawbacks, Firstly, due to tolerances in the resistors and diodes used in the circuit, there may be devi- ations from a true logarithmic response. This means that the two channels of the meter may not give the same reading when fed with the same input voltage. However, potentiometers PI and PI allow accurate calibration of the full-scale reading of both channels, so any mismatch will only be apparent at small input levels, where it is not so important. The second drawback of this system is that the meter only has a range of just over 20 dB (a voltage ratio of 10 to 1 ). However, this is comparable with the 23 dB calibrated range of a VU meter, or the 28 dB calibrated range of a BBC PPM, and since the circuit is intended principally for indication of peak signal and overload levels, this relatively small range is not a great disadvantage. If the PPM drive circuit is used with a UAA 180 LED voltmeter, then each of the 12 LEDs represents a step of approximately 2 dB, as shown in figure 4. Construction The use of a 324 quad op-amp allows a two-channel version of the meter drive circuit to he accommodated on a single, compact printed circuit board, the com- ponent layout and track pattern for w'hich are given in figures 5 and 6. The p.c. board is the same size as that for the two-channel UAA 180 LED voltmeter described elsewhere in this issue, so the two boards may be stacked Figure 1. The peak meter drive circuit consists of a rectifier, peak storage capacitor, sensi- tivity control and logarithmic amplifier. The circuit may be used to drive either moving coil or LED voltmeters. Figure 2. Complete circuit of the PPM meter drive. An active rectifier built around A1 rectifies and inverts negative half cycles of the input waveform and charges C2 to their peak value. The feedback arrangement around A2 progressively reduces the gain as the input signal increases to provide a 'piecewise linear' approximation to a logarithmic curve. Figure 3. Transfer characteristic of the logar- ithmic amplifier. Since the diodes turn on gradually the transition from one portion of the curve to another is smooth, without any sharp break points. Figure 4. If the UAA 180 LED voltmeter is used with the PPMi drive circuit, each of the 12 LEDs can represent a 2 dB step. Figure 5 and 6. Printed circuit board and component layout for the PPM drive circuit. This is the same size as the LED voltmeter board so that the two may be stacked together to form a compact unit. (EPS 9860). together to form a compact, two- channel PPM Alternatively t the meter drive circuit may be used with a pair of moving-coil meters such as 100 /jA meters with 100 k series resistors or 1mA meters with 1 0 k series resistors. However, if moving-coil meters are used it is im- portant to remember that the response time of the meter will be affected by the mechanical inertia of the meter movement, and overshot?! may also occur if the movement is poorly damped. Testing and calibration The PPM requires a power supply of between 12 volts and 18 volts maxi- mum. If moving-coil meters are used then the supply should be capable of providing about 30 mA, but if the LED voltmeter board is used a 100mA supply is necessary. Before the outputs of the meter drive circuit are connected to the inputs of the LED voltmeter, the latter must he calibrated to read 10 V full-scale. This is achieved by connecting the L and R inputs of the LED voltmeter to a variable bench power supply, together with a multimeter set to the 10 V range (or nearest suitable range). The power supply output is adjusted until the multimeter reads 10 V, and P3 and P3 on the LED voltmeter board are adjusted until every LED in each column is lit. If moving-coil meters are used with the specified resistor values this procedure is unnecessary . The outputs of the meter drive board may now be connected to the inputs of the voltmeter hoard. Pi and PI on the meter drive board can be used to set the desired sensitivity of each channel. This will obviously depend on the peak programme meter elektor January 1978 — 1-19 -10 -is gfw-Ai ■! intended application of the meter. 1 he meter scale can be calibrated linearly from - 18 to +4 dB, the portion of the scale above 0 dB being marked in red to indicate overload. If the LED voltmeter is used then green LEDs can be used up to -2 dB and red LEDs from 0 to +4 dB so that an overload condition can easily be seen, \s it stands, the meter drive circuit will accept a maximum input of 5 V RMS. If larger voltages are to be measured (such as amplifier outputs! then a resistor must be included in series with each input to form a poten- tial divider with R 3 , For example, j 180 k resistor would allow input volt- ages up to c5 V RMS to be measured. K 0° y Parts list for left channel: duplicate for right channel Resistors R 1 - 47 k R2 r R4 - 470 k R3= 220 k R 5 - 1 k R6 - 100 n R7 = 15 k R8 - 12 k R9 = 1 k3 R10 = 10k R1 1 - 1 k2 R 1 2 - 1 k5 R13 - 120 n Pi - 100 k preset Capacitors : C1.C2 - 10^,16 V □IX □33 Semiconductors D 1 to D5 ■ 1N41484N914 A 1 ,A2 - %IC1 - 14324 1-20 — elektor January 1978 UAA 180 LEO voltmeter UAA 180 LED voltmeter l The Siemens UAA] 70 and UAA ISO ICs, which have been available for some time, are electronic replacements for a conventional moving coil voltmeter and are designed to drive a column of LEDs in response to an input voltage. As the input voltage is increased the LEDs light up in turn, thus indicating the voltage level. The principal difference between the two ICs is that the UAA 170 lights only one LED at a time (i.e. whenever a LED lights the previous LED in the column extinguishes) whereas the UAA 1 80 provides a thermometer-type indication (he, the LEDs, once lit, stay lit so that at full scale the whole column is illuminated). Both these ICs may be used to replace conventional voltmeters m applications not requiring fine resolution, which is limited to one-sixteenth of the scale length in the case of the UAA 170 and one-twelfth of the scale 1 eng (it in the case of the UAA 180. However the resolution can be improved by using more than one 1C and LED array, as will be described in detail later. The UAA 180 is most useful in appli- cations requiring rapid reading of the meter, such as in audio level meters, since it is much easier to judge the length of a column than it is to estimate the position of a moving point of light. The UAA 170, since it lights only one LED at a time, consumes less power and is thus more suitable for appli- cations where a thermometer- type indi- cation is not necessary, for example, in tuning scales for varicap FM tuners and other general voltmeter applications. Figure i illustrates the basic principle of the UAA 180 LED voltmeter 1C. A number of analogue comparators have their non-inverting inputs joined together and connected to the input voltage that is to be measured. The inverting inputs are connected to reference voltages derived from a potential divider chain between points A and B. A is assumed to be at a higher potential than B. With no input voltage, all the LEDs will be extinguished. When the input voltage exceeds the voltage on the inverting input of the first comparator, the com- parator output will swing positive and D1 will light. When the voltage exceeds Various meters based on the Siemens UAA 170 LED voltmeter 1C have previously been featured in Elektor (issue 12, April 1976 and issue 17, September 1976), This article describes a voltmeter circuit using a companion 1C — the UAA 180. The voltmeter can be combined with the PPM drive circuit described elsewhere in this issue ('peakmeter'} to make a compact audio level meter, and a novel board layout allows extension of the meter to any required number of LEDs. that at the second comparator input. D2 will also light and so on. An interesting feature of the UAA I 80, which readers may remember also applies to the UAA 170, is that both ends of the potential divider chain are accessible, and point B need not necess- arily be at ground potential. This means that the voltage range of the meter is determined by the potential difference between points A and B, but the zero point of the meter may be indepen- dently adjusted by varying the volt- age at point B. This facility makes it extremely simple to build a 'suppressed-zero’ voltmeter, which can be useful in applications where the voltage to be measured never drops below a certain value, and where the lower part of a normal (zero = 0 volts) meter would be wasted. For example, in equipment using a 9 V dry battery the minimum usable battery voltage might be, say, 8 \ Battery volt- ages below this are of no interest since the battery is then defunct, so the voltmeter need only read over the range of, say, H V to 10 V. Using the UAA 180 to measure only over this 2 volt range obviously gives much better resolution than using it as a 0 to 3 0 V meter. Another use ot the suppressed zero facility is to allow extension of the display to 24 LEDs, or even more. Take, for example, a voltmeter that was to read from zero to 2.4 V in 0.1 V steps. This would require two UAA I 80s, The inputs would be joined so that both were fed with the same voltage, but the first 1C would have point B connected to ground and point A to 1,2 V, whilst the second 1C would have point B at 1.2 V and point A at 74 \ Fhe first IC would display voltages from zero to 1.2 V, above which the second 1C would take over. There are other similarities, and also other differences, between the UAA 170 and UAA 180* Both the UAA 170 and U A A 1 8 0 have p rovisio n for c o nn e c 1 ing a light-dependent resistor to vary the display brightness to suit ambient illumination. One feature possessed by the UAA 170 however, which the UAA 180 lacks, is a reference voltage output. An external reference voltage UAA 180 LED voltmeter elektor Ja nuary 1978 — 1-21 Figure 1* Basic principle of the UAA 180* A series of voltage comparators measure the input voltage against reference voltages derived from a potential divider chain. When a reference voltage is exceeded the appropriate LED lights. Figure 2. Complete circuit of a two-channel LED voltmeter. Table 1, Absolute maximum ratings of the UAA 180, which must not be exceeded. Table 1 Absolute maximum ratings of the UAA 180, Ail voltages referenced to pin 1 . Supply voltage at pin 18: +18 V Input voltage at pin 17, reference volt- ages at pins 3 and 16; +6 V Operating temperature range: -2h to +80"'C. 1-22 — elektor january 1978 UAA 1B0 LED voltmeter must be provided for the UAA ISO, Another small disadvantage of the UAA ISO is that it is housed in an 18- pin DIL package. This can be a problem if [C sockets arc to be used, since 18- pin sockets are not exactly common! However, a solution exists in the form of Solder con socket strips, which can be ciil to any desired length. Complete voltmeter circuit Figure 2 shows a complete circuit for a two-channel voltmeter using UAA 180 ICs. This is especially intended as a stereo audio level meter for use with the PPM drive circuit described else- where in this issue. However, for single- channel applications IC1 and all com- ponents marked with an apostrophe (') may be omitted. The most striking difference between this circuit and the basic circuit given in figure 1 is that the LEDs are arranged in three series- connected groups of four. This means that when all four LEDs in a group are illuminated, the same current flows through all four from the supply, thus reducing the current consumption by a factor four compared to the arrange- ment of figure 1, where there is an independent connection to each LED. 1'he high and low r reference voltages for both channels are derived from a zener diode, D14. The high reference voltage is taken from a wiper of PI and is applied to pin 3 of each !C, The low r reference voltage is taken from the wiper of P2, to pin 16 of each IC. Since P2 derives its voltage from the wiper of PI the low reference voltage can never be higher than the high reference voltage. Parts List Resistors. R1 = 1 k5 R2 - 1 M R3 = 1 k R4, R4“ -10 k R5 - LDR PI = 10k preset P2,P3,F3' - 100 k preset Capacitors: Cl - 10 ju 35 V tantalum Sem i conductors: D1 , . 012,01“ . . . D12 r = LED {e.g. T I L 209) □ 13,013' = DUS 014 = 4V7/400 mW zener 1C1 JC2 = UAA I SO Miscellaneous: 2x18 pin IC sockets or 36 way socket strip. D14 also provides input protection for the ICs. The maximum allowable input voltage to the UAA 180 is 6 V. If the voltage on the wiper of P3 exceeds this then the voltage at the IC input will be clamped to about 6 V by 1313 and DI4, fhe same is true of the R input. Input voltages in excess of 6 V are accommo- dated by using P3 ox PV io attenuate the input voltage to below 6 V at the IC input. Automatic adjustment of the display brightness to suit ambient illumination is provided by connecting an LDR, R5, between pin 2 of the ICs and posit vt supply. R2 and R3 serve to limit the range of display brightness. If [Ins facility is not required then these components may simply be omitted, leaving pin 2 floating. Construction Printed circuit board and component layouts for the voltmeter module and LED array are given in figures 3 and 4.. The LED board is simply mounted at right angles to the main b ard using wire links. As mentioned earlier, the suppressed -zero facility makes it possible to extend the display length by using two or more ICs per channel. To facilitate this the supply connec- tions, brightness control (A), and 1. and R inputs are duplicated at both ends of the p.e. board so that two or more boards may be stacked together as shown in figure \ The following com- ponents need be provided only on one of the boards, and may be omitted from the other board - P3> P3\ R2, R3, R5, UAA 180 LEO voltmeter elek tor January 1978 — 1-23 ooooooooooooooooooooo 1> <4 S IS i£3 SO oo oo o o — oo — — 0 0 o d 0 n 0 ro 0 0 0 ofRTlo ji ® oiE^Jo - liiiiTlwfSi . ooo jopoooooo S4 n > a s — OO — -OOrr O 0~ > oooopooooooooo O O 6o{b£|o DI3 and D13\ These components have been omitted from the left-hand board of figure 5, It is also necessary to connect links in place of the omitted P3 and P3' as shown, and to provide links between the adjacent +, 0, L, R ? and A connections on the two boards. Calibration The LED voltmeter may be calibrated against an ordinary multi-range test meter. However, a small complication exists because the value of the reference voltage affects the manner in which the LEDs turn on. When the voltage between pins 2 and 16 of the 1C is the maximum (5.6 V) then each LED will turn on abruptly. At lower reference voltages, however, the LEDs turn on more gradually. The calibration pro- cedure given uses the maximum possible reference voltage, and if other reference voltages are required to suit individual requirements then the pro- cedure must be adapted accordingly, it is assumed that both channels of the meter will be calibrated to the same range. For input voltages up to 6 V the cali- bration procedure is as follows: 1* Using a multimeter, set the wiper voltage of PI about 10% below the maximum input voltage. 2. Feed in the maximum input voltage to be measured and adjust P3 until all LEDs are lit (top LED in the column being just lit). 5, Feed in the minimum input voltage that is to be measured and adjust P2 until only the first LED is lit. 4. Check the adjustment of P3, Table 2. Typical operating parameters of the UAA 180. Supply voltage: 1 0 to 1 8 V Supply current, excluding LEDs: 5.5 mA Input currents (reference voltage be- tween pins 3 and 1 6 less than 2 V): pin 3: 300 nA pin 16: 300 nA pin 1 7: 300 nA Current per LED: 1 5 mA Current consumption of the two-channel voltmeter board: 1 2 V supply: 75 mA, all LEDs lit 1 5 V supply: 80 mA, all LEDs lit Figures 3 and 4. Printed circuit boards and component layouts for the UAA 180 LED voltmeter (EPS 981 7L The display board is mourned at right-angles to the main board using wire links. Figure 5. The display may be extended to 24 or more LEDs by stacking two lor more) boards end-to-end as shown in this figure. Table 2, Typical operating parameters of the UAA 180. 5. Repeat (2) for the right channel. For input voltages in excess of 6 V the following procedure is adopted: 1 . Set Pi to maximum. 2. Feed in the maximum voltage to be measured and adjust P3 until all LEDs are lit (top LED in the column being just lit). 3. Feed in the minimum voltage to be measured and adjust P2 until only the first LED is lit. 4. Check the adjustment of P3. 5. Repeat (2) for the right channel. For extended versions of the voltmeter the adjustment procedure must be duplicated for the voltage ranges covered by the first and second modules. Care must be taken to ensure that the first LED of the second module lights at exactly the right voltage. It should not light before the last LED of the first module lights, nor should there be too great a gap between the last LED of the first module lighting and the first LED of the second module. Applications Use of the LED voltmeter with the PPM drive circuit to form an audio level meter is described elsewhere in this issue. Another interesting application is to use the voltmeter with a temperature- to-voltage converter (such as circuit No. 5 in the July /August 1977 issue) to form a novel thermometer. H 1-24 — elektor January 1978 experimenting with the SC/MP (3) experimenting with the SC/MPcs Interrupt operations An interrupt operation occurs when an externally generated control signal causes the suspend main programme The interrupt request will typically be issued by a e,g., a display which requires refreshing. When the CPU acknowledges such an interrupt, it jumps from the main pro- gramme to a special routine to service the interrupting device - after saving the return-to-main programme address. Once the interrupting device has been serviced, the CPU automatically resumes main programme execution. This pro- cess is illustrated in figure 1. Note that, in principle, an interrupt routine is quite similar to a subroutine call, except that the jump is initiated externally by an interrupt request rather than by an instruction in the programme, The situation becomes more compli- cated when the CPU receives several interrupt requests more or less simul- taneously and has to choose between more than one interrupt routine. When this happens the CPU basically has two ways of servicing the interrupts: the first is to run the routines sequentially, the second is to 'nest' the interrupt routines in order of priority. In the former case the CPU first determines the source ot the interrupt request then jumps to the appropriate routine* Whilst the CPU is executing this routine the interrupt input is inhibited, so that it will not respond to any further interrupts. Once the service routine has been completed the CPU resumes main programme However, if the CPU is then presented with a second interrupt request, it will once more automatically branch to the required subroutine. The problem of several interrupt requests occuring whilst the CPU is already executing an interrupt routine is solved be means of a priority encoder which assigns a differ- ent priority to each interrupt source. The CPU must therefore be able to interrogate the encoder so as to deter- mine the relative priority of the various interrupting devices. Figure 2a show's the sequence of subroutines; in this example routine has the highest priority. In the case of nested interrupts, the interrupt system is re-arnied immedi- ately upon the CPU branching to an interrupt routine, so that a second interrupt request can be acknowledged by the CPU at any time. Assuming, for example, that the CPU is already executing an interrupt routine when it detects a second interrupt request from a higher priority source, it will first branch to the routine which will service that device, then return to complete the initial interrupt routine, and only then return to the main programme (see figure 2b). During a routine the CPU will not acknowledge an interrupt request from a lower priority source. Jumping from one programme to another does not, in itself, present any special problems; one must simply ensure that the contents of the various CPU registers are not lost when branching to a subroutine, otherwise the original programme could not be executed properly . To preserve the status of the CPU's internal register values, they are stored in a stack. This stack consists of several general- purpose registers which store data on the principle of last-in/first out Ulifo ). Some microprocessors possess an inte- grated stack register, whilst others have instructions* which permit a stack to be programmed into the RAM SC/MP interrupt system The SC/MP h as only one interrupt input (Sense A), When the internal interrupt enable (IE) flag is set, by- executing either an Enable Interrupt Instruction (IHN> or a Copy Accumu- * In several microprocessors one machine instruction results in the CPU carrying out a large number of separate steps. Think, for example, of how many operations are in- volved in a DlY instruction in the case of the SC/MP The name for the total number of steps involved in a single machine instruc- tion is a 'micro-programme \ Computers and some microprocessors are 'micro-pro- grammabief i.e the micro-programme, and so The instruction set, can be altered to suit a special application. Naturally this technique requires a profound knowledge of the architecture of the CPU in question. SC/MP temporarily to execution, rpically be peripheral device such as, internally current This, the third article in the SC/MP series, introduces the memory extension card, which, in addition to containing % k of RAM and X k of PROM, also houses the multiplexer and priority encoder. The latter hardware allows the SC/MP to handle interrupt requests from more than one peripheral device. The article also examines the software involved in interrupt operations. H. Huschitt ICS 74143 main programme experimenting with the SC/MP (31 0000 interrupt elektor January 1978 — Figure 1. Diagrammatic representation of an interrupt operation. Figure 2, These examples illustrate the two basic methods of servicing multiple source interrupts, Figure3, Complete circuit diagram of all the hardware housed on the second Euro card. 1-25 -W C o. ICi sift im V M 7 I 1C 6 RAM MM7117 Mil nIB NRDS4 NWDS|-* Li C£ RAM imi*' Cihfifi.C/i-h =14 DA A RUS • DDRF6& BUS AUUHF-XH KliS ■ pj-i l.i-i A l L 0 1, S C TO RAM MM 7117 IC11 RAM MM7I1? 1C 14 MO v pAftw. Q--r HL P.l\f[i?04O N'I.Ni.W;g.N12.NT3.MB 'Cl 4049 Ni.m.Mg N 2 o ica - 4tm Nf, IMB.N1B.NI7 IC3 - 740B NIQ.NlI N 14 N I 5 iCrt- 7A0Q a/-- - - - - H _ u - 5 0^ O O © © -7V +SV L -0 TC.I != IC3 IC2 M ■ • ,1 ILJ 15 icia HAM m v j ' yj (CIS HAM WM2T1 S L-'A A HUS interrupt 0 = highest priority interrupt routine 0 interrupt routine i nterrupt routine 2 interrupt routine 3 mam programme 995 a. 2b ns m 4- 1 4-J ” S. CL 'C □ =1 3 1-26 — elektor january 1978 experimenting with the SC/MP (3s address x - don't care 3367 j 1 k byte's = 1 024 tayte's lator to Status Register Instruction (CAS), the Sense A line is enabled to serve as an interrupt request input. Upon detection of an interrupt request (SA is high) the SC/MP first completes the current instruction which is being executed before acknowledging this request. There then follows an internal DINT Instruction which resets the status register flag (IF), thus preventing the SC/MP from responding to any further interrupt requests. At the same time the contents of the programme counter are exchanged with the contents of pointer register 3. The next instruction which the SC/MP executes is that which is found under the address (PC) + 1. This means that before the interrupt se- quence begins, pointer register 3 must be loaded with the start address of the interrupt routine minus 1 . The return from interrupt to the main programme is effected by two instruc- tions: first Enable Interrupt (IKN), followed by Exchange Pointer 3 with Programme Counter (XPPC 3). The latter instruction copies the original contents of the programme counter, which for the duration of the interrupt routine were stored in PTR 3, back into the PC, allowing main programme execution to recommence. Since during the interrupt routine the programme counter is being continually incremented, this means that PTR3 will no longer contain the start address of this routine, but rather the address immediately following the end of the routine. Thus in order that this routine can, if necessary, be repeated, the subsequent address must contain an instruction to jump back to the start address of the routine (see table I ). Multiple interrupt capability The number of interrupt inputs of the SC/MP can be extended fairly simply to 8. All that is required is some extra hardware in the form of a 74148 ( ICS in figure 3) that is used as a priority encoder. The output (pm 13) of the priority encoder goes high whenever a L (T appears at one of its eight inputs. This *r is used to take the interrupt input (Sense A) of the SC/MP high, causing the CPU to acknowledge the interrupt request. The BCD outputs of the encoder indicate which of the inputs is low. This information is routed out onto the data bus via three buffers (IC9). The 0 input of the encoder has the highest priority, he, when this input is taken low, interrupt requests appearing at any of the other inputs arc ignored ■ by the SC/MP. To be able to service several interrupting devices requires not only the hardware of a priority encoder, but also ad- ditional software. This is particularly true in the case of the SC/MP, which does nth have any stack registers. Thus, in the event of an interrupt routine the contents of the SC/M P’s internal registers must be temporarily stored in an external 'software stack 1 , This is basically a programme which loads the contents of these registers into a section Figure 4, This diagram shows a detailed breakdown of the current page-address structure of the SC/MP system's memory. At a later stage the RAM I/O card will become redundant, and the CPU card will move up to occupy the first page of memory. Table 1, This table shows the various steps involved in entering and exiting an interrupt routine in the case of the SC/MP, Table 2. Example of a programme designed to process a number of interrupt requests. Table 3. A programme which enables the contents of the CPU registers to be checked by having them displayed on LEDs. Table 1 UNIT : ; label of interrupt routine • • ► interrupt routine proper * • IEN ; enable interrupts XPPC3 ; return to mam programme JMP INT ; jump to start address of in- terrupt routine of memory reserved for this purpose, and after the routine, loads them back into the original registers. A part of this programme will be discussed later in this article. Before the interrupt programme can begin the CPU must first interrogate the state of the priority encoder. An example of suitable interrupt software is shown in table 2. The actual inter- rupt routines and the way in which the interrupt requests are handled will of course depend upon the type of periph- eral devices which require servicing. For this reason it is impossible to provide universally applicable interrupt routines, these must be developed by the individual user in accordance with the requirements of his particular pro- gram me. Multiplexer In addition to the 8 interrupt inputs, main programme execution can also be influenced by a number of other inputs, namely the 8 inputs of the multiple xu IC7 (see figure 3). The logic stale of each of these can be tested by applying the ‘address 1 of the input concerned (BCD-coded) to the select inputs of the ‘Mux' (pins 9 , . . 11). The inverted version of the selected inpul signal then appears at the output (pin b) of the multiplexer. This output data hit is then pulsed onto bit 07 of the data bus via a tri-state buffer. Bit 07 w r as chosen since the status of this bit can easily be tested by means of the Jump If Positive (JP) instruction. The SC/MP experimenting with the SC/MP <31 efetuor january 1 978 — 1 -27 Table 2 Table 3 MAIN PROG; START = 0000 DINT ; disable interrupts 0000 08 NOP • 0001 C454 LDI L • section of main programme with (SAVSTA)— 1 * SC/MP inhibited from detecting 0003 33 X PALS * interrupts 0004 C400 LDI H LDI L (STACK} (SAVSTA) B J- load PTR 3 with the XPAL2 K load PTR 2 (stack pointer! with 0006 37 XPAH3 address of SAVSTA LD} H (^TACK) address of RAM stack 0007 if the programme under test XPAH2 is loaded from 0007 on LDi L load status back into CPU * BYTE • 0090 00 PI L: T etc. * J * BYTE IEN only +or sequential interrupts 0091 m P1H; XPPC3 * BYTE JMP INTIN . only (or sequential interrupts 0092 00 P2L: * BYTE 0093 00 P2H: 1-28 — elektor January 1978 experimenting with the SC/MP (3) USIM mutmth - i*:*:*- W jssstma ■»WSW«# J tl R Rid 567 RIO R19 R13 lllll WwM Parts list to figures 3 and 5 Semiconductors: IC1 = 4049 Resistors: (C2 - 4011 R 1 ,R2 = 2k2 ICS - 7405 R3 - 470 n* \ C4 - 7400 R4 - 220 P- * IC54C6JC10 .103 2112 R5, . > R20 - 4k7 IC7 - 74151 I CS - 74148 Capacitors: IC9 - 74125 Cl - 2.2 p/16 V 10 4 - MM 5204G C2 = 1 p/16 V- IC15 = 79G* C3 . C6 - ISO n * omitted for SC MR 1 1 experimenting with the SC/MP {3) elektor January 1978 — 1-29 will jump if this bit is 4 0’ tie. a positive number), and continue main programme execution if it is ( 1\ An example of the software involved when utilising the multiplexer is shown in the pro- gramme listed in table 3, Page-address structure As the volume of system hardware continues to grow with the addition of the memory card { shown in figure 3), so the need to clarify the address structure of the system becomes more urgent. The CPU card already contains a large portion of the memory capacity of the system (e,g. the PROMs for the monitor software), which, naturally enough, must be capable of being addressed. The CPU card is therefore supplied with an address decoder, With the advent of the additional memory capacity rep- resented by the circuit in figure 3 the page-address structure of the system’s memory takes the form show- n in figure 4. Half of the first memory page (0000 0EFF) can be addressed by the address decoder of the RAM I/O card. However, since the address decoding on the RAM I/O card is incomplete (AD 1 1 is not decoded) the second halt" of this page identical to the first half and cannot therefore be used for additional hardware. The second memory page contains everything which can be addressed by the address decoder of the C PU card. This consists firstly of the l wo PROMs tor the monitor software. To provide 'he option ot expanding the monitor ■> >f l ware, space is provided for a third • ik! PROM MO 14 in figure 3). The • . • c t ion f r o m 1600 to 17 F F i s re se r v e d : >r the multiplexer with priority en- . Jer and for the hexadecimal input/ output (HEX I/O) hardware which will be appearing shortly. The remaining lines of the second page are taken up by a Ik RAM, 3 4 k of which is present on the CPU card, with %k situated on the memory card as shown in figure 3. Once the hexadecimal input/ output has been incorporated into the SC/MP system, the RAM I/O card will become largely redundant. Once the user has acquired a certain degree of proficiency with the system he can be expected to dispense with the RAM I/O card completely. For this reason it is also possible to construct the system without the RAM I/O card. This is done by switching the wire links a and b {shown as dotted lines in figure 3) to their alternative positions, so that the first memory page is now addressed by the address decoder of the CPU. Every- thing which in figure 4 lies between 1000 and ]FFF is then situated be- tween 0000 and 0FFF + Board interconnections The complete circuit diagram of the memory-extension card is shown in figure 3. The track pattern and com- ponent layout of the printed circuit board for this card are shown in figure 5. This hoard, like the CPU card, is double- sided with plated- through holes, and conforms to Eurocard dimensions. It should also be fitted with a 64-way edge connector. The board houses a regulator 1C (1C I 5) which supplies the negative voltage for the earlier, PM OS version of the SC/MP (see Elektor 32, p. 3 2-08: SC/MP If and the p.c. board). If SC/MP [I is used, IC15, R3, R4 and (2 can be omitted; the wire link adjac- ent to the position for this IC is then connected to the point marked 4 5 V\ In order to interconnect the various Figure 5* The track pattern and component layout of the printed circuit hoard for the memory card {EPS 9363). Particular attention should be payed to the wire link to the right of IC15' when using the earner P-MQS SC/MP, this link should be in the '—7 V' position; for SC/MP II it should be in the '5 V' position. Figure 6. To interconnect the Eurocards, a connector bus can be formed by joining the corresponding pins of the socket con- nectors using wire links. Eurocards, a connector bus' is necess- ary. This basically nothing more than a number of socket connectors with the corresponding pins (all pins la, ail pins lb, etc.) interconnected. This arrangement is illustrated in figure 6. Whilst it is entirely possible to make all the necessary connections in this fashion (using e.g, ‘wire-wrapped’ links), such a method is both time-consuming and error-prone, For this reason a 'bus board', which can accomodate three socket connectors, was designed (see figures 7 and 8). The CPU card and the memory card can then be intercon- nected by simply plugging them into the bus board. Fhis bus board must of course be able to communicate with the RAM I/O card. Since the RAM I/O card does not use edge connectors, these connections must be hardwired. The wiring details are provided by figure 9. Termination points to the bus lines are provided at each end of the board so that several bus boards can be stacked end to end and linked to extend the system. The layout of these termination points allows the addition of extra 64- way sockets so that external con- nections need not be hard wired. The only major limitation to large scale expansion of th rt system is that imposed by the power supply. Anyone who plans a large system should bear in mind that each page of memory (4 k) consumes a current of approximately 1 A, A suitable 5 V/3 A, —12 V/0.5 A supply will be published in the near future. Software It will not have escaped most readers that the role played by software in this series of articles is gradually growing in importance. The reason for this is twofold: firstly the 'intelligence' of a 1'30 — elektor January 1 970 experimenting with the SC/MP (3) Figures 7 and S. A more convenient method of linking the Euroeards is to use this "bus board' (EPS 9857) which can accomodate up to three 64 pin edge connectors. Several bus boards may also be stacked together to expand the memory capacity of the system even further, Figure9, This diagram shows the wiring connections between the RAM I/O card and the bus board. Figure 10. By means of a multiposition switch it is possible to display the contents of each CPU register in turn on the LEDs, computer system is largely determined by the number and lype nt programme at its disposal: secondly, it is through developing his own software that the user can he si appreciate the true poten- tial of his system. To this end, the present article con- cludes with a short "debug' programme which wall display the contents ot the CPU registers at an\ stage during the programme under test. This is done by replacing the instruction which immedi- ately follows the 1 suspect' section of the programme by XPPC 3 l3F), The programme under test is then started as normal, after an NR ST instruction. When the programme reaches the XPPC 3 it jumps to a 'save status routine" and writes the contents of the CPU registers into the RAM. The Mux inputs are con- nected to a multiposition switch (see figure 10), by mean> ■ >! which the register whose contents are to be displayed on the LFDs can be selected, in this way the contents of each CPU register, with the exception of course of PTR 3 and th e PC, can be examined in turn. The programme in its present form can only be exited from by means of an NRST instruction A more sophisticated and convenient version of the pro- gramme will later be incorporated into the monitor software. experimenting with the SC/MP (3) missing elektor january 1978 — 1-31 + BV EMIN N HOLD AD 10 ADld RAM - I/O ■ ADdd DB07 0 DB00 nrst{] cont(] NADS^ MWDS^ CE RAM I/O § DB 07 0 DB00 ^ MRST ^ CQIMT [} NADS [) NWDS 0 ee A n{ ^ lC 1 cK 1 )cr c 2 (pC - £R F >TR 1 L F TR 1 H yj — 'TR 2 L s 0- — 3 TR 2 H 6 0 BE The listing for this programme is given in table 3, The "save status routine' can also be used for interrupt operations* In this case the section of RAM from 008 D . . . 0093 is used to form a soft- ware stack. The majority of the instructions con- tained in this programme have already been discussed and require no further explanation. One important exception however is the "indirect 7 address mode utilising the extension register. As explained in part 1 ("address modes'), indirect addressing describes the address mode whereby the effective address (EA) is obtained by incrementing the contents of a pointer by the contents of a byte taken from the RAM. In the case of the SC/MP this can only be done with the aid of the extension register. When the displacement value is X'80, then (for memory reference instruc- tions) it is no longer used to obtain the effective address, but is replaced by the contents of the extension regis- ter, The contents of the extension register are not known at the time of entering the programme, but are determined during execution of the programme. In this programme the indirect address mode results in a considerable saving in the number of instructions. N ( to be continued ) Modifications to Additions to Improvements on Corrections in Circuits published in Elektor Formant — the Elektor music synthesiser Parts 4 and 5, October and November 1977. In Part 4, R1 in the input adder circuit (figure 7) is shown as 1 00 k. This gives the ‘Octaves, coarse' control (PI) a range of + 7 x h octaves. At a later data, it was decided to reduce this range to ± 5 octaves, hi pail 5, this modifi- cation was carried out: in figure 2a, Rf is shown as 15Qk; the front panel lay- out (figure 3) shows a "coarse’ range from -5 to +5. However, we forgot to point out that the value of Rl shown in figure 7 of part 4 is "in correct'. Furthermore, and more seriously, the Octaves/ Volt adjust- ment was based on the original value of Rl, Thi s means that the Octaves/ Volt adjustment procedure described in part 5 is incorrect. For the adjustment procedure using a DVM and a frequency counter, the connection between the slider of PI and Rl must first be unsoldered* The free end of Rl is then connected to ground, and the slider of PI is connected to the KOV input with SI in position V. Adjustment can now proceed as de- scribed in the original article. The second adjustment procedure, using the beat note method, is correct as originally described* M 1-32 — elsktor January 1978 A/ an invitation to investigate, improve on and implement imperfect but interesting ideas third octave filters In the article on the Elektor equaliser (see elsewhere in this issue) it was noted that third octave filters represented a more ideal solution to the problems of room equalisation , but that their complexity meant they were prohibitively expensive to build and use in a system which covered the entire audio spectrum . For this reason they were discarded as a suitable basis for the -0 — 7 Eire Elektor equaliser. Rather than throw out the baby with the bath water h o w ever, v aria u s l e ss comp lex arrangements of a third octave filter system were examined in an attempt to find a more financially viable application. 7 h e fo 1 1 o w ing c ir c u i t provides a reasonably acceptable solution . The basis of the compromise solution is shown in figure 1; this response differs from that shown in figure 2 of the equaliser article inasmuch as ah the rectangular passbands He below the O-dB line. The filter therefore applies only cut, which falls to zero at the top end stop of the potentiometer in question. The frequency response curves shown in figure 2a coincide with the lower half of figure 4 in the equaliser article; at the band edge frequencies f\ and fs , the attenuation is -3 dB for the lower end stop (-7) of the appropriate filter potentiometer. Similarly, the curves showm in figure 2b correspond to the ejektor elfiktor january 1978 — 1-33 3a 20.. 30V _x‘ C 1 r-- C -4 0 lower half of figure 5 in the equaliser article. The filters here, however, have one-third octave centre-frequency spacing, he. the frequencies f-j and f 2 are nearer the centre frequency fo than in the case of octave filters. As in the case of the Elektor equaliser, the relatively less selective filters (lower Q, see figure 2b) offer superior musical performance. Figure 3a shows the circuit diagram of two cascaded filter sections. The filter proper consists of a series resonant cir- cuit .in series with the potentiometer resistance R s , The filter input voltage Uj is supplied by an emitter follower. The output voltage u Q is taken from each filter section via the wiper of the potentiometer. Alternate PNP- and NPN-transistors should be used w r hen cascading the different sections, since the base-emitter voltage drops of alter- nate stages are then of opposite polarity , and cancel out. The emitter followers can be replaced by op-amps, which should then be con- nected as voltage followers; quad op- amps in particular should prove suitable for this task. To electronically syn- thesise the inductance, the circuit shown in figure 1 1 of the equaliser article (be, the modified version of the circuit in figure 9) is an obvious solution. A discrete-component equivalent is shown in figure 3b — basically a gyrator circuit. A complete equaliser w r ould consist of a cascade of 30 of these filter sections. However an alternative answer that is particularly appropriate in the case of a long filter chain, since it both reduces the number of active components and offers a superior signal-to-noise ratio, is to combine a number of fitters whose centre frequencies fo are sufficiently far apart in the manner shown in figure 4. The values of R should be several times larger than those of The frequency response curve of a filter section expressed mathematically is as follows: , where The gain at f = fo is Ki : The value of K 2 is fixed, whilst Kj depends upon the potentiometer setting j3, and may vary between 1 and K 2 . The maximum attenuation at fo is A dB = -20 log K 2 dB, I he parameters of the series resonant circuit are: lo - 2tt\/LC ■ q = T\ A. ’ ^ R e V C ’ L= R e R g C g ; C = ■ V ; 6 “■ 27rtoQR L ' r = _ 0 g 2jri'o Rg The value of Q is determined by the choice of the 3 dB points and by K 2 , therefore by the maximum attenuation at the centre frequency fo- In the case of third octave filters and with the 3 dB points shown in figure 2b Q 1 equals 4,32 * K 2 Af- 2 If R s =4k7 and R e =lk2, then K 2 =4,92 and A= 1 3 .83 dB. from which follows that Q = 4 t 51, The values of the capaci- tors C and Cg can now be calculated. Figure 1 can serve as basis for the choice of the centre frequency. It may be necessary to change the value of R^ somewhat in order to obtain a suitable value for Cg. Literature: J, Eargle: Equalising the Monitoring Environment; Journal of the Audio Engineering Society, March 1973. D . Davis and Zb Pahnquist: Equalising the Sound System to match the room; Electronics World y January 1970 . tie ktor equ al ise r, see elsewh ere in th is issue. 1-34 — etektor January 1978 formant synthesiser (7) It is often not realised, even by mu- sicians, how much the character of an instrument is determined by the dy- namic amplitude and harmonic behaviour, rather than by the steady- state harmonic content of the instru- ment. If the attack and decay periods of a note are artificially modified, then the whole character of the sound is altered. An interesting and amusing experiment is to record the sounds of several musical instruments, but to remove the attack and decay periods by bringing up the recording level after the note starts and fading it down before the note ends. Then ask some musical friends to identify the instruments. They will no doubt be amazed how characterless the sound of an instrument becomes when robbed of its particular amplitude envelope. On the other hand, starting with i single basic waveform such as the triangle out- put of the Formant VCO. a whole range of instrument sounds can be produced simply be varying the amplitude envel- ope, ranging from ‘soft’ sounds such as flute and some organ voices, to "hardy percussive sounds such as piano and xylophone. Envelope control of the harmonic content using the VCF allows even greater variation in the character of the sound. Types of envelope curves I he envelope shaper of the synthesiser must be able to simulate the envelope contour of conventional musical instru- ments when the synthesiser is used in an imitative capacity, and also to produce envelopes that are purely synthetic in character (i,e. not found in sounds produced by normal acoustic methods). Fortunately, there are relatively few types of envelope contour that are musically important, and these are all fairly easy to generate electronically. 1. Attack/decay contour The simplest type of envelope curve is that consisting only of attack and decay periods. The envelope contour rises to a peak when the note is played, and begins to decay immediately the peak is passed (see figure 1 ). By varying the attack and decay times a wide variety of sounds can be produced. For example s if a rapid attack and slow decay is applied to the VGA control, then a percussive sound like a piano results. Applied to the VCF in the low- pass mode, the same envelope contour can produce very bright, metallic sounds, depending on the input wave- form . If the attack period is made long and the decay period short, then applying this to the VC A will produce com- pletely synthetic "fantasy' sounds simi- lar to those obtained by playing a recording backwards. Applying this type of envelope contour lo the VCF can produce sounds similar lo those of a brass instrument played staccato However, the main use of this type of envelope curve is for the production of percussive sounds such as xylophone, marimba, glockenspiel, bells and gongs, cymbals, and struck or plucked strings such as guitar, banjo, harp, other string instruments played pizzicato, harpsi- chord, and o I course, piano. 2. Attack -sustain -re I ease contour The attack/decay characteristic pre- viously described is typical of instru- ments where the sound is initiated by a short pulse of energy (e g. by striking or plucking a string), after wtuc h Ih e sound dies away since there is no further excitation to sustain it. The envelope contour shown in figure 2 is typical of instruments in which a note is sounded and sustained, such as a pipe organ, woodwind instruments, and bowed siring instruments. In a pipe organ the note builds up fairly rapidly after a key is depressed as standing wave modes are established in the pipe, and the note is sustained by virtue of the fact that air is continuously blown into the pipe When the supply ot air stops on releasing the key the note terminates more or less rapidly. The same basic contour applies to woodwind instruments and to string instruments played with a bow. since the note is here again sustained by blow i ng or bo w mg . 1 1 owe very w i Lh s u c h instruments much greater expression can be obtained by modulation of the The ADSR (Attack-Decay-Sustain- Release) shaper described in this article can be used to control the VGA and VCF to impart a wide range of tone colour and amplitude dynamics to the VCO waveforms. C. Chapman formant elektor january 1978 — 1-35 U(V1 Amplitude steady -state level, since this is deter- mined by the player, and not by a mechanic a 3 blower as is the case with a pipe organ With a synthesiser, a degree of ex- pression can be obtained by modulating the VC A using the low-frequency oscillators or noise source. 3 Attack-decay-release contour \ variation on the attack-decay contour :> shown in figure 3. Here the slow Figure 1, The attack -decay envelope contour is the simplest contour found in music. Figure 2. The attack-sustain-reiease contour is used to simulate instruments where the note can be sustained at a constant level, such as organ, woodwind, and bowed string instru- ments. Figure 3. Instruments such as the piano can be simulated using the attack-decay-release contour, As long as the key remains de- pressed the decay path is followed, but once the key is released the note is ended more abruptly, following the release contour. decay is allowed to continue for only a certain time, and the note is then terminated by a more rapid release. The most common example of this type of contour is provided by our old friend, the piano. When a note is sounded and the key remains depressed, then the damper is held off the siring and the note decays over a period of a few seconds, If, however, the key is released after playing a note, the felt damper contacts the string and the note termin- ates after about 500 ms. 4, Attack -decay -sustain -re lease contour Most of the examples given so far relate to envelope control of the VC A, since the amplitude contour of a sound is somewhat easier to visualise than its dynamic tone colour behaviour. How- ever, the most complex envelope contour, shown in figure 4, is a good illustration of envelope control of the VCR Many brass instruments, such as the trumpet, are characterised by a rapid build-up of harmonics during the attack period of the note, which gives the instrument a very strident sound. Once the note is established, however, the harmonies die away somewhat, and the tone is much more mellow during the steady stale period, f inally, during the release period at the end of the note, the n o te dies a w ay fa i r 1 y ra p i d 1 y . This type of characteristic can be obtained by using the VCF in the low- pass mode and controlling it with an envelope contour similar to that shown in figure 4. As the control voltage rises during the attack period, so the turn- over frequency of the VCF increases, passing more harmonics. During the decay period the VCF turnover fre- quency falls until the steady-state value is reached, and finally, during the release period the VCF turnover fre- quency drops very rapidly. Envelope shaper requirements It is apparent from figure 3 that the envelope contours shown in figures 1 to 3 are merely special cases of the more general a t tack -decay-su stain-release con- tour illustrated in figure 4, Any of the four contours can be generated by an envelope shaper having the following four functions: • variable attack time (A) • variable decay time ( D) • variable sustain level (S) • variable release time(R) These four parameters can be preset manually using the ADSR controls of the envelope shaper. The envelope shaper is controlled by the gate pulse output of the keyboard. When a key is depressed the gate output goes high and this initiates the attack-decay sequence. The output of the envelope shaper then remains at the sustain level until the key is released, when the release period begins. 1-36 — dektor January 1978 formant Block diagram The required exponential attack, decay and release characteristics are easily obtained by charging and discharging _ capacitor through resistors, and the sustain level by clamping the capacitor voltage to a preset D.C. level during the sustain period. The basic principle of the envelope shaper is illustrated in figure 6, The gate pulse is fed to a volt- age follower A 1 . and when the gate pulse is high C charges exponentially through P2 and D2 (and 13), At the end of the Attack period, ‘switch' T3 is opened and T6 is closed. Capacitor C now discharges through D4 and P3 (Decay), until the Sustain level is reached. This level is maintained until the gate pulse finishes, either when the key is released or when a preset time has elapsed. When the gate pulse finishes, the output of A1 goes to zero volts, and C dis- charges through D1 and PI (Release). The capacitor cannot discharge fully, ADSR adjust mem ranges; Attack period (A) 10 tins, ,20 s Decay period (□) 10 ms . 20 s Sustain level (S 1 0.5 V . 5 V Release period (R) 10 ms 20s since D I ceases U> conduct once she voltage on ( has fallen to aboui IJ.5 V. but this is not important as it merely constitutes a D.C. offset which can be compensated lor l he attack, decay arid release times may be adjusted bv means of P2, P3 and PL Complete circuit The co ni pic I e circuit, which is shown :n figure 7, is, of course, more complicated The envelope shaper has two modes of operation, ADSR and AD, which are selected by means of SI. With S3 in position 'IT (.ADSR) the circuit operates as follows: When a key is depressed the gate pulse output goes to +5 V. KT has a gain slightly greater Ilian unity, so about ' fp V appears at its output. ITie leading edge of the gate pulse also triggers monostable T1/T2, which pro- formant elektor January 1978 — 1-37 V5V * tantalum T1 . . . TB = BC108C.BC109C □ 1 . . . D5.D7 - 1N4143 D6 = LED IC1 ... IC5 = jiA 741C,MC1741CP1 (Mini DIP } am i mo P5 iiyk , . [in T& V D6 ]% 15V © Ft?? EOS C ^ ADSR pi 2 ENV M-^uIl Ol duces a short pulse to set flip-flop T4/ T5 (T5 turned on and T4 turned off). The collector voltage of T4 thus rises, turning on T3 and allowing C2 to charge from the output of 1C 1 through T3, P2, R] 7 and D2. This is the attack period. The voltage on C2 is fed to voltage- follower buffer IC4 } which is connected to the outputs EOS and ENV and also to the non-inverting input of IC3* This IC functions as a comparator, with its inverting input held at about 4.7 V by R24 and R25. When the voltage on 02, and hence at the output of IC4, exceeds this value, the output of IC3 swings positive, resetting flip-flop T4/T5, turning off T3 and terminating the attack period. T6 is turned on, initiating the decay period when C2 discharges through 04, R21, P3 and T6 into the output of IC2 until the sustain level, set at the output of voltage follower JC2 by P4, is reached* The output of the envelope shaper then remains at the sustain level until the key is released, when the output oflCl goes to zero volts and C2 discharges through 01, R] 3 and PI (release period). Diode 07 protects C2 in the event of the output of IC 1 going negative for any reason, when the voltage across C2 is clamped to a maximum of —0*7 V r A LED indicator constructed around IC5 allows visual monitoring of the envelope contour. The brightness of the LED follows the envelope voltage. Two outputs arc provided from the envelope shaper; an external output to a front panel socket (EOS), and an internally wired output (ENV), The full ADSR envelope contour is, of course, produced only if the key is depressed for a period longer than the attack plus decay time, and if the sus- tain level is greater than 0%, If the key is released before the sustain level is reached then the release period is initiated prematurely, and either AR or ADR curves may be produced. If the Figure 4. The a ttack-decay-su stain-release contour is the most complex envelope shape provided by the Formant envelope shaper, lA/hen applied to the VCF it is useful for imi- tating brass instruments, where the harmonic content of the note rises initially to a large value, then reduces to a lower level during the steady -state part of the note. Figure 5* By varying the sustain level the envelope contour can be changed from an AD contour at 0% sustain, through various ADSR contours to an ASR contour at 100% sustain. T is the time for which the key remains depressed* Figure 6. This simplified diagram illustrates the basic principle of the envelope shaper* C charges through D2 and P2 during the attack period. It then discharges through D4 and P3 to the (adjustable) sustain level; finally, it discharges through D1 and PI during the release period* Pi, P2 and P3 can be used to vary the release, attack and decay times, Figure 7. Complete circuit of the Formant envelope shaper. sustain level is 0% then only AD or ADR curves may be produced, depending on when the key is released. If the sustain level is 100% then, of course, only AR or ASR curves may be produced, depending on when the key is released, since the decay period is inhibited. Triggered AD mode It is sometimes useful to be able to pro- duce AD envelope contours that are unaffected be releasing the key, that is to say, once the key is depressed, a fixed attack -decay sequence is initiated, which is completed whether the key is released or not. this triggered AD com tour is obtained by selling Si to pos- ition 'a 5 and selecting 0% sustain level. The input of IC 1 is now connected to the junction of R1 and R2, so its output is permanently at about +6 V, irrespect- ive of the gate input. When a key is depressed, the gale signal triggers the monostable, setting the flip-flop and turning on T3. At the end of the attack period, comparator IC3 resets the flip-flop, turning on T6 and initiating the decay period. C2 will now discharge through D4, R21, P3 and T6 to the 0% level (sustain is set at 0%). Even if the key is released before this sequence is complete, the release period is inhibited since the output of IC 1 is permanently at +6 V, so C2 cannot discharge through Dl, R13 and PL. Construction There are no special requirements with regard to resistor tolerances in the envelope shaper circuit, and ordinary, good-quality 5% carbon film com- ponents are quite adequate; C2 should be a tantalum electrolytic capacitor for low leakage, and Cl the usual polyester or polycarbonate type. Semi- conductors should all be from a repu- 1 '3B - elektor January 1 973 formant "ormant elektor January 1978— 1-39 Parts list tor figures 7 and 8 Resistors: R1,R9,R23 “10k R2,R25=4k7 R3 r R7 = 5kb R4,R6,R8,R16,R18 - 100 k R5,RlO r Rl 1 r R22 = 33 k R 1 2, R26, R27 - 470 H R13.R21 - 1 k R14,R20 - 27 k R1 5 r R19 = 6k8 R17 = 220 n Potentiometers: PI ,P2,P3 = 1 M log, P4 = 10 k lin. P5 - 25 k preset Semiconductors: T1 . , , TG = BC108C, BC 1 09C or equivalent D1 . . , D5,D7 - TN4148, 1N914 □6 = LED (TIL 209 or similar) I Cl , . IC5 * juA 741 C r MC1741 CP 1 (MINI DIP) Capacitors.: Cl = 10 n C2 - 10/1/16 V tantalum C3.C4 - 10 ju/16 V Miscellaneous: 31 -way Euro connector (DIN 41617} 1x3.5 mm jack socket 4 x 13 15 mm collet knobs with pointei J 3 • :,i 1:>K' manufacturer , and it is a good idea lo test 13 and 16 for leakage, using the method detailed in pari 5 r \ printed circuit board and component layout for the envelope shaper are uven m figure 8, and a front panel a you I is shown in figure 9. Connections :o the front panel are fairly simple, the nly front panel-mounted components ocing the four potentiometers for attack inu\ decay time, release tune and sustain level, switch SI. the external 'Ut put socket and the envelope indi- cator LED Testing and adjustment Fo rest the envelope shaper agate pulse •iust be available from the 91 A \ \ out- put of i he infer laec receiver board The OS output of the envelope shaper is •mi to red on an oscilloscope with the V seiisili\ily set to about 1 V/div and c l tine base set lo about 10 rns/div. >r the first test, the sustain level is set ' ■ zero, S: is set to the “AIV position and the attack and decay poten- ■ meters are set to hast'. 1 he release lent tome ter has no effect during this 9, If a ke}, is depressed at short ie reals then a sitorl AI3 envelope curve .3 be seen, which rises and falls ‘ e tween about 0.? V and 5 V, The out- Figure8, Printed circuit board and com- ponent layout for the envelope shaper (EPS 9725-1 ) . Figure 9, Front panel layout for the envelope shaper module. pul of IC3 can also be monitored, to check that it swings briefly between 1 5 V and +1 5 V when the peak of the attack curve is reached. The only adjustment required lo the envelope shaper is to set the 1 00% sustain Level, using 1*5, to correspond with the voltage on C’2 at the end of the attack period. If it is too low, then there will always be a decay, even at 100% sustain level; if it is too high then the calibration of P4 will be inaccurate, since 100% sustain will be reached before maximum rotation of the poten- tiometer. To make the adjustment, the sustain level is set to 100% and medium attack and decay times are selected. Preset P5 is then adjusted until there is just no decay after the attack period (he. the attack period blends into the sustain level with no dip). The adjust- ment can be checked by turning P4 slightly to the left, when a slight dip after the peak of the attack period should be noted. As P4 is turned further anticlockwise then the decay down to the sustain level will become greater and greater, until finally, at 0% sustain level, pure AD curves will be produced. The envelope shaper is now ready for use. (to be continued) H 1-40 — elektor jartuary 1978 simple function generator simple function Most commercially available function generators suffer from the distinct, dis- advantage that they represent a pretty hefty investment for the amateur con- structor, who, unlike a service workshop for example, is unlikely to ever make full use of the wide range of facilities offered by a professionally produced instrument. For this reason, the circuit described here, which incorporates a special function generator 1C, type XR22Q6, was designed to strike the right balance between cost and perform- ance. Although lacking hop-notch' speci- fications, it offers a wide range of wave- forms, is both simple to build and calibrate, and is extremely easy to operate. The function generator can switch between sine, square, triangle, sawtooth and rectangular pulse waveforms. It has a linearly calibrated frequency scale which covers a range of 9 Hz to 220 kHz. In addition to a special output stage which ensures a low r output impedance, three calibrated output voltage ranges are provided; 0 ... 10 mV, 0... 100 mV and 0... 1 V (RMS). The circuit can be calibrated without the assistance of an oscilloscope, and the compact design means that it can be easily mounted in a neat case. The XR 2206 The circuit utilises the purpose-built IC function generator, type XR22G6 (Exar), the pin configuration and internal block diagram of which arc shown in figure F The heart of this 1C is the VCO (which in fact is a current controlled oscillator, CCO, though the manufacturer’s data calls it a VCO). The frequency of the oscillator is deter- mined by the external capacitor and resistor, C ext and Rexi* V control current, If, is switched via integrated current switches to one of the two cur- rent outputs (pin 7 or 8) of the IC, depending on the logic level of the selector input (pin 9), thus providing the possibility of frequency shift keying (FSK). The VCO output is buffered by an integrated transistor, the collector of which is accessible at the synchronis- ation output, pin 11. This output pro- vides a rectangular pulse waveform. In addition the VCO signal provides the basis for the signal generation carried out in the multiplier and sine converter- section. Pins 13... 16 allow adjustment of sine purity (distortion factor) and symmetry. The DC level at the signal output can be adjusted via pin 3. The sine, triangle and sawtooth wave- forms are buffered via a voltage fol- lower, and are brought out at the low impedance output, pin 2. The amplitude of the sine/triangle out- put can be varied linearly by a control voltage at AM input pin 1 of the 1C. This makes amplitude modulation of the oscillator signal possible. The voltage between the current con- nection pins 7 and 8 is stabilised to 3 V (typically) within the IC As this refer- ence voltage exhibits only a very small temperature coefficient (6x lO" 5 V/°C), the temperature stability of the oscil- lator frequency is also very good. The control current l( may vary be- tween I /a A and 3 niA however, opti- mum temperature stability is obtained in the range between 1 5 juA arid 750 fJtA, The frequency ot the VCO is determined by this current If and the value of the external capacitor C eX f, the control current being adjusted by means of a resistance Rf between pins 7 or 8 and earth. The equation for the frequency is as follows: If f = _ • The amplitude of the output signal may be varied by means of P2 and P3, The adjustment is carried out separately for sinew ave (T2) and triangle/sawtooth (P3) in order that the peak-peak value of all three voltages be the same; S3a allows for switching between P2 and P3. The symmetry of the triangle and sine waveforms can be adjusted by means of potentiometer P4, whilst the distortion factor of the sine signal can be varied by means of P5, Switching between sine and triangle waveforms is achieved by S3b. When switch S4 is closed a sawtooth signal is present at output A. The inte- grated current source will then switch belw r een pin 7 and 8 at a rate equal to the frequency of the rectangular pulse signal at output B, thus providing an ‘automatic’ frequency shift keying. The slope of the trailing edge is determined by the value of R8, which should be not lower than 1 k„ 1-42 — elektor January 1978 simple function generator Figure 4a. The complete circuit diagram of the function generator section. Figure 4b t The output stage ensures that the generator has a low impedance output and allows precise adjustment of the output volt- age. Figure 4c. The power supply is built round an integrated voltage regulator. Figures. Component layout and track pat- tern of the primed circuit board for the function generator (EPS 9453). 1-44 — elektor January 197B simple function generator The output stage A prerequisite of a good signal generator is a low output impedance and a precise, easily adjustable output voltage. Both these conditions are met. by the output stage shown in figure 4b. The sine, triangle and sawtooth signals from output A of the generator stage are fed via switch S5 to the base of TL The square wave and pulse signals are present at output B of the generator, this output being the collector terminal of a buffer transistor contained within the IC (see figure 3 ). R9 is the collector resistor of this transistor, and at the same time, together w r ith RIO, forms a voltage divider which limits the ampli- tude of the square wave signals to approximately 4.5 V. This ensures that the sync, output is both TTL compat- ible and short-circuit proof and may therefore be used to drive TTL cir- cuitry , as well as to provide synchronis- ation and trigger signals for an oscillo- scope. Tl ? which is connected as an emitter follower, buffers the relatively high impedance outputs of the gener- ator (600 £2 and 2000 12). The division ratio of the voltage divider R l 1 , . , R13 are 1,10 and 1 00. thus dividing the out- put amplitude into three switchable (by means of S6) decade ranges. The output voltage can be varied continuously within these ranges by means of P7, The actual output stage itself consists of transistors T2 to T5, which together form a DC coupled voltage follower. T2 and T3 form an emitter-follower con- sisting of a complementary Darlington pair, u'hich ensures that the output stage has a high input impedance and that the output transistors T4 and T5 ? which are also a complementary pair, arc driven from a low impedance source. The high input impedance reduces the load on P7 and allows a non-elect roly tic capacitor to be used for C7. Via diodes D1 . , . D3 transistors T4 and T5 receive a base bias voltage which causes a quiescent current of approx. 30 mA to flow through the emitter resistors. This measure effectively reduces the distor- tion of the output stage, C9 AC couples the output signal. The impedance of the AC output is approximately 5 £2, which means that it can be connected direct to a loudspeaker. The AC output is also short-circuit proof. The power supply The supply (see figure 4c) is quite straightforward, being built round an 1C regulator which produces a stable 12 V output. Since the supply, generator and output stage are all mounted on the same board, the only external connec- tion required is the mains transformer (approx, 1 5 V/0.5 A). LbD D8 provides on /off indication. Printed circuit board and front panel The entire generator is mounted on a j single printed circuit board (see fig- simple function generator elektor January 1978 — 1-45 ure 5), thus considerably facilitating construction. Figure 6 shows a suggested design for the front panel The individual controls and sockets are arranged in functional groups for ease if' operation. The power indicator LLD, 08, is mounted above the on/off switch. To the right of them is potentiometer PI which controls the signal frequency. The large easily-read scale allows fine :requcncy adjustment. The desired fre^ -jeney range can be selected using the Hz 1 switch (x 1, x 10, x 100, x 1000); e. 10 ... I 10 Ilz : 100 Hz. . . 1.1 kHz, kHz . + , 1 1 kHz, 10 kHz ... 1 10 kHz : ach of these frequencies can be . ubled using the f x 2 switch, so that c it frequency ranges are available in Figure 6, The ergonomically designed front panel facilitates operation of the function generator, figure?. Wiring diagram for the sockets, switches and potentiometers situated on the front panel Figures. The single multiposition switch used to select the desired waveform can be replaced by three separate switches (S3a, S3b, S4 and $51 Figure 9. Accurate frequency calibration can be achieved by using this simple supplemen- tary circuit. all. The selector switch for the various waveforms is situated to the right of the frequency controls. The output voltage is continuously vari- able between 0 ,,, 10 mV, 0 ... 100 mV, and 0 ... 1000 mV, the appropriate range being selected by means of the l mV’ switch (x 1, x 10 and x 100), The output signal is taken from the L AC ter- minals, and the synchronisation signal from the "sync" terminals. Wiring and construction To further facilitate construction of the function generator a wiring diagram (see figure 7) is provided. In particular, the wiring of the selector switch for the 1-46 — etektor january 1978 simple function generator various waveforms seems fairly compli- cated at first sight. A 4-pole, 5 -way switch is required, which must first be wired "internally" and then soldered to the appropriate connections on the printed circuit board (see figure 7). It is recommended that screened wire be used for switch 85, since this will pre- vent crosstalk from the square wave sig- nal on these leads. The wiring for switches SI, S2 and S 6 , as well as that for the AC and sync outputs, should present no special problems. Components A wirewound potentiometer is rec- ommended for PI , since this type gener- ally has a superior linearity to that of carbon potentiometers. Of course the use of a 10 turn potentiometer with slow motion drive, w r hich would provide extremely accurate adjustment of fre- quency. is also possible; however this would naturally involve somewhat more expense. Only close tolerance capacitors (MKM) should be used for Cl . * . 04. It is also worth mentioning that it is, of course, possible to replace the multi- position switch used to select the desired waveform by three separate switches (see figure 8). This solution does complicate the operating pro- cedure slightly, and whether it proves cheaper or not will depend on the type of switch which is used. Calibration After the components have been soldered onto the circuit board and the external switches and potentiometers have been wired up, the entire construc- tion should be carefully checked. Once this has been done power can be applied and the on load supply voltage measured. This should not vary more than 10% from 12 V. Amplitude calibration • First of all switch S6 should be set to position 1 (x 100) and potentiometer P7 turned fully clockwise (maximum amplitude). • Select a sinewave signal with a fre- quency of approx. 1 kHz. • Set P 2 for minimum amplitude, he, turn the wiper to earth. • Set P4 and P5 to their mid-position. • Connect a universal multimeter with an AC voltage range of 2 V RMS to the AC output of the generator, and adjust P2 for an output of either 1 V or 2 V RMS. The above step requires a little clarifi- cation, The advantage of selecting the higher output voltage of 2 V RMS is offset by a resultant deterioration in the quality of the waveform at high fre- quencies (above roughly 50 kHz), Thus, in order to obtain a reasonably pure waveform for frequencies up to approx, 200 kHz, it is recommended that the output voltage be set to 1 V. To achieve the low distortion factor of typ, 0.5% specified in the ICs data sheet, further calibration using a distor- tion factor meter is required. In this respect it should be mentioned that, in spite of the carefully designed board layout and the use of screened leads to and from switch S5, there is the likeli- hood of crosstalk (largely within the 1C itself) between the square wave- and sinew r ave outputs. At increased fre- quencies this results in pulse spikes being superimposed upon the sinewave signal For applications which require a minimal distortion factor the simplest solution to this problem is to short out the squarewave output, thereby remov- ing the source of the distortion. • Coarse adjustment of the output sig- nal for distortion is achieved using P5, whilst P4 provides fine adjust- ment. If no distortion factor meter is available, then setting F4 and P5 to their mid -position should give satis- factory results. * The amplitude of the triangle and sawtooth signals can be adjusted by means of P3. Sw r itch to the triangle waveform, and adjust P3 until the multimeter reads approx. 0.8 V. Of course the adjustment procedure can also be carried out using an oscillo- scope: sine: by means of P2 set the amplitude to 2.82 V pp (the equivalent of 1 V RMS) or 5.65 V pp (2 V RMS), triangle: by means of P3 set to 2,82 Vpp or 5 .65 Vpp* Frequency calibration There are basically two methods of cali- brating the function generator fre- quency scale. The first is to use a frequency counter connected to the synchronisation out- put, set PI to 100 Hz, and by means of P6 the frequency can be adjusted to correspond with the scale reading* The second method is to use the circuit of figure 9* The AC voltage of roughly 6 . * * 12 V supplied by the bell trans- former is rectified and fed via a 1 k resistor to a loudspeaker. This results in a pulsed DC voltage which has a fre- quency of 1 00 Hz, and which is clearly audible, being applied to the loud- speaker. In addition the loudspeaker is fed via a 100 ohm resistor with a 100 Hz sinew r ave signal taken from the function generator (AC output). Since these two signals add, a beat note is produced as they drift in and out of phase. By means of P6 the frequency of the function generator can then be adjusted until zero beat occurs. In only a very few cases will an absolute zero beat be produced, since both the mains- and the generator frequency are subject to periodic fluctuations. For this reason it is sufficient to reach a low beat fre- quency of under 5 Hz. N market elektor January 1 978 — 1 -47 Security lock uses identity cards An extra feature to their elec- tronic digital door lock is intro* duced by A.R.C Europe Ltd - an integral card or key reader unit. In its basic form the lock will operate an electric door strike if the correct four figure code is entered on the recessed keyboard. Arrangements are included for secret alarm signals, entry of incorrect codes, changing of codes at will and so on. With this new model the user has to first push a personal card or key into a slot in the lock and then enter a code which corre- sponds to that on the card or key. The code is different for each individual and has to be memorised. - it could for example be the figures of an important date. Cards are plastic and are coded magnetically - like bank or cash cards - and keys, about car key size, are of nylon with codes cut as notches. For extra security, however, the code in the card or key is not the same as the one entered. It is itself de-co ded by an electronic matrix before compari- son with the entered code is made. Matrix boards are plug-in units, so changing them means that from time to time people’s memorised codes can be altered without exchanging keys and vice versa. As on the basic model to allow for operator errors the lock can be set to accept one or several incorrect codes before an alarm sounds. Even when a correct code is entered the door strike is not operated until the card or key is removed from the lock, to prevent its being forgotten. Electronic controls for the locks are all on the inside wall. If an intruder were to wrench the whole lock from the wall the door would remain shut. All locks are ind ep end e ntly in st ailed Just plugging into a convenient power socket. No central electronic processor is required. Price, depending on specification is from £ 495 plus VAT. ARC Europe Ltd. Shakespeare Industrial Estate Watford Hertfordshire WD2 51ID England (620 M) Timer-counter to 35 MHz New p from Gould Advance Ltd., the TC 320 is a rugged, 5 -digit timer-counter offering frequency measurement up to 35 MHz. Extensive use of low-power C-MOS and Schottky circuitry, plus thick-film resistor networks and an openplan component arrangement, gives high reliability, easy access for maintenance and low cost of ownership. Facilities offered by the TC 320 include frequency, single-period, multiple-period and ratio measurement, together with counting and totalising. Frequency measurements up to at least 35 MHz can be easily made with the clear, 7-segment Beckman-type display. The single- or multiple-period facilities can be selected for lower-frequency measurements, and the count mode totalises regular or random events up to a 35 MHz rate. The high-impedance 1 0 mV input is enhanced by slope selection facilities and a "disciplined 7 trigger function; the display indicates zero if a signal is insufficient for correct operation of the amplifier. Input facilities include automatic gain control, with sensitivity automatically adjusted to give optimum triggering. The instrument is housed in a rugged case measuring 8b mm high x 258 mm deep x 280 mm wide, and weighs 2.27 kg. A multi-positioned carrying handle allows the instrument to stand at varying angles. A battery-powered option is available, using five rechargeable nickel-cadmium cells to give 8 hours' operation. The batteries are trickle -charged during normal AC operation. A temperature-controlled crystal option is also available, increasing the crystal accuracy from one part in 1 0 fi to one part in 10 7 and the temperature stability from one part in 10 s (0-35 C) to two parts in 10 s (0-50° C). Gould Advance Limited Roebuck Road Hainault , Essex England (618 Ml Silicon photovoltaic cells Recently introduced by NSL is a new family of Silicon photo- voltaic cells having not only good stability and high efficiency but also excellent short circuit current lineality over wide ranges of illumination. Being fully compatible with simple transistor amplifiers and eminently suitable for both power generation and light sensing appli- cations particularly at low light levels, this family of cells also feature low Leakage currents of 10 uA maximum when reverse biased by only 1 .5 volts and fast response rates of typically 8 jrs. Although these Silicon cells are generally of N on P construction, reverse polarity PN cells can be provided with, in each case, a choice of low capacitance high speed 800 material, or alterna- tively 700 type material giving higher open circuit voltages. Available in T018, T05 and 1 / 3 " diameter hermetically sealed packages, both the 700 and 800 series devices will operate over a temperature range of -60“ C to | +125°C. I Na t io nal Sent ico ndu ctors L id, \ Stamford House Stamford New Road, Altrincham Cheshire y WA 141 DR England (619 m High-voltage fast- switching power transistors RC A Solid State-Europe has launched a new range of high- voltage fast- switching power transistors designed for applications such as switch -mode power supplies and motor control from rectified mains supplies. The devices cover a range of collector currents from 3 A to 15 A, have fall times of 300- 800 ns, can sustain collector- emitter voltages up to 450 V, and will operate at up to 30 kHz. RCA Limited /Solid State-Europe Sunbury-on-Thames Middlesex, England (627 M) 1-48 — elektor January 1978 market Precision low power op-amps Precision Monolithic* have intro- duced two new precision low power op-amp* which are pin-for - pin improved replacements for the popular LM 1 08/ 308 series. GP-08 is externally compensated, while QP-12 is internally compen- sated allowing replacement of 108 type and the 30 pi capacitor. Since elimination of this capacitor is extremely important in hybrid applications, OP- 12 is offered in both packaged and chip versions. Major improvements over the LM 1G8A/308A include three times lower offset voltage drift. The total worst case input offset voltage over 55°C to + 1 25 "C is only 350 pV, In addition, the OP -08 and OP-12 can drive a 2 k ll load which is 5 times the current capability of the 108A. This excellent performance is achieved by applying Precision Monolithic*' ion implanted super- beta process and on-chip zener t rim m i.ng ca p a b ili tie s . Bourns ( Tr import Limited Hod ford House, 1 7/27 High Street, Hounslow, Middlesex TW3 1TE England (614 M) Temperature operated switch The Therm otrigger solid state temperature operated switch, suitable for operation in tempera- ture ranges between 50' C and SO 1 " 1 C, is now available from Lee Green Precision Industries Limited, This device is made from vanadium pentoxide, the resist- ance of which changes abruptly from high values at low tempera- ililrilN 4 & « 7 8 ture to low value at high tempera- ture, The temperature coefficient of resistance changes: in the pre- transition region it is about -5%, in the transition region 8%, and in the post-transition region - 20 %, The Thermotrigger is especially suitable for many temperature control applications, such a* temperature control in electronic equipment, water temperature control, fusing circuits, battery overcharge prevention etc. One especially in teres ling application is in the control of water cooling systems for petrol engines to achieve low fuel consumption. As this device is solid state, there are no mechanical contacts and L he re to re no prob lem s as so e iat cd w ith bounce, arcing and switching transients. Circuitry is simplified. Lee Green Precision Industries Limited Grotes Place , Blackheath LONDON SE3 ORA England 1613 MJ Coaxial connectors A new range of low -loss 50 n subminiature coaxial cable plug* and jacks which combine the speed and ea*e of assembly of push fit connectors with the firm retention characteristics of screw and bayonet types, is announced by Suhner Electronic* Limited, Designated QL, the connector* were originally developed for the UK Atomic Energy Authority, and have been tested and approved by CLRN in Switzerland. Nevertheless many other applications exist, particu- larly in the instrumentation and communications fields. Suitable f o r freq u e n c ies u p t o 1500 MHz and for cables with a dielectric diameter of L5 mm or less, QL connectors employ an unusual quick latch mechanism whereby three independent beryllium copper springs on the plug sleeve locate into a ring groove inside the jack socket. Disconnection is effected simply by pulling on the plug sleeve, but tension on the cable cannot disengage the connector. Since no twisting action is involved, these connectors are particularly suitable for areas where access is restricted, as well as for equipment with high packing densities. Moreover when rigid copper tube cable is used, it is even possible to insert a QL plug into its socket from a dis- tance of about 4 inches by pushing on the cable, thus facili- tating installation through instrument panels. Whereas most coax connectors are tested to only 500 mating cycles, even for military specifications, the Suhner QL range is guaranteed to 20,000 cycles due to the use of tempered beryllium copper and beryllium bronze latch springs and contacts. 4 • f ..$* * £| : : k : • ffejj If The absence of slots in the con- nector body results in a surface transfer impedance at 500 MHz of 5 mfl compared with 600 to 700 mfl for conventional slotted connectors, thus drastically reducing unwanted radiation of RF energy. Contact resistance of the screen circuit is 0,2 mO ; and that of the inner circuit is 2 mil. The Suhner range of QL connec- tors comprises straight and angle cable plugs, straight and bulkhead cable jacks, a bulkhead jack, and various adaptors and accessories. Time-saving crimp cable entries or moisture-proof pressure sleeve entries may be specified, Su h n er E lec tro nics Ltd The Technical Centre Jefferson Wav, Thame Oxfordshire OX9 SUN England 1616 M) Precision quad op-amp Precision Monolithic* have intro- duced the OP-1 1 , a precision quad op-amp w ith matched input offset voltages and matched CMRR, Individual amplifiers have input offset voltages as km as 500 u V , symmetrical slew rates in the positive and negative-go ittg direc- tions, low noise, and low drift. The OP-1 1 is ideal for precision instrumentation amplifier designs, active filters and other appli- cations needing small size and high accuracy in a single chip quad op -amp with a LM 148/ HA 4741 standard pin out. Bourns {Trim pot) Limited Hod ford House 17/27 High S tree 1 . 1 loan slo w Middlesex TW3 1TE England 2400 LSI data modem The Data Communications Division of Penril Corp. has announced the introduction of a new 24 00/ 1 200 bp s synchro nou s LSI modem offering superior performance and reliability. The 2400 LSI is designed for 2400/ 1 200 bps operation over 2- or 4-w r irc dedicated or dial net- works. The modem employs a four -phase modulation technique conforming to CCITT Type A or B and is fully on-line compatible with the Bell System 201 B or C Data Sets, most other PSK modems, as well as the Bell 801 Automatic Calling Unit, The modem features fast synchron- isation for use in multi-station polled networks and point-to- point applications. The 2400 LSI is equipped with an equaliser that is strappable in either the transmit or receive sections. Strap options are- provided for selecting transmitter output levels, carrier detect level, internal or external clock , carrier detector response time, RTS/CTS delays, and equalisation. When operating over the Direct - Distance-Dial Network, automatic answer circuits enable unattended call answering wTien connected via a Type CBS or CBT data coupler. In the Auto Answer mode an answer tone of 2025 Hz is generated for 3 seconds to switch over 801 devices or alert manual calling stations of call completion, depending upon application. Built-in local digital and analog loopback diagnostic capabilities reduce the time required to localise system malfunction. A built-in test pattern generator and receiver pattern detector greatly simplify on and off line testing and troubleshooting. No external test equipment is required to install or troubleshoot the Penril 2400 LSI. Basie modem functions are implemented in four MGS/ LSI chips, providing reduced size and increased reliability, contained on one compact printed circuit card, The modem card measures 5 inches by I 2 inches (12,5 cm by 30 cm), mounted in a free- standing enclosure. The enclosure contains an integral power supply and measures 3 -inches high by 7-3/8 inches wide by 12-5 ■■ 8 inches deep (7.5 cm by 18.5 cm by 31 .6 cm). Penril Corp L 5520 Randolph road , Rockville Maryland 20852 USA (61 5 M) (617 Ml TIL. 74 1.C.'s By TEXAS, NATIONAL, ITT FAIRCHILD Etc. 7400 14p 74 1J 30p 7432 25p 7454 15p 7490 35p 7401 14p 7414 60p 7437 25p 7450 ISp 7491 /Sp 7402 14p 7416 30p 74 J8 25p 7470 30© 7492 45p 7403 14p 7417 30p 7440 15p 74 7? 25p 7493 40p 7404 14p 7420 ISp 7441 65p 74 73 30p 7495 60 p 7405 14p 7422 70p 744? 65p 74 74 30p /496 7 Op 7406 40p 74 2 J 25p 7445 80p 74 75 30p 74103 95p 7407 40p 7475 25p 7446 85p 74 76 30p 74104 40p 7408 20p 7425 25p 744 7 75p 7481 85p 74105 40p 7409 20 P 7427 25p 7448 70p 7485 lOOp 74107 30p 7410 15p 7478 40p 74 5C ISp 7486 30p 74109 SOp 741 1 TOp 7430 15p 7451 15p 7439 ?50p 741 18 90p 7417 TOn 7453 15p 74120 y-o 74121 25p 74139 lOOp 74156 70p 74174 lOOp 74 - • 350p 74122 40p 74141 60p 74157 7 Op 74175 75p 74190 1 40p 74123 60p 74142 270p 74160 90p 74176 lOOp 7419’ 1 40p 74125 SOp 74143 770p 74161 90p 74177 lOOp 74197 1?0p 74126 50p 74144 270p 74162 90p 74178 MOp 94193 120p 74130 1 30p 74145 /5p 74163 90 p 74179 14 Op 74194 lOOp 74131 lOOp 74147 ?30p 74164 125p 74130 lOOp 74195 75p 74132 65p 74148 I60p 74165 125p 74181 200p MV3F lOOp 74135 lOOp 74150 120p 74166 125p 74182 75p 74197 lOOp 74136 80p 74151 65p 74167 325p 74184 150p 74198 165p 74137 lOOp 74153 65p 74170 200p 74185 1 50p 74199 1 85p 74138 125p 74154 120p 74173 150p 74108 350p 74155 70p SEMICONDUCTORS by MULLARD, TEXAS, MOTOROLA, SIEMENS, ITT, RCA AA113 10p HA 1 38 16p BC168R A A 1 44 10p BA 148 16p HC168C AA/17 30p BA 1 54 12p HC169C AC121 30p BA 157 ISp BC171 AC126 19p HA i 7 : ISp BC177C AC127 19p BA716 18p BC178A AC1 27/01 2Sp BA 116 16p BC182 AC 1 28 18p BA X 1 3 Sp RC187J. AC 151 2Sp BAX 16 Sp HC183 AC 153K 40p BB105A 36p BC183L AC176 70p HB110 45p BC184 ACY1 7 3Sp BC107 lOp BC1B4L ACY21 30p HC10H lOp BC1H6 AD149 60p BC10BC ISp BC705 AD 16 1 40p BC109 lOp BC712 AD16? 40p BC109C 14p BC712L AD161/7MP 90p BC113 12p •0131 AF 114 22p BC11 7 19p BC214 AF115 22p BC125 20p BC214L AF 1 16 22p BC126 ISp HC258 AF117 22p HC146 12p BC794 AF 1 18 SOp UC147 lOp BC303 A» 175 2Sp BC148 lOp BC317 AF 139 35p BC148C 14p 8C323 AF739 45p HC149 lOp HC376 ASY26 40p BC157 10p HC338 BA1 14 9p BC158 lOp HC.516 BA121 9P BC159 lOp 8C517 BC167A 12p BC547 BC54 78 80648 14p BC548C !4p BF 123 45p ISp BC549B 13p BF 125 45p ISp BC-.549C. 14p BF127 SOp 12p BC557 13p BF 166 30p 12p BCY34 SOp BF 167 25p 16p BCY70 ISp BF 1 73 25p 1 0p BCY/1 20p BF 1 79 35p 12p BCY7? ’Sp Bf 1 79C 40p 10p BD121 85p BF180 30p 12p BD123 lOOp BF 181 35p lOp BD124 85p BF 182 30p 12p BD131 36p BF 183 30p 34p BD137 39p BF 186 25p 14p BD133 45p BF 194 lOp lip BD135 40p BF 195 lOp 12p BD136 40p BF 196 lOp 12p BD137 40p BF 197 lOp 13p 80139 38p BF 198 2Sp 14p BD140 40p BF 199 2Sp 13p BD181 SOp BF 200 30p 3Sp BD182 SOp BF224 20p 55p BD707 7 Op BF225 20p ISp 80733 SOp BF241 16p 60p BD263 65p BF244H 35p 13p BDY10 1 f>0 o BF 257 26p 12p 80Y62/01 90p BF 258 26 p 60p BF 120 SOp BF259 30p 65p 12p 13p 12p BF 121 45p 8F274 BF 324 ISp 30p BF336 35p BY164 SOp OC140 BF337 3Sp BYX94 8p 0C1 71 BF 368 60o Cl 120 30p OC200 BFS94 30p Cl 164 20p 0PP12 HFW58 ?0p £’00 42p TIP29 BFX29 30p E201 50p TIP29A BFX31 lOOp E204 45p TIP29C BFX84 25p E300 47p TIP30 BFX85 30p E310 60p TIP30A BFX87 25p E420 180p TIP30B BFX88 25p £430 125p TIP30C BFY60 20p MJE340 4Sp T1P31 BFY51 20p MPSA06 2Sp TIP31A BFY52 20p OA 10 40p TIP31B BFY90 12Sp OA47 15p TIP31C BR 101 35p OA90 6p TIP37 BRY39 35p 0A91 6p TIP32A BRY58 35p OA202 8p TIP32B BSX70 20p OC23 200p TIP32C BSY40 25p OC25 lOOp TIP33 BSY95 20p OC23 75p TIP33A 8T100A 80o OC35 75p TIP33B BUI 05 150p OC42 35p TIP33C BUI 33 75p OC43 3Sp TIP34 8U208 2?0p OC45 35p TIP34A BY100 ?0p OC71 2Sp TIP34B BY 126 ISp OC72 30p TIP35 8Y127 ISp OC75 30p BY133 22p OC81 30p OC810 2Sp OC83 50p 0C84 SOp 150p TIP35A 230p 2N1305 30p 40p TIP35C 260p 2N1036 38p BOp TIP36A 350p 2N 1 308 SOp 70p TIP41A 70p 7N1711 22p 45p TIP41B 7Sp 2N2219A 25 p 47p TIP41C 80p 2K 2483 30p 75p TIP42A 80p 2N2906 16p 55p TIP47B 85p 2N2907 20p 58p TIP2955 7 Op 2N3053 20p 65p TIP3055 55p 2 43054 50p SOp T1S90 25p 2M3055 60p 55p TIS91 25p 2N3439 SOp 58p IN914 5p < N3702 lip 65p IN3754 lOp : N3703 12p 7 Op IN4001 Sp VN3704 lip 60p IN4002 5p 2N3705 12p 65p IN4003 Sp 2N3706 lip 85p IN4004 Sp 2N3711 12p 90p IN4005 Sp ">N3715 300p lOOp IN4006 6p 2N1772 175p 105p IN4007 7p 7N3819 25p 1 15p IN414B 4p < N3866 95o 150p IS44 4p 2N3904 ISp 1 15p 7N456A 90 p 2N4062 14p 1l8p 2N607 15p 2N4126 20p 145p 2N929 20p 2N6081 3Sp 225p 2N930 20p 2N5163 35p 2N1302 25p 2N6027 50p 250234 50p BZXG1 Zrnttri I 3W 6VB 1 Sp 9V1 ISp 11V Up 13V ISp ISV 15» ??V 15,. 30V 15p 39V 15p 43V 15p 47V ISp 51V 1 bp 55V 15p 68V 15p 77V 15p 160V 15p BZY38 400M W Zene i OV/ lOp 2V4 lOp ?W lOp 3 70 lOp V3 lOp 3V6 lOp 3V9 lOp 4V7 10p 5V1 lOp 5V6 lOp 6V? 1 0p 6V8 lOp /VS lOp 8V2 lOp 9V1 lOp .OV lOp 11V lOp BZ88 12V lOp 13V lOp 15V lOp 18V 10p 20 V lOp 22V lOp 27V lOp 30V 10p 33V lOp 100 Mnu-rt Zen*ft 850p BZY88 /rrurr\ 850p WESTING HOUSE THYRISTORS Type: CS21L/AC. 300 V at 16 amps. £2.50 I.C's LM301 AM 8 Pm Oil LMJOHN 8 Dil LM309K T03 l MSS5 8 Pm D.l LM556 14 Pin Oil LM/10 14 Pm D.l LM7/3 105 LM723 14 Pm D.l LM741 8 Pm D.l LM/41 14 Pm D.l LM747 14 Pm D.l LM 748 8 Pm D.l Nfc5558 P.n D.l ME 556 14 Pm D.l T AA300 I AA350 TAA550 65p 130p 190p 35p lOOp 50p 60p 75p 25p 30p ROp 45p 35p lOOp 150p 190p 50p 270p 47/50 75p .01 390p 6p .02? TAD100 250p 6p .033 200p 6p 6p .1 300p lOp .27 TBA550Q 400p 6p .33 TBA560C 400p lOp TBA720 230p 12p 1BA990Q 2S0p 20p TCA270Q 2SOp TCA270S ?S0p 6 8 MC1310P 18Sp ELECTROLYTIC CAPACITORS uF/V 4 7/75 1/16 1/25 1/50 7 7/75 2 2/35 3 3/25 4 7/10 4 7/16 4 7/25 4 7/SO 6 8/25 10/10 10/16 10/25 10/35 10/50 22/6V3 22/10 27.' 16 77/75 22/35 22/50 33/6V3 33/16 33/25 33/40 33/50 47/10 47/16 47/25 4 7/35 18p 12p 16p 18p 330/25 15p 330/35 16p 330/50 470/10 470/25 470/35 470/50 22p 680/25 25p 1000/16 25p 1000/35 26p 2200/10 78p 2200/16 35p 2200/63 6 Op 2200/1 001 20p 3300/16 4 Op 3300/25 42p 3300/63 80p 4700/25 45p 4700/40 50p 4700/63 1 20p POLYESTER CAPACITORS Mullard or Erie uF/V .001 .0022 .0033 0047 0068 5p 5p 5p Sp 5p 5p Sp 5p 5p 6p 7p 9p 12p 20p 25p 35p 40p SPECIAL OFFERS 8C147 8C148 100 BC149 ASSOR 8C157 TED 8C158 FOR BC159 £7. SOp BF 194 8F195 BF 196 BF 197 IN4148 100 fO' 150p 741 Op AMPS 100 for £20 555 TIMERS 100 for £30 RELAYS 24V 3CO H D 1 1 Pm Plug 1 N £1 00 Each ASSORTED POLYESTER CAPACITORS 100 fO' £3.00 BYX94 DIODES 1250 PIV 1 Amo 100 for £6 00 MULLARD 45O0 64V C 432 Sere* Terminal 75p SPECIAL OFFER DL 707 DISPLAYS 65 p EACH XEROZA RADIO 306, ST. PAUL'S ROAD, HIGHBURY CORNER, LONDON, N1 Telephone 01-226-1489 Easy access to Highbury via Victoria Line (London Transport) British Rail RESISTORS CARBON FILM 5% .25W 2.2 - 4 7M 2p 5 W 7.7 4 7M 2.5p 1 W 2.2 10 MEG 3.5p MULLARD POT CORES LA3 lOO-SOOKHz 75p LA4 10- 30 KHz lOOp LA5 30 - 1000 KHz lOOp LA7 10 KHz 1 00 p LA 13 For W.W. OSCILLOSCOPE 200p POTENTIO- METERS 1 K Lin 30p !• K Lm 30p 10 K Lm 30p 25 K Lin 30p 50 K Lm 30p 100 K Lm 30p 250 K Lm 30p 500 K Lm 30p 1 Meg Lm 30p 2 Meg Lm 30p 5 K Log 30p 10 K Log 30p 25 K Log 30p 50 K Log 30p 100 K Log 30p 250 K Log 30p 500 K Log 30p 1 Meg Log 30p 2 Meg Log 30p ROTARY SWITCHES BY LORLIN 1 P 1 2W 40p 2P 6W 4 Op 3P 4W 4 Op 4P 3W 40p A.E.I. MOTORS Type BP. 1303/B 100 125 V Single phase, 60 cycles H.P. 3 watts R.P.M. 1550 £3.50 C M OS 400C 20p 4001 20p 4002 4006 20p 120p 4007 20p 4009 70p 4011 20p 4012 20p 4013 55p 4015 90p 4016 55p 4017 11 Op 4018 250p 4020 140p 4022 180p 4023 20p 4024 lOOp 4025 20p 4026 200p 4027 86p 4078 155p 4029 130p 4030 60p 4032 150p 4043 ?20p 4048 150p 4047 116p 4049 70p 4050 50 p 4054 130p 4055 1 40p 4056 145p 4060 1 30p 4066 55p 4069 30p 4071 30p 4072 30p 4081 20p 4082 30p 4510 14Sp 4511 200p 4616 140p 4518 110p 4528 130p BRIDGE RECTIFIERS 1 A 50V 25p 1 A 100V 30p 1 A 200V 30p 1 A 400V 35p 1 A 600V 40 p 2A 50V 3Sp 2 A 100 V 50p 2A 200V 55p 2A 400V 60p WIRE WOUND RESISTORS BY VTM 5 K 9 Watts 10 for £1.00 IX) for £6.00 SKELETON PRE-SET POTENTIO- METERS 0.1W 50N-5M mini vert & hori? 7p 25W 100N 3.3M honz. 7p 2SW 200N-4.7M vert 8p TANTAMLUM BEAD CAPACITORS 36V; 0.1 uF, 0.22, 0.33. 0.47. 0.68. 1.0, 2.2.3.3,4 7, 6.8, 25V 1.5. 10. 16V 4.7, 10, 22.47. 10V 4.7. 15, 25. 33 Pr»ee 14p each LATE EXTRA TBA 800 lOOp TBA 641 A12 2 SOp SU7653 op 185p TAA661B 175p TAA 790 250p CA 3085 55p CA 3090AQ 400p Filter CFT 455B SOp 3817 I.C. 750p FND500 130p MM 3314 430p LM 317K 385p LM 309K 80p LM 3900N 90p TBA 120T 120p LEDS RED 125 ISp GREEN 125 ?5p YELLOW 125 2Sp RED 2 15p GREEN .2 25p YELLOW 2 25p DL 747 Duplay 200p DIL SOCKETS 8 Pin 13p 14 Pin 14p 16 Pin ISp PLEASE NOTE ALL PRICES INCLUDE POSTAGE AND V.A.T. AT 8 OR 12%% AS APPROPRIATE LARGE STOCKS OF NEONS, NUMICATOR TUBES, SINGLE ANO MULTIPHASE HIGH CURRENT RECTIFIER STACKS CAPACITORS OF ALL TYPES INCLUDING PHOTO-FLASH AND MOTOR START, TV TUNERS ALSO SOME HIGH TECHNICAL EQUIPMENT AND PARTS FOR INDUSTRIAL USERS AND SCHOOLS FOR PERSONAL CALLERS. manufacturers (large and small) we welcome your enquiries, overseas buyers/agents etc. let us know your requirements.