1 up-to-date electronics for lab and leisure eUBHTOr HI April 1977 45p / USA $ 1.50 Slotless model car track LED VU/PPM Morse decoder with DDLL(2) 4-04 — elektor april 1977 dekoder ekeHTDr Editor : W. van der Horst Deputy editor : P. Holmes Technical editors : J. Barendrecht G.H.K. Dam E. Krempelsauer G.H. Nachbar Fr. Scheel K.S.M. Walraven Art editor : C. Sinke Subscriptions : Mrs. A. van Meyel U.K. editorial offices, administration and advertising; 6, Stour Street, Canterbury CT1 2XZ, Kent, U.K. Tel.: Canterbury (0227) 54430. Telex: 965504. Bank: 1. Midland Bank Ltd., Canterbury, A/C no. 11014587, Sorting code 40-16-11, Giro no. 3154254. 2. U.S.A. and Canada: c/o World Way Postal Center P.O. Box 80689, Los Angeles, Cal. 90080, A/C no. 12350 04207. Assistant Manager and Advertising : R.G. Knapp Editorial : T. Emmens Elektor is published monthly on the third friday of each month. 1. U.K. and all countries except the U.S.A. and Cananda: Cover price £ 0.45. Number 27/28 (July/ August), is a double issue, 'Summer Circuits', price £ 0.90. Single copies (incl. back issues) are available by post from our Canterbury office, at £ 0.60 (surface mail) or £ 0.95 (air mail). Subscriptions for 1977, January to December inch, £ 6.25 (surface mail) or £ 1 1. — (air mail). Subscriptions from May to December 1977, £4.15 (surface mail). 2. For the U.S.A. and Canada: Cover price $ 1.50. Number 27/28 (July/ August), is a double issue, 'Summer Circuits', price $ 3. — . Single copies (incl. back issues) $ 1.50 (surface mail) or $ 2.25 (air mail). Subscriptions for 1977, January to December incl., S 18. — (surface mail) or S 27. — (air mail). Subscriptions from May to December 1977, $12.00 (surface mail). All prices include post & packing. Change of address. Please allow at least six weeks for change of address. Include your old address, enclosing, if possible, an address label from a recent issue. Letters should be addressed to the department concerned: TQ - Technical Queries, ADV = Advertisements, SUB =■ Subscriptions; ADM = Administration; ED = Editorial (articles submitted for publication etc.); EPS = Elektor printed circuit board service. For technical queries, please enclose a stamped, addressed envelope. The circuits published are for domestic use only. The submission of designs or articles to Elektor implies permission to the publishers to alter and translate the text and design, and to use the contents in other Elektor publications and activities. The publishers cannot guarantee to return any material submitted to them. All drawing, photographs, printed circuit boards and articles published in Elektor are copyright and may not be reproduced or imitated in whole or part without prior written permission of the publishers. Patent protection may exist in respect of circuits, devices, components etc. described in this magazine. The publishers do not accept responsibility for failing to identify such patent or other protection. National advertising rates for the English edition of Elektor and/or international advertising rates for advertising at the same time in the English, Dutch and German issues are available on request. Distribution: Spotlight Magazine Distributors Ltd, Spotlight House 1, Bentwell Road, Holloway, London N7 7AX. Copyright ©1977 Elektor publishers Ltd — Canterbury. Printed in the Netherlands. What is a TUN? What is 10 n? What is the EPS service? What is the TQ service? What is a missing link? Semiconductor types Very often, a large number of equivalent semiconductors exist with different type numbers. For this reason, 'abbreviated' type numbers are used in Elektor wherever possible: — '74T stands for pA741 , LM741, MC741, MIC741, RM741 , SN72741 , etc. — 'TUP' or 'TUN' (Transistor, Universal, PNP or NPN re- spectively) stands for any low frequency silicon transistor that meets the specifications listed in Table 1 . Some examples are listed below. — 'DUS' or 'DUG’ (Diode, Uni- versal, Silicon or Germanium respectively) stands for any diode that meets the specifi- cations listed in Table 2. — 'BC107B', 'BC237B', 'BC547B' all refer to the same 'family' of almost identical better-quality silicon transis- tors. In general, any other member of the same family can be used instead. (See below.) For further information, see 'TUP, TUN, DUG, DUS’, Elektor 21, p. 160. Table 1. Minimum specifications , for TUP (PNP) and TUN (NPN). v CEO, max 20V 'C, max 100 mA h fe, min 100 Pfot, max 100 mW f T, min 100 MHz Some 'TUN's are: BC107, BC108 and BC109 families; 2N3856A, 2N3859, 2 N 3860, 2N3904, 2N3947, 2N41 24. Some 'TUP's are: BC177 and BC1 78 families; BC179 family with the possible exeption of BC159 and BC179; 2N2412, 2N3251 , 2N3906, 2N4126, 2N4291 . Table 2. Minimum specifications for DUS (silicon) and DUG (germanium). — Tous DUG Vr, max •F, max Ir, max Ptot. max CD. max 125V 100mA IpA 250mW 5pF 20V 35mA 100/iA 250mW lOpF Some 'DUS's are: BA127.BA217, BA218, BA221 , BA222, BA317, BA318, BAX13, BAY61 , 1N914, 1N4148. Some 'DUG's are: OA85, OA91 , OA95, AA116. BC107 (-8,-9) families: BC107 (-8, -9), BC147 (-8,-9), BC207 (-8, -9), BC237 (-8. -9), BC317 (-8, -9), BC347 (-8,-9), BC547 (-8, -9), BC171 (-2,-3), BC182 (-3, -4), BC382 (-3, -4), BC437 (-8, -91, BC414 BC177 (-8, -9) families: BC177 (-8, -91, BC157 (-8, -9). BC204 (-5, -6), BC307 (-8, -9), BC320 (-1, -2), BC350 (-1,-2), BC557 (-8, -9), BC251 (-2, -3). BC212 (-3, -4), BC512 (-3, -4), BC261 (-2, -3), BC416. Resistor and capacitor values When giving component values, decimal points and large numbers of zeros are avoided wherever possible. The decimal point is usually replaced by one of the fol- lowing international abbrevi- ations: P (pico-) = 10' 1J n (nano-) = io- 9 M (micro) = 10 -6 m (milli-) = 10' 3 k (kilo-) = 10 3 M (mega-) = 10 fi G (giga-) = 10’ A few examples: Resistance value 2k7 : this is 2.7 kn, or 2700 n. Resistance value 470: this is 470 n. Capacitance value 4p7: this is 4.7 pF, or 0.000 000 000 004 7 F . . . Capacitance value 10 n: this is the international way of writing 10,000 pF or .01 n F, since 1 n is lO" 9 farads or 1000 pF. Mains voltages No mains (power line) voltages are listed in Elektor circuits. It is assumed that our readers know what voltage is standard in their part of the world! Readers in countries that use 60 Hz should note that Elektor circuits are designed for 50 Hz operation. This will not normally be a problem; however, in cases where the mains frequency is used for synchronisation some modifi- cation may be required. Technical services to readers . — EPS service. Many Elektor articles include a lay-out for a printed circuit board. Some — but not all — of these boards are available ready-etched and predrilled. The 'EPS print service list' in the current issue always gives a complete list of available boards. — Technical queries. Members of the technical staff are available to answer tech- nical queries (relating to articles published in Elektor) by telephone on Mondays from 14.00 to 16.30. Letters with technical queries should be addressed to: Dept. TQ. Please enclose a stamped, self addressed envel- ope ; readers outside U.K. please enclose an I RC instead of stamps. — Missing link. Any important modifications to, additions to, improvements on or correc- tions in Elektor circuits are generally listed under the heading 'Missing Link' at the earliest opportunity. contents elektor april 1977 — 4-05 Si Volume 3 Number 4 In the slotless model car track the infra-red LEDs are mounted on a column in the centre of the race track, while the photo- diodes are mounted on the roofs of the cars. advertiser's index 4-08 selektor 4-12 formant — the elektor synthesizer 4-14 slotless model car track (1) 4-15 Model car race tracks have been popular for many years. However, all of the commercially available tracks have the disadvantage that the car is guided by a pin which engages in the slotted track, and the 'driver' controls only the speed of the vehicle. In the Elektor race track the cars are freed from this restraint and both speed and steering are remotely controlled using an infra-red link. In the LED VU/PPM columns of up to 20 LEDs are used in a thermometer-type display. morse decoder with ddll (2) — E.H. Leefsma 4-20 The first part of this article dealt with the general operating principles of the morse decoder. In this part the complete circuit is considered in detail, together with some methods of displaying the decoded message. digisplay on board 4-29 LED VU/PPM 4-32 VU (Volume Unit) meters and PPMs (Peak Programme Meters) are used widely in the recording industry and by the amateur to monitor audio signal levels. In this circuit the conventional moving coil indicator is replaced by a column of LEDs that indicate the signal level. The simple auto slide changer senses the drop in signal level of the commentary when the commentator stops speaking for more than one second, and uses this information to control the slide projector. signal injector - J.W. van Beek 4-41 simple auto slide changer — W. Frose 4-42 hinil drive for light sequencer — H.P. Blomeyer 4-45 variometer tuner (2) 4-46 This FM receiver uses a double conversion i.f. and PLL demodulator, and offers performance equal to the best commercially available tuners. albar mk II - a b.f.s 4-54 market 4-55 up-to-date electronics for !ao anc' leisure BLiBHTDP SM AW* f.VT i / USA S 1.50 LEDs, electronics and model cars are the main ingredients of the slotless model car track. Stockists mode! cai track LEO VU/PPM Morse decoder with DDLL (2) 4-14 — elektor april 1977 formant the elektor synthesizer The availability of printed circuit boards and front panels make it a true home construction project for the reasonably skilled and sufficiently enthusiastic amateur. The module system pioneered by R.A. Moog is used — the unit con- sists of 8 basic modules that can be plugged into printed circuit board con- nectors. More than one of each module can be used in one instrument, of course. The complete project will be described in a series of seven articles, starting next month. The first part is a fairly exten- sive introduction into the subject of music synthesizers, the rest of the series don’t see why it shouldn’t!) the last explains the electronics and the mechan- instalment should be published in ical construction of the Formant. January 1978 . . . If all goes according to schedule (and we slotless model car track (1) elektor april 1977 — 4-15 Model car race tracks have been popular for many years. However, all of the commercially available tracks have the disad vantage that the car is guided by a pin which engages in the slotted track, and the 'driver' controls only the speed of the vehicle. In the Elektor race track the cars are freed from this restraint and both speed and steering are remotely controlled using an infra-red link. The proportional control system used can easily be extended and used in a radio control system for other models, as will be described in a future article. Infra-red links are now becoming well known in applications such as cordless headphones for audio use, and figure 1 shows the Siemens BPW34 photodiode and LD241 infra-red LED used in these devices (and in the car race track!). Infra-red radiation is ideally suited to short range links where the photodiode is always within ‘sight’ of the infra-red emitter. In the Elektor race track the infra-red LEDs are mounted on a column in the centre of the race track, while the photodiodes are mounted on the roofs of the cars. One advantage of using an infra-red link is that no Post Office licence is required. The Track It would be nice to make the car com- pletely independent of the track by having it battery powered, but unfortu- nately the small size of model race cars precludes this. It is not possible to mount large enough batteries to give the required speed and range, plus all the control electronics, inside the car. Instead, the car picks up its power from a special track by an ingenious contact arrangement. The underside of a car with steering servo is shown in figure 2, while figure 3 shows the track lay- out and details of the contact arrange- ment. The track consists of parallel metallic strips alternately connected to positive and negative supply rail. The contacts on the car are arranged at the apices of an equilateral triangle with a fourth contact at the centre. This arrangement ensures that at least one contact is always touching a positive strip and at least one is always touching a negative strip, whatever the position of the car. Eight diodes, shown in figure 4, ‘decode’ the voltage so that the same polarity always appears at the + and — output terminals, independent of the car position or the direction in which it is facing. The track can be made by glueing alu- minium foil to a baseboard using contact adhesive. When the adhesive is set the track pattern can be scored in the foil using two modelling knife blades with a 5 mm spacer between them. The un- wanted sections of foil can then simply be peeled off. For straight runs a metal straightedge should be used to guide the blades. Even in bends it is advisable to keep the gaps between the tracks running in parallel straight lines. This implies that if the gaps run lengthwise along one straight, they will be at right- angles to the track after the curve. If required the track can be made up in several sections to facilitate storage. To make electrical connection between the various sections each section can be equipped with a miniature two pin 4' 18 — elektor april 1977 sfotless model car track (II points A and B or A" and 3T and infrs - r6d radiation tailing upon it. A complete system. Control commands inserting a milliameter. The current resonant circuit consisting of LI, Cl from the joysticks are fed into the should be set to 180 mA maximum. and C2 is tuned to the operating fre- multiplex encoder., block 1, and thence Figure 7b shows two arrays of infra-red quency of the transmitter and rejects to the transmitter, block 2. The received LEDs mounted in a reflector housing any low frequency interference such as signal is demodulated in the receiver, to direct the radiation down onto the 100 Hz pickup from room lighting. block 5, and fed to the de-multiplexer, t rac k. Since the leakage current of the BPW34 block 6* The decoded pulses from this is small the circuit impedance is fairly are fed to the servo and drive motor high so a source-follower FET T1 is control amplifiers, blocks 7 and 8. Jnfra-red Receiver provided as an impedance converter. T2 Power for the whole in-car system is Figure 8 shows the circuit of the infra- amplifies the signal to a level suitable derived from the track via the contacts, red receiver. This is extremely simple for the FM amplifier and demodulator the diode bridge, block 3, and a stab 11 - and compact. The receiving element is a IC1. The demodulated pulse train ised power supply, block 4, The control BPW34 infra-red photodiode* The leak- appears at pin 8 of this IC h circuits will be discussed in detail in the age current of this diode varies with the Figure 9 shows a block diagram ot the following parts oi this article. M se-e text slotless model car track (1) 1 1 rm Figure 6, Showing the principle of 'digi proportional' servo control. Figure 7a. Circuit of the infra-red transmitter Figure 7b. An array of infra-red LEDs mounted in a reflector housing. Figure 8. Circuit of the infra-red receiver Figure 9. Block diagram of the complete infra-red control system. BP W 34 4-20 — elektor april 1977 morse decoder with ddlt \ 2 E.H. Leefsma The first part of this article dealt with the general operating principles of the morse decoder. In this part the complete circuit is considered in detail, together with some methods of displaying the decoded message. At the end of last month's discussion the general principles of the decode: had been established, terminating with _ description of the operating sequence on receipt of a morse message, which is controlled by the status register and decoder. Practical circuit Figure 1 shows the circuit diagram of the status decoder. The register proper consists of four D-flip flops (FF5...FF8) in which S 0 > comparator result, signal level, and are stored (S 0 and Si are the "previous status’ bits). The outputs of the flip flops are connected to a decimal decoder which supplies the eight drive signals via a diode matrix. Some outputs are active low, i.e. when the output is "0*, hence there is no "bar over these functions. The matrix diodes must be of the germanium type because the threshold voltage of silicon types is too high reliably to pull down a TTL input to a "0 s level. The IDLE-LED lights up when the decoder is idle (status condition A), and the MARK-LED lights up when the level of the input signal is * 1 \ The small loudspeaker is driven by gated bursts of clock signal and makes the input signal audible in the form of an audio tone. The drive signals obtained are used for the dot/dash/space duration logic and to feed the results into the shift register. Because ail this cannot take place simultaneously, the clock pulse generator, as mentioned earlier, is four phase (see fig. 2) and each output gives a clock pulse in turn. When the apparatus is switched on a "master clear pulse 7 is given, so that all important initial logic states are fixed and the decoder will always start from the idle position. To ensure reliable functioning of the morse decoder, the input signal applied to the input of the status register must be free from interference. This is achieved by means of the suppressor cir- cuit in the morse decoder (see figure 3). This circuit consists of a shift register and a flipflop (1C 10 and PF7); the flip- flop (FF7) forming part of the status register. At each clock pulse (T3) the level of the (digital!) input signal is moved along the shift register. The outputs of this shift register (pins 12 . . , 15) and the original input signal are fed to a binary adder (IC22) which in turn stores the most often occuring level in the status flipflop. Its output is connected to the adder, so that in fact there are six different versions of the input signal. If the majority of them are ‘1\ the output is 'I s ; if there are as many "Cfs as "Ts, the (status) result remains unchanged. Figure 4 show's the complete circuit diagram of the morse decoder. A large part of it is occupied by the dot/dash/ space duration logic. The x2 multiplication of counter 1 indi- cated in the block diagram is obtained by applying the bits of this counter (IC19 and IC20) to the selector input (IC15 and IC16) shifted one position further. The x2/3 multiplication is ■’ orse decoder with ddll (2) elektor april 1977 — 4-21 Table 1 QS Q7 Q6 Q5 Q4 Q3 Q2 Q1 1 1 1 0 1 0 1 1 1 1 1 1 1 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 1 1 0 0 0 0 1 0 0 0 1 1 0 0 0 1 1 1 □ 0 0 1 1 0 1 Q2 Q1 1 0 1 1 0 1 1 1 . 0 1 0 1 0 1 1 1 1 1 0 0 1 0 1 1 0 0 0 1 0 0 0 1 1 0 0 1 1 1 0 0 1 0 1 0 0 0 1 0 0 0 1 1 0 0 0 0 0 0 1 0 1 1 1 1 1 1 1 1 0 1 0 0 1 0 0 0 1 1 1 0 D 1 Figure 1. Practical circuit of the status decoder. The diodes used for the matrix are germanium types because they have a low threshold voltage. Figure 2. Circuit diagram of the clock-pulse generator. An external oscillator may also ce used, so the morse decoder can be adapted :□ extremely low or extremely high signalling speeds. -igjre 3. Diagram of the signal processor. In c- i^ciple six 'samples' are taken of the input s>gnal r after which the most occurring 'level' s decisive. Table 1, Showing the morse decoder output codes for tetters, digits and punctuation -narks. This table may be used to programme tne character decoder (upper half of figure 8}. Eaeh transistor corresponds to a particular ;raracter, and to turn a transistor on all the C outputs which are '1 r must be connected to base by diodes together with all the Q out- outs which are J 1\ 'sole 2. Showing which segments are lit for a canicular character. This table may be used :o zogramme the display decoder (lower "■a if of figure 8). The emitter of the appro- or ate transistor is connected via a diode to each segment-line whose logic state is '1\ morse decoder with ddtl !2 4-22 — elektor apr il 1977 master dear load rEqi S-te-i reset counters Ml . Nft - ICt = 74D4 N7 ... MIC, Nil . . Nit - \C2, IC3 = 2*7402 N15. Nie, N2B = IC9 ' 7400 N17 ■ PJ2D - ICb - 7400 N21 . . N24 " IC6 - 7400 N20 . . . N31 - 107 - 7410 M3S.. N26 ' 103 - 7420 FF13 FF16FF17. . FFJ0- tCIt, I Cl 2 =2x7475 FF5 + Ffb, FF7 + FFB, FF9* FFlO. FF»1 4 FF £2 - IC25, IC26, IC27, IC20 - 4x7474 IC4 = 741 53 IC13, 1C 1 4 . 1C 1 9, IC20 -4x7493a IC 15 IC10 2x74157 1C 1 7. iC1& - 2 x 74BS lC5t - 7414& IC22 - 74B3A 1C 1 O'. »C23, 1C 24 3*74195 D S n -Ki &- Ifi BP P a LrJ £ ; Ji > _ ► i- a morse decoder with ddll (21 elektor april 197? - 4-23 v R arpiufir LF Pulse _n Morse DATA Display shaper Decoder 9759 5 5V Figure 4, Complete circuit of the morse decoder. The circuit uses 28 TTL ICs. Figure 5. Example of a receiving chain of which the morse decoder forms part. Figure 6. This circuit can be used to convert the tones received into pulses. Figure 7. Modified alphabet for use with seven-segment displays. The text becomes quite legible after some practice. 4-24 — etektor april 1977 morse decoder with ddll < 2 i achieved by blocking the count input of counter I! one clock period after each two dock pulses (T2). This is done by means of flipflops FFli and FF12 which are connected as a divide-by-two counter* Other important parts of the morse decoder are the gates N17...N24, N29...N3L It may happen that counter I (in one of the stable con- ditions B, C, or D) counts over the maximum counter reading or that the ‘number* to be loaded into the register does not ‘fit in\ In either case decoding is no longer reliable, and the gates then ensure that the ‘overdo w-LED’ lights up as a warning that signalling is too slow or that a continuous signal (CW) is received. The output of the decoder consists of IC24 and IC23 which together form an 8 -bit shift register. At the output the letters received appear in the code which can be transferred at the time of ‘data-ready* (Table 2). Application As already explained in the first article, the morse decoder is only a link in a receiving chain. Figure 5 gives an example of a possible receiving chain. After reception the detected Tones* must be converted into pulses to drive the decoder. The circuit of such a pulse shaper is given in fig- ure 6* This particular circuit will only work reliably is the input signal is relatively ‘clean’, and preferably of fairly high frequency (above 1 kHz), There is room for improvement here! The letter code obtained can be further processed in various ways. One of the methods is a running display using seven-segment displays. Although this is certainly not the most elegant solution, reading is easy after some practice. Figure 7 shows the format of the ‘seven-segment alphabet*; Table 3 indi- cates which segments light for a given letter. Now the output of the morse decoder must be converted into the seven-segment code. There are 26 letters, 10 digits and 5 punctuation marks. In total 41 characters. The letter code is translated into the seven-segment code 1 morse decoder with ddlt (2) elektor april 1977 — 4-25 by means of two diode matrices. The first matrix has 7 inputs and 4 1 outputs, the second has 41 inputs and 7 outputs. Figure 8 shows the converter. Tables 1 and 2 show where the diode connec- tions must be made. For example: If a letter A is read out at the output of the decoder, the output code is as follows: QS Q7 Q 6 Q5 Q4 Q3 Q2 Q1 111110 10 To turn on the 'A* transistor all £ QV --ch are * 1 ’ and all fi QY w r hich are l V i-t : onnected to its base via diodes, Fne following diode connections are :hus made in the first matrix: Q2 Q3 Q4 Q5 Q6 Q7 Only if letter A is received, will the base of the first transistor be positive. In the second matrix the connections must now be made which ensure that the right segments of the displays light up; in this case all segments except T (Table 2), On the output of the last matrix the information is now available to drive a display. In practice a read-out with only one display will not be very intelligible. Therefore figure 9 gives a method of extending the read-out with several displays. The letter received is made visible on the first display, and during the data-ready pulse the data is stored in the first memory, so that the letter appears on the second display. The Figure 8. Two diode matrices convert the code at the output of the morse decoder into a letter code for the seven-segment displays. Figure 9. Circuit diagram of a running-script display. 4-26 — elektor april 1977 morse decoder with ddU 0 / Mt I® Parts list for figures 4 and 1 1. Resistors: R1 . . . R8 = 4k7 R9,R1 0 = 330 H R1 1 r R1 2,R1 3 = 1k5 R1 4 . . . R17, R24 . , . R26 = 3k3 R18 P R21,R22 = 1 fc R 1 9 = 150 n R20 - 47 H R23 - 100 n R27 = 220 O R28 . . . R36 = 1 a Capacitors: Cl - 560 n C2 - 1 0 ji/6,3 V C3 , . . Cl 3 - 100 n Semiconductors: T1 - TUP D1 . , . D23 = DUG D24,D25,Q28 = LED D26,D27,D29 = DUS I Cl * 7404 IC2JC3 - 7402 IC4 = 74155 IC5JC6JC9 = 7400 IC7 - 7410 ICS = 7420 IC1 GJC23JC24 = 74195 IC1 1 ,IC1 2 = 7475 IC13JC14, ICl9 r lC20 = 7493A IC1 5 JC1 6 ~ 741 57 IC17 r tC1S - 7435 IC21 = 74145 IC22 - 7483 A 1025 . . . SC28 = 7474 Sundries: S = Switch SPOT R |_ * 1 00 „ , . 200 n loudspeaker mm Figures 10 and 11. Component layout and p.c. board for the circuit of figure 4 IE PS 9759) 1 . Mote, that D19 is not mounted f the same way round' as its neigh* hours! Some wire links are required on the copper side of the board: 'a' is connected to V, 'b' to V, etc. Tl «HMu '-*6 8S *88 0{r3b}0 liMUl sag ■Bn mm IS Ijli : : ' IS II : * “ : t§ 1, « ^ J1 I IS EXT. IN u. mi- i m Si® r3i ';T' CLOCK ! NT. OUTPUT rnfM *1 iT: m % EXT. m ilAM 9759 - 12 Figure 12* External connections to the board. Figure 13. Suggested power supply, it can be mounted on the TV tennis 5- volt supply board (EPS 921SB). Table 3. The ASCII code (American Standard Code for Information Interchange). 4-28 — elektor april 1977 morse decoder with ddll '2 MARK ^ IDLE OVERFLOW NPUT morse decoder with ddll (2) digisplay on board elektor april 1977 —4-29 letters are constantly shifted further so that ‘running-script’ results. As already said, this form of read-out is intended only as an example. High signalling speeds require a large number of displays to keep the message received legible An alternative solution would be to use a teletype or ASCII display. The latter may be of particular interest, and the code is shown in Table 3. The ‘double diode matrix’ principle of figure 8 can be used. Construction A printed circuit board layout is shown in figures 10 and 11. For reasons of economy, a double-sided board was not used, so a large number of wire links are required. The long links with ‘0’ marks at each connection are the supply lines; these are mounted last as ‘overhead wires.’ ft is advisable to use solder posts and interconnect them with insulated wire links. A further set of (insulated!) wire links is required on the copper side of the board. These interconnect points ‘a’ and ‘a’, ‘b’ and ‘b’ etc. As with all TTL circuits, it is advisable to mount the unit in a screened box. The external connections to the board are shown in figure 12. A suitable power supply is shown in figure 13. The IC should be adequately cooled; the heatsink should have an area of at least 40 cm 2 (6in. 2 ). M The ‘Digisplay’ circuit described in El 3 (May 1976, p.538) has proved to be more popular than we had expected. Whenever the working model was dem- onstrated at radio shows we received a large number of requests for a design for a printed circuit board. A suitable board design is shown in figures 1 and 2. Since the unit will normally be used as a more-or-less self- contained measuring instrument, a suitable power supply was included on the board. The circuit of this supply is shown in figure 3. The fuse and trans- former are not mounted on the board, of course. Photo 1. This logic tester displays the states of sixteen binary signals (either ‘O' or '1') simultaneously, in the convenient form of a 4 x 4 matrix on an oscilloscope screen. 4-30 — elektor april 1977 digisplay on boarc i ■ ■W me C5 ho OiB o oooooo ■ 'v-. Rt5 -Q Ol& -0 O 14 R14 R^3 1C 7 tJCUUUUU A BCD o o o o PH n ia f o 0ia |mT}o source could be recorded on to tape o: disc with no reference to the signal level. The only adjustment necessan would occur during playback, when the volume control of our (perfect) ampli- fier would be adjusted to give a sounc level acceptable to our ears. Regrettably the real world is very imperfect. To begin with, the lowest signal level that can usefully be recorded is determined by the noise generated b> the recording and/or transmitting medium itself in the absence of a signa. In the case of disc this is caused by the surface texture of the disc material, in the case of tape, by random orientation of the magnetic domains. At the other extreme, the maximum signal level is determined for tape by the level at which the tape ‘saturates’, i.e. when the signal level versus magnetisation graph becomes non-linear, thus distorting the signal. In the case of disc the upper limit is determined by the tracking ability of the cartridge and, to a lesser extent, that of the cutting lathe, I The electronic circuits used to process the signal also have upper and lower signal limits due to noise on the one hand and distortion due to clipping on the other. In a re cording/ re producing situation however, the limits imposed by the recording medium are much narrower than those imposed by the electronics. This brings us to the concept of dynamic range, which is simply the ratio between the highest usable signal level and the lowest. This may be expressed as a ratio, e.g. 1000:1, but is more usually expressed in dB (decibels). There is nothing mystical or magical about the use of decibels, a voltage ratio expressed in dB is simply 20 logic Vi/V 2 . Thus the dynamic range of 1 000: 1 expressed in dB would be 20 log 1000. Log 1000 - 3 so the dynamic range - 60 dB. It is most important to note that this tells nothing about the magnitude of the quantities involved, merely their ratio. If some absolute quantity such as a voltage is to be ex- pressed in dB then a reference voltage must first be chosen against which to express it. Thus, for example, the mains voltage (250 V) could be quoted as 60 dB (reference 0 dB = 250 mV). There are many reasons for using dB rather than simple ratios. Firstly, human sense organs tend to respond to stimuli in a logarithmic fashion, so if electronic instruments such as signal level meters are scaled in dB then their response will be similar to that of the human ear (disregarding frequency response). Secondly, when dealing with dynamic ranges of many thousands of times, it would be impossible to construct an instrument with a linear scale that could read both the lowest and highest signal levels without changing ranges. Thirdly, since ratios expressed in dB are simply logarithms, to express the product of several quantities such as the overall gain of, a number of amplifiers and/or led vu/ppm elektor april 1977 - 4-33 attenuators In cascade it is necessary merely to add up the dB gain/attenu- ation of each stage rather than multi- plying the straight ratios. This makes life much simpler. Meter Characteristics This may seem something of a di- gression from the why’s and wherefores of signal level meters, however, from the foregoing it is possible to make the following conclusions, 1, Signal level meters are required for the following reasons: a. To set the limits of the signal within the dynamic range of the recording/ reproducing chain, b t Wh e n m aking mu It ic ha n n e I r e co rd ings (including plain stereo) to set the balance between the different channels. This is particularly important in re- cordings studios using multi-mike techniques and umpteen-channel mixing desks, c. The dynamic range of live sound is much greater than can be accomodated by the recording medium. It is around 120 dB, determined on the one hand by the softest sound that can be heard, and on the other by the threshold of pain. Since a tape or disc might have a dynamic range of only 50 to 60 dB it is necessary to turn up the recording level during quiet passages so that the signal is not masked by the noise, and to turn the level down during loud passages, so that the equipment is not overloaded. Of course automatic compressors can be used in this situation, but manual adjustment by a skilled engineer is frequently less noticeable. 2. From the earlier discussion it is evident that a signal level meter should have a logarithmic (dB) scale. PPM or VU? Since the average value of a symmetrical AC waveform is zero, before the signal level can be measured with any sort of DC meter it must first be rectified. If the rectified waveform is fed to a DC voltmeter then, due to the mechanical inertia of the meter movement, the meter will tend to indicate the average voltage of the waveform. So that the meter needle does not tend to follow the waveform at low signal frequencies (w T hich would make it difficult to read) the inertia of the meter must be fairly high. This is the basic principle of the VU meter. However, a meter that is sufficiently well damped for the needle not to jitter at low frequencies will also be unable to repond quickly to short transients. This means that the meter will tend to underestimate peaks, Figure 1. Showing the principle of full-wave rectification of the audio signal, and the attack and decay time constants. Figure 2. Complete circuit of the rectifier section of the meter. Figure 3. Showing how, in a VU meter the voltage on C4 builds up until equilibrium is reached between charge and discharge. Figure 4* For half-wave 'VU' measurement the inverting half of the rectifier section may be omitted. 4-34 - elektor april 1977 leading to overrecordiug, and this has been a criticism of the VU meter. That this criticism is to some extent justified is verified by the fact that cassette tape recorders, which almost invariably use VU-like meters, have lately begun to sport peak indicator lamps that light when the maximum safe recording level is exceeded by a few dB, even on short transients. Anyone possessing one of these cassette decks will have noted that when making live recordings, especially of speech, the peak indicator lamps can be made to flash merrily on and off with the VU meters barely leaving their end stops. This simple experiment indicates that ‘live 1 sound frequently has only a low average level* but a much higher peak level he. a very large peak to mean ratio. To give a truer indication of the peak signal level the BBC (and other) PPMs (Peak Programme Meters) were devel- oped. These instruments were designed to respond very quickly to increase in signal level, but for ease of reading the meter reading was made to die away at a much slower rate. To put it another way, the PPM has a short attack time constant and a long decay time constant. The original PPMs had an extremely fast attack. However, later experiments in psycho-acoustics showed that the ear is unable to perceive distortion caused by short duration clipping (e.g. less than 10 ms). This means that overrecording of short transients is allowable, provided no undesirable effects occur such as blocking of amplifiers or rX bursting, and in this situation the original PPM tended to overestimate the effect of peaks. Accordingly the attack time constant was modified, and the present (BBC) standard is an attack time constant of 2.4 ms and a decay time constant of 1 s. the ballistics of the meter movement play a great part in determining the time constants and are very tightly specified. However, the present design is to use a line of LEDs instead of a moving coil instrument, so the meter characteristics are determined entirely by the elec- tronics. Figure 1 shows a block diagram of the drive circuit ry for the PPM. The signal is first buffered and if necessary amplified before being fed to two rectifiers. The signal to one rectifier is inverted so that this rectifier operates on negative half- cycles of the waveform, and a full- wave rectified version of the waveform appears at the junction of the outputs of the two rectifiers. This output is used to charge a capacitor C through resistor R a , the time constant R a * C being the attack time constant. C can Figure 5. Complete circuit of the LEO display with logarithmic divider chain. This gives a DlN-type PPM scale (Table 4) but cart be adapted to give a BBC-type PPM (Table 3 1 or VU scale (Table 2). Meter Drive Circuits In the conventional VU meter and PPM led vii/ppm eisktor april 1977 — 4-35 discharge through R a and R|^. R a , Rb and C are chosen to give the time constants mentioned earlier. To avoid the output from C being loaded by the following circuits a high input ;mpedance buffer is provided. Operation of the VU meter drive cir- cuit is very similar, The only difference is that Rfr is very small compared to R a . so the attack and decay time constants are virtually identical. Figure 2 shows the complete circuit of the meter drive. IC1 functions as a non- inverting amplifier with a gain of R2 + R3 —= — . Bootstrapping of Rl at the K2 junction of R2 and C2 increases the in- put impedance. The reverse series connection of C2 and C3 is necessary since the DC voltage on pin 2 of IC 1 can be either slightly positive or slightly negative. A 22 ju reversible electrolytic could also be used here. 1C 2 functions as a unity gain inverter, and the inverted and non-in verted sig- nals are fed to 1C3 and IC4 respectively, which form the two halves of the rectifier circuit. The rectifier circuit is of the active type, and eliminates the effect of the forward voltage drop of the diodes, which could otherwise cause gross errors in the measurement of low signal levels. The rectifier configuration shown offers some interesting possibilities, depending on how diodes D3 and D4 are connec- ted. With the diodes connected as shown the circuit operates as follows: assuming that the circuit is in the quiescent condition the outputs of IC3 and IC4 are at zero volts. Since there is no bias voltage across D3 or D4 1C3 and IG4 have no feedback and are thus operating open-loop. On positive half- cycles of the input signal the output of 104 will swing rapidly positive until it clips. D2 will conduct, T1 will conduct and C4 will charge rapidly from T1 via R9 t IC5 is connected as a voltage follower, so its output will be equal to the voltage on C4. This provides negative feedback to the inverting input of IC4, The voltage on C4 will thus stabilise at a level equal to the peak value of the input voltage. During this time the output of IC3 will swing negative, forward biasing D4 so that IC3 simply operates as a voltage follower following the (negative) output of IC2. D1 is thus reverse-biased, so no negative voltage appears on the base of TL On negative half -cycles of the waveform the situation is reversed. The output of IC3 swings positive and causes DJ and 11 to conduct, while the output of 104 goes negative, reverse -biasing D2. The discharge path for C4 is through R9, RIO and Rl 1, since 1C5 presents a very high input impedance and thus has little effect on the discharge time. The ratio of RIO to Rll is chosen so that the voltage on C4 cannot cause reverse emitter base breakdown of TI in the absence of a signal SI can be opened to remove the dis- charge path and provide a 'peak memory’, which can be useful when making 'dry-runs’ of recording sessions to find the maximum level of the signal This ‘Elektor Standard’ system provides a peak rectifier with an extremely fast rise time. However, to make a rectifier with characteristics corresponding to that of a BBC or DIN PPM or VU meter, several changes must be made. If the circuit is to be used as a VU meter then T 1 is not required since the attack and decay times are relatively long (300 ms). The charging current of the 4*36 — eiektor aprif 1977 led vu/ppm 1 Table 1 . Component values for different response characteristics. Eiektor PPM DIN PPM BBC PPM VU METER R9 R10 Rll C4 270 n Sk2 18 k 10 ju< lJ 68 n 10 k 18 k 22 loo n 12 k 33 k 22 100 k o n 10 k 330 n D3 D4 1N4148 1N414S 2 x 1N41 A 2 x 1N41 A ■8 reversed 18 reversed 1 x 1 N41 48 reversed 1 x 1N4148 reversed Tl BC549C or equivalent omitted 111 Tantalum 20% tol. 16 V. capacitor is smaller and can easily be provided by the outputs of IC3 and 1C4 direct. RIO is replaced by a wire link, R9 becomes 100 k, Rll -10k and C4- 330 n. Diodes D3 and D4 are reversed. This may seem a little strange, but the circuit now operates as follows; on positive half cycles the output of SC4 will swing positive in an open-loop mode until D3 begins to conduct, after which 1C4 will operate as a voltage follower. The voltage at the output of IC4 is thus equal to the input voltage plus the forward voltage of D3. D2 is forward biased and its voltage drop cancels that of D3, so the voltage at its cathode is equal to the input voltage to IC4. C4 will charge from 1C4 through D2 and R9. On negative half-cycles the output of IC3 will swing positive and C4 will charge via D1 and R9. D4 discharges through R9 and Rll, but since Rll is small compared to R9 the attack and decay time constants are similar. The problem now arises as to how to calculate the required time constants. The specification for a VU meter states that if a 1 kHz sine wave, whose amplitude is such as to give a steady state reading of 100, is suddenly applied, the reading on the VU meter shall reach 99 within 300 ms. If it was a simple step function that was applied to the meter then the time constant would be simple to calculate. Unfortunately, since the meter is dealing with a full-wave rectified sinewave 04 does not charge continuously, but only when the rectified voltage is present. In the ‘gaps 1 between it actually discharges, as shown in figure 3, The steady state reading is reached when the capacitor is dis- charging at the same rate as it charges.. This makes the calculation of the required time constant rather compli- cated and by far the easiest way to find it is empirically. Alternative drive circuits Many cheap so-called VU meters use only half wave rectification (figure 4) and assume that the signal waveform will be symmetrical The circuit of figure 2 may be so modified by omitting 1C2, 103 and their associated com- ponents. This does result in a slight saving in cost, but the meter does not have a true VU characteristic. Since a conventional VU meter is simply a rectifier instrument relying on the meter ballistics for its attack and decay times it is fairly easy to simulate electronically. However, the attack time of a conventional PPM is determined by two factors, the meter ballistics and the electrical time constant of the rectifier circuit. This makes it more difficult to simulate accurately without using a more complex double time constant circuit. Some compromise must thus be found. The standard for a BBC type PPM {BS 4297 ;1 968) states that if a 5 kHz signal (whose steady state value would give a reading of mark 6 on the PPM scale) is applied in a 5 ms hurst the reading will be 4 dB below mark 6. To obtain this reading the attack time constant must be about 2.5 ms. Since the attack of a genuine PPM is more complex this n ece ssa rily re suits ins ma 11 errors at other scale points. The decay time constant of the BBC PPM is 1 s, and the decay time is largely indepen- dent of the meter ballistics, since these are swamped by the long time constant. The specification for a PPM to DIN 45406 states that a 5 ms tone burst shall give a reading 2 dB below the steady state reading. Here an attack time constant of 1.5 ms is a reasonable compromise. The decay time of the DIN PPM is 650 ms. Since the attack time constants of PPMs ire much shorter than that of a VU meter the value of R9 and/or C4 must be reduced. To obtain a value of R9 that could be driven direct from the 741 outputs C4 would need to be less than Ip, which means that Rll would be greater than 1 M. This is unacceptable as the input currents of ICS would then have a significant effect on the decay time. To overcome this Ti must be reinstated so that smaller values of R9 and Rl! and a larger value of C4 can be used. However, the base emitter junction of Tl then introduces an extra voltage drop, which must be eliminated. The solution is an extra diode in series with both D3 and D4. To clarify matters the component values for the ‘Eiektor Standard 1 PPM, BBC and DIN PPMs and VU meter are given in Table 1. These values may of course be altered to suit personal taste and/or any other standards. Display Having obtained a voltage which in- creases and decreases with the appropri- ate time constants in response to a signal, the question now is how to convert it to a suitable display. Estab- lished readers of Eiektor will be familiar with the ICs manufactured for this purpose by Siemens. These consist of a chain of analogue voltage comparators each having one input tied to a refer- ence voltage from a potential divider, the other input being connected to the signal voltage and the output to a LED. When the input voltage exceeds the reference voltage the comparator output goes low and lights the LED. The higher the input voltage the more LEDs in the chain there are lit. Unfortunately the response of these LED voltmeters is linear, whereas a logarithmic response is required for a signal level meter. One possibility would be to precede one of the Siemens ICs by led vu/ppm elektor aprit 1977 -4-37 Table 1 . Component values, to modify figure 2 for different response characteristics* Figure 6. Printed circuit board and com- ponent layout for the rectifier section {EPS 9419-1). Parts list for figures 2 and 6: Resistors: R1,R3= Sk2 R2' lJ = 390 H R4(2)_F!5(2) = 10 k R6<2) = 4k7 R7^,R8 = 2k7 R9 ~ see text and Table 1 RIO = see text and Table 1 R1 1 - see text and Table 1 Pi ( l) = 1 Q0 k P2 - 1 0 k Capacitors: C1 (l1 - 100 n C2(ll,C3 - 47,11/16 V C4 = see text and Table 1 C5 f C6 r C7,C3 = 47 n Sem [conductors: 1C1.IC2 121 , IC3l 2l ,IC4,IC5= 741 Dl(3)_D2 = 1 N41 48 or 1N914 D3 D4 ld = see text and Table 1 Ti 13) = BG109C, BC549C or equivalent Miscellaneous: SI* i * 3 * * = DPST switch Notes: 1 : See 'Calibration". 2: Omitted for half-wave VU measurement, 3: Omitted for VU measurement. i logarithmic amplifier. However, these :an be somewhat critical in operation, and a much simpler solution Is to design a LED voltmeter using discrete com- p arators but, unlike the IC versions, having a logarithmic potential divider network. . he complete circuit for a stereo version of the indicator is given in figure 5, The 'law 5 of the logarithmic potential divider is shown in the second column of Table 4. It will be seen that around the 0 dB level the steps are 1 dB, but at lower levels the steps are made much coarser (5 dB). This means that the meter has the best resolution in the most useful part of its range. This corre- sponds to the scaling of a DIN PPM, but can easily be altered to give BBC PPM and VU type scales as will be explained later. Construction A printed circuit board and component layout for a two-channel version of the rectifier circuit are given in figure 6, and for the display unit in figure 7. The 4-38 — elektor april 1 977 lad vu/pprn Table 2. Components for VU type display scale. Resistors: R21 - R28- 0 ft {wire links) R29.R31 ,R33,R 55 = 1 k R3G,R53 = 270 ft R32 * 100 n R34,R48 H R6Q = 00 R35 = 820 ft R36 r R44,R50 = 68 ft R37,R43.R45,R47 = 470 ft R38,R49,R51 = 330 ft R40 = 390 ft R41.R57 - 560 ft R42 h R 58= 56 ft R46 - 33 ft R52 = 22 ft R54 = 47 ft R56 = 120 ft R67 = 18 0 R59 = 820 ft R61 — R 84 and R 61 ' — R64 J = omitted Semiconductors: IC20JC21 (K1 — K4 and K1 f - K4 H ) = omitted 021 - 024 and D21 J - D24' - omitted Alt other components as per figure 5. Table 3. Components for 'BBC PPM' type display scale. Resistors: R21 — R30 = 0 ft {wire links) R31 - 3k3 R32 = 2k7 R33 - 1 k2 R34,R37 = 560 ft R35 = 1 k R36,R4Q r R43 = 270 ft R38 = 39 ft R39 r R41 - 330 ft R42 - 68 ft R44 r R46 - 33 ft R45 = 220 ft R47 = 180 ft R48,R54 = 22 ft R49 = 150 ft R50,R52,R56 r R58,R60 = 0 ft R51,R53.R57,R59- 1 20 ft R55 = 100 ft R61 - R65 and R61 r - R65 J - omitted Semiconductors: IC20JC21 (K1 - K4 and K1 r - K43 = omitted D21 - D25and D21 J - 025 r = omitted All other components as per figure 5, 1 ? s accompanying photographs show the construction of the complete unit. Of course it is not mandatory to mount the LEDs on the back of the p.c. board as shown. If panel width is at a premium the LEDs can be mounted separately on a piece of vero board and the p.c. board can then be mounted edge on to the front panel. Calibration Having checked for any wiring errors, before connecting the rectifier and display boards a few simple checks can be carried out. Firstly, apply power to the display board and check that, with the input grounded, no LLDs are lit, and with the input connected to +15 V all LEDs are lit. Secondly, with power applied to the rectifier board, adjust P2 in both channels to give a null (zero volts) at the output of IC5. It may be necessary to swap around the 741s to find the one with the best offset characteristics, so it is a good idea to mount them in sockets. Having carried out these checks the two boards may be Linked together and the whole system tested by applying a sine wave signal to the inputs and varying the input level. Calibration of the meter led vu/ppm elekt^r april 1977 — 4-39 Parts list for figures 5 and 7. Resistors: R20 = 560 H R21,R23,R25- 820 H R22 ~ 270 H R24,R34 = 1 50 ll R26.R46 - 47 n R27, R29, R43 = 680 S2 R28 - 82 H R30 = 0 H (link) R31,R33,R45 = 390 H R32,R47 = 220 H R35 = 470 12 R 36, R 60 = ISO R37,R39 = 330 n R38,R44 = 100 12 R40 = 56 H R41,R61^ - RSO^ = 1 k2 R42 = 1 SO n R48 = 27 H R49 - 1 20 12 R5G, R55 = 18 H R51 = 68 12 R52 = ion R 53 = 39 n R54 - 407 R56 - 608 R57 - 1 2 12 R58 = 108 R59 = 207 Capacitors: C20 - 1 00 m/25V Semiconductors; IC20 - IC29 ( K 1 — K2Q r ) = LM324 D20 = 1 2 V 400 mW zener D21 - D25 ~ red LED^) D26 - yellow LED^ D27 — 040 = green LED^ Notes: 1 : See 'Adaptions' and Tables 2 and 3 < Note that, for the stereo display, R61 to R80 and 021 to 040 are duplicated. I C20 to I C29, R20 to R60, C20 and D20 serve both channels. mm tjik v' 1 :;. viy- 7 •. . W ; - m ™<4r n . m ■ w*. •M* Jrmmr ■s. . ” . m* • JhkW WOW;. W l MU) — \v.. ML 'r . v ■il ** i i 1 1 £ 5 £ % I \ 4:.;: * 11 J* i ? . f* *c • S :i | M 1 -i If § 4 m f h\. depends on the maximum signal level likely to be encountered in the system. In any event, to calibrate the PPM version the unit must be fed with a sine wave signal having a peak value equal to the maximum peak signal level. PI is then adjusted until all LEDs up to the 0 dB LED are lit. To calibrate the YU meter it must be remembered that the average value of the signal is being measured, so the peak value of the input signal must be tt/ 2 OI- LS? times higher, if the half-wave 'VlT circuit is used then, since only half the waveform is utilised the peak value of the input signal must be n times greater. To put it another way, if the same signal level is to give a 0 dB reading on both PPM and VU measurements then Pi will need to be turned up further in the VU meter. If the meter is to be used to monitor the Figure 7. Printed circuit board and com- ponent layout for the display (EPS 9419-2). Table 2, Component modifications for VU* type display. Table 3, Component modifications for BBC PPM- type display. 4'40 - elektor april 1977 tad vu/ppm output of a high signal circuit such as a power amplifier then there is no need for 1C 1 to have any gain and it can simply be used as a voltage follower. In that case C2 and C3 are omitted, R2 is replaced by a wire link, PI becomes 10 k and Cl becomes 1 ju. Adaptations 1 . The scaling of the meter is —50 dB to +5 dB in accordance with DIN 45406 and the IEC proposal of September 1970. However, the display p.c* board is almost universal in scope and can be altered to give virtually any desired scale. This is achieved simply by altering the resistors in the potential divider chain and/or the number of compara- tors and LCDs. The modifications required to give a VU-type scale (-20 to +3 dB) are given in Table 2. Modifi- cations to give a BBC PPM -type scale are given in Table 3, 2 . The time constants of the rectifier circuit can be adjusted to suit personal taste. There arc various ‘standards' for peak reading meters, none of which agree with one another. If one wishes to stick to a particular standard, however, then the appropriate time constants should be u^ed, The appropriate display scaling should also be used. 3 . The current through the LEDs (and hence their brightness) can be varied by choosing different values for the series resistors. However the resistors should not be chosen so low that the current rating of the LEDs is exceeded. 4 . The +15 volt supply must be well decoupled by a 100 ./i/35 V capacitor to avoid transients appearing on the supply line due to the large current drawrn by L I and the LEDs on transient signal peaks. The +15 V supply should be capable of delivering 500 mA. The — 15 V consumption, however is only 5 . Since 741s have a limited slew rate 6, The LED colours may be changed to the h.f. response of the rectifiers may be suit personal taste, but it is a good idea somewhat limited, since for the first to arrange that the display colour part of their output swing the 741s are changes to indicate an overload e.g. operating open loop. Perfectionists may green below 0 dB, yellow at 0 dB and like to experiment with faster op-amps red above 0 dB t N such as the 748 or 531. Such lCs, of course, require external frequency com- pensation, so would-be experimenters are advised to consult the manufac- Table 4. Scale layout of the three types of about 20 mA. turer's data. Table 4. PPM Level (dB) 'DIN PPM' type scale parameters. PPM Level Absolute volta 9 e LED (HR1 applied to display input (volts) D40 -50 .021 D39 -45 .033 D38 40 .067 D37 -35 .12 D36 -30 .21 D35 -25 .38 □34 -20 .67 D33 -15 1 ,2 032 -10 2.1 D31 -J5 3.8 D30 -4 4.3 D29 —3 4.8 □ 28 -2 5.4 D27 “1 6 D26 0 6,7 D25 +1 7.6 D24 +2 8.5 D23 +3 9.5 □22 +4 10.7 D21 +5 12 'BBC PPM' type scale parameters. VU type scale parameters. LED dB Level PPM reading Absolute level at display input (volts) □40 -20 unspecified 0,12 D39 -14 1 0.24 D38 -11 1.5 0,34 D37 -8 2 0.48 D36 -6 2.5 0.6 D35 -4 3 0.75 D34 -2 3.5 0,95 D33 0 4 1.2 D32 +2 4.5 1.5 D31 +4 5 1.9 D30 +6 5.5 2.5 029 +8 6 3 028 +1 1 6.5 4.26 027 +14 7 6.01 □26 +20 unsDecified 12 LED D40 □39 □38 □37 □36 D35 D34 D33 D32 D31 D30 □ 29 D28 D27 D26 D25 VU Level ”-20~ -16 -10 -9 —3 -7 -6 —5 —4 -3 —2 -1 0 +1 +2 +3 Absolute level at display input (volts) 0.85 1,51 2.68 3.01 3,38 3.79 4,28 4.77 5,36 6.02 6.75 7.57 8,5 9.53 10,7 12 signal injector elektor april 1 977 — 4-41 J.W. van Beek Many people consider the use of signal injectors to be something of a 'brute force' method of faultfinding. However, a signal injector can easily be carried around in the pocket and is often the service engineer's 'first line of defence' when undertaking repairs in the customer's home. Certainly a signal injector is much more portable than an array of sophisticated signal generators. Parts List Resistors: R 1 „R2,R5,R6 - 10M R3 - 100 k R4 = 470 O R7 = 27 k PI — Ik preset Capacitors: Cl = 100 4/6 V C2,C3 - 470 n C4,C5= 100 p C6 = 100 n/250 V (see text) Semiconductors: IC1 = 401 1 T1 = TUP T2 r T3 = TUN D1 ,D2 = DUS {see text) D3= LED (e g. TIL209) Miscellaneous: SI = SPST switch 4 x 1 A V mercury batteries Most of the cheap signal injectors on the market produce a square wave output at about 1 kHz. Since the squarewave is rich in harmonics extending up into the Megahertz region these are useful for testing rX circuits, as well as using the fundamental for audio testing. The signal generator described here differs slightly in that the 1 kHz square wave is keyed on and off at about 0.2 Hz, which makes it easier to trace. Figure 1 shows the complete circuit of the signal injector. The keying oscillator consists of an astable multivibrator built around two CMOS NAND gates N I and N2. This switches on and off TL, which drives a LED to indicate when the signal is on. The 1 kHz squarewave generator also consists of an astable multivibrator, which utilises the tw r o remaining NAND gates in the 401 1 package. This astable is gated on and off by the first astable. The output of the 1 kHz oscillator is buffered by transistors T2 and T3, the output being taken from the collector of T3 via a potentiometer PI which serves to adjust the output level. The maximum output is approximately equal to the supply voltage (5.6 V). Diodes DI and D2 provide some protection for T2 and T3 from external transients, and C6 isolates the circuit f r o m an y DC volt age in the circuit under test. If the signal injector is to be used to test circuits having high voltages present, especially mains (e.g. television sets) then C6 should be rated at 1000 V working, in which case it will be too large to mount direct on the p.c. board,, the layout for which is given in figure 2. If is also a good idea to mount the complete circuit inside a box made of insulating material, especially when working on live chassis equipment such as a television set. Dl and D2 should be types capable of handling any transient voltages and currents likely to be encountered. Power for the circuit can be provided by four 1 ,4 V mercury batteries. The exact type of battery chosen is up to the individual constructor, but should neither be so small that the battery life is too short, nor so large that the complete instrument is too bulky. M ■I =s7mA f V 4*1.4 V Cl R2 1 nr.i i |T N2 Fit 1 DM IQGJU RS i DM N1...N4 = IC1=CD4011 simple auto slide changer 4-42 — elektor april 1977 meW u seiu ' s used to j. 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Big jigl | MB :|| iilp S p'i!: : 2 p.x n.a* :ii;t g j L^ i - x A V £ ■ ' X m m-m ■ ■ -x-:- • : ‘ y -x ■ Ifi « Mm ■v>: ....y.. v-x •• Xfi x.. R .RR . W ^ m M •f :R- . X- ..... •- X P ..xx . : RR . • • v : -v • • afr. v a •• - , ; B"’# :: 4'.-* ^4 #: ii, %. --rx ■ v.X : ,Jl S ' » P ti-S:" ■ |m.* ,r-j hi nil d rive for light sequencer elektor april 1977 — 4-45 (H.P.BIomeyer) Readers may remember that the J Disco! ights r featured in Elektor September 1975 used TTL shift registers to drive triac switches. This is fine so long as great care is taken with layout and supply decoupling, otherwise noise from the triac switches can interfere with the sequencing of the TTL shift registers due to the poor noise immunity of the latter, i he circuit described in this article shows an application of a new family of High Noise Immunity Logic (MINI L) from Siemens. This has the advantage (in this type of application) that the logic thresholds are much Higher. The switching speed is also lower than that of TTL, so the circuits are much less likely to be affected by short transients. The circuit of the light sequencer is given in figure 1 , Clock pulses are provided by an oscillator based on a Schmitt trigger, which readers will no doubt have encountered before in TTL circuits, PI controls its frequency, ST2 buffers the output of the oscillator. Since the HINTL shift registers can sink more current than they can supply the circuit is arranged to operate from a negative supply voltage so that negative* going pulses can be used to trigger the :riac switches. It will be noted that the 0 V rail of the sequencer is connected to the neutral side of the triac dimmer shown at the :op left corner of the circuit. Thus when in output of the shift register is high I (he. at 0 V) the triac has zero gate voltage and the lamp is extinguished. When an output of the shift register goes low (he, dow r n to —12V) the triac receives a negative gate voltage and the lamp lights on negative half-cycles of the supply waveform. Note that since the triac is controlled only on negative half-cycles the lamp receives only half power and is thus not operating at full brightness. This should not really be a disadvantage for the intended appli- cation. The operation of the sequencer is as follows: at the start of a sequence the outputs of all the shift registers are high and all lamps are extinguished. The serial input of IC1 is tied to -I2V (he, logic 0) so on each clock pulse a *0' will be loaded into the serial input, The four shift registers are connected in cascade and as the 0’s shift through the lamps will light in turn until all are illuminated. On the clock pulse after the last lamp is illuminated pin 6 of IC4 will go low, causing the output of N4 to go high. The output of N4 is connected to the 'parallel load' inputs of 1C l to IC4. so the data on these parallel inputs will appear on the outputs. Since the parallel inputs are ail connected to 0 V they are at logic 1 so l’s appear on all the outputs and the lamps extinguish. The whole cycle then repeats. At any time the sequence may be reset manually by SI, which inhibits the clock generator and makes the output of N4 go high. Buffer stages N1 to N T 3 are included on outputs pin 12 of IC1 to fC3 as other- wise these outputs would have to drive both a triac and the serial input of the next shift register, which would exceed their fanout. Extensions This, of course, is simply one example of what can be done. It is quite possible to extend the circuit to control many more lamps by increasing the number of shift registers. The only point to remember is that the fanout of N4 is limited to 30, and the input loading of the parallel load input of an FZJ161 is 4, so N4 will drive only 7 shift registers. This can be overcome by connecting additional gates in parallel with N4. Other sequences can also be devised, limited only by the ingenuity of the constructor. Some useful suggestions are given in the previously mentioned a rti cle 4 Dis c o ligh ts ' . Warning Since the positive side of the sequencer power supply is connected to mains neutral the entire circuit should be treated as having mains voltages on it. The unit should be mounted in an insulated box, or if in a metal box should be isolated from the box and the box earthed. PI should have an insulated spindle and none of the metal parts of PI should be accessible. Final note The Siemens triac type TXC 01140 may not be readily available. Any other reasonably sensitive triac may be used, however. H Figure 1, Circuit of the Light Sequencer, Each shift register output drives a triac switch (one only shown!. 4-46 eiektor april 1977 manometer tuner {p@iif^ 2j Having dealt with the construction of a mono FM receiver using the Variometer front-end, the second part of this article describes the design of a stereo tuner. This tuner uses a double conversion i.f. and PLL demodulator, and offers performance equal to the best commercially available tuners. As mentioned in the first part of this article, to provide a satisfactory signal- to-noise ratio in stereo, a PLL demodu- lator must operate at a low i.f. This ensures that the ratio of the VCO frequency deviation to VCO free- running frequency is as high as possible, so that phase noise in the VCO does not degrade the s/n ratio. Since the i.f. output of the variometer front-end is at 10.7 MHz this means that a second frequency conversion must be performed to convert this to a much lower intermediate frequency such as 450 kHz, hence the term ‘double- conversion'. There are, on the market, double conversion PLL FM 1.1. systems integrated into a single 1C. However, it was felt that considerably superior performance could be obtained using a PLL made from discrete components (though admittedly containing one 1C) such as the OTA PLL used in the mono receiver described last month. Ibis also means that any readers who have built the mono receiver can, if they wish, extend it to a stereo version without wasting the OTA PLL already built. Block Diagram As the simplified block diagram of figure 1 shows, the mono tuner is converted to the stereo version simply by inserting the new i.f. converter and 450 kHz i.f. amplifier between the Vario tuner and the existing OTA PLL, and by adding a birdy filter and stereo- decoder to the PLL output. The OTA PLL must, of course be slightly modified to operate at 450 kHz. Figure 2 shows that this 'simple 1 circuit is somewhat more complex when examined in more detail. The i.f. converter is the section enclosed by a dotted line. The 10.7 MHz output of the Vario front-end is first converted down to 450 kHz by mixing with a second local oscillator signal at either 10.25 or 11.15 MHz. Either of these frequencies (iow T or high frequency heterodyning) will give a difference frequency at the mixer output of 450 kHz. Of course, other unwanted products also appear at the mixer output, so the mixer must he followed by a bandpass filter that allows through only the wanted 450 kHz signal. The filtered signal is then fed to three cascaded limiting amplifiers. The outputs of these three stages are detected (rectified), summed and used to drive a signal strength meter. Its principle of operation is as follows: at low signal levels none of the three amplifiers will be limited. An increase in signal will cause an increase in each amplifier output, which will cause an increased reading on the meter. Most of the drive to the meter will, of course, be provided by the third stage, since this has the greatest output. Once the third stage has limited it will make no further contribution to an increase in the meter reading, and most of the incremental meter drive will be provided by the second stage. This in turn will limit, so that only the first stage will be contributing to an increase in the meter reading with increased signal strength. The meter will thus start off at low signal levels by giving a large increase in the reading for a small increase in signal strength. Once the third stage has limited it will require a much larger increase in signal strength to give the same increase in meter reading and so on. In other words the sensitivity of the meter will reduce as the signal level increases, that is to say it will have an ap p r o xim a t e ly 1 o gari th mic r esp on se . This is an ideal type of response for a signal strength meter, which may have to deal with a change in signal strength of around 100 dB (100,000 times), since it allows small signals to be seen without overloading on large signals. The meter drive circuit also provides the drive to a muting circuit that switches off the output signal from the OTA PLL when the signal level falls below a certain value, which can be varied by means of the 'mute leveT potentiometer. This means that interstation noise is not heard when tuning between stations. The output of the third limiting amplifier is fed to the OTA PLL, which provides further gain and also demodu- lates the signal in a similar manner to Figure 1. Simplified block diagram of the double conversion stereo tuner. Figure 2. Block diagram of the stereo tuner showing details of the i.f. converter. Figure 3. Circuit of the i.f. converter. L ■R l variometer tuner elektor april 1977 - 4-47 270^H 10 QK BC177 3k 3 © CD 401 1 GND 1 0n '00n 100. i 5FE107M A 4 5 2?0ljw BF494 2k 2 squelch level BC107 *see test X_tal RC107 N1 . . . N4 - CD 4011 320.VH J2 P bOi^A converter Variometer i ! n — | — ^ Front end 4-H J- ! - MIX % A Lim XTQ DET DET DET Squelch level Decode r OTA RLE 4bU kHz ObT Stereo filter 944 7 J 4-4B - elektor apri! 1977 variometer tuner that described last month, the only difference being that the OTA PLL is now operating at 450 kHz instead of 10.7 MHz. The demodulated output from the PLL is fed back into the muting section of the i.f. converter, then out again into a ‘birdy’ filter. Because of the large i.f. bandwidth required for stereo reception, and the close channel spacing (200 kHz) in the FM broadcast band, the i.f. stages cannot be relied upon completely to suppress adjacent channel interference, which is apparent as warbling and chirruping noises - hence 'birdies 1 . The only way to suppress these is by a low pass filter, cutting off at 55 kHz, after the output of the demodulator. The output of the birdy filter is then fed to the stereo-decoder, which utilises the well-known 1310 TC. Complete Circuit of the i.f. Converter Figure 3 shows the full circuit of the i.f. converter. The output of the Vario front-end is fed to a ceramic filter, which provides some 10.7 MHz selec- tivity. The mixer stage is built around Tl, while a crystal controlled oscillator built around T2 provides the oscillator signal, which is injected into the base of T 1 via C 1 . The crystal can be either a 10,25 MHz or 11.15 MHz type, either being equally suitable. Capacitor C x is not required in every case, and should be added only if the oscillator fails to start without it. It should not be added if the oscillator functions without it since this could lead to too high an oscillator amplitude w r ith consequent production of unwanted harmonics in the oscillator waveform. The mixer output is filtered by L1-L3 and C7-CI 0 before being fed into the first limiting amplifier. The limiting stages are somewhat unusual in that they make use of CMOS logic gates as linear amplifiers. These were chosen because they are extremely cheap, if biassed into their linear region they make perfectly adequate amplifiers, and they have excellent limiting character- istics, Three of the gates in a 4011 package are used as limiting amplifiers, the fourth being used in the muting circuit. Parts list for figures 3 arid 4, Resistors: R1 ,R7,R8 = 330 n R2,R4,R1 4,R 1 6,R 1 8 f R23 - 4k7 R3,R 5 = 33 k R6.R9,R1 1,R20,R22 = 2k2 R10 = 1 k R12 = 1 k5 R1 3,R1 5,R 1 7 = 100 k R19 h R 24 = 15 k R21,R25 = 10 k R26 = 390 n R27 P R28 = 3k3 Capacitors: Cl = 22 p C2,C19 = 10 n C3= 180 p C4,C5,C14 = 100 n C6 = 56 p C7.C10- lOOp C8.C9 = 1 50 p C11 P C13 P C15,C16,C17 H C18 = 1 n 012 = 470 n C20 = 47 ju/1 6 V C21 = 47 m/10 V C x - 100 n (see text) Semiconductors: Tl ,T2 = 8F 494 T3.T5 = BC 107, BC 547 T4 = BC 1 77, BC 557 N 1 . + , N4 = CD 4011 D1 r D2,D3 = AA 1 19 D4,D5 = 1N4148 Miscellaneous: Ll r L3 = 270 mH R.F. choke L2 = 820 mH R.F. choke PI = 2k2 !in r pot. P2 = 100 k I in * preset SFE 10.7 MA or CFSA 10,7 ceramic filter 10.25 MHz or 11,15 MHz XTAL * 25 kHz 50 mA meter The output of the third limiting stage (N3) is taken out via Cl 9 and R23 and clamped by D4 and D5. Since only a small input signal should be applied to the OTA PLL for optimum operation, the signal is attenuated by R25 and R26 before feeding to the PLL. The output of the PLL is fed back into the muting switch, comprising T5 and R24. When T4 is off the signal passes through R24 unhindered. When T5 is on the signal is shorted to ground. The outputs of the limiting amplifiers N1 to N3 are rectified by Di to D3 and used to charge Cl 4. In the absence of a signal the voltage on Cl 4 will be below the threshold level of N4, so the output of N4 will be high. T3, T4 and T5 will be turned on and the signal from the PLL output will be shorted to ground by T 5 , The muting level potentiometer PI provides a variable bias on the anodes of Di, 02 and D3 via R14, R16 and R18. This puts a preset voltage on C 14 in the absence of a signal, wdiich determines how T much additional signal will be required from the limiting amplifiers before the threshold of N4 is exceeded variometer tuner and T3, T4 and T5 will turn off, allowing the signal to pass. For example, with no preset voltage on C 1 4 the output signal from N3 would have to equal the threshold level of N4 plus the forward voltage of D3 before the muting switch would open. With the voltage on C14 preset to just: below the threshold of N4, on the other hand, very little signal would be required to switch N4. Note that with PI turned fully clockwise (max, muting level) the slider is grounded. The signal strength meter is also connected to C14 via P2, which adjusts its full-scale deflection. However, if the negative end of the meter were connected to ground it would register the preset bias on C14* even in the absence of a signal. To avoid this the negative end of the meter is taken to the slider of PI, so that in the absence of a signal the positive end of the meter is at more or less the same potential as the negative end (except for the forward voltage of D1-D3). The meter reading will thus be almost zero, in fact very slightly negative, in the absence of a signal. Printed Circuit A printed circuit board and component layout for the i.fxonverter are given in figure 4 and require little comment. The only point is that 10.25 and 11.15 MHz crystals are likely to be relatively expensive, and it may be worth trying 30.75 MHz ± 75 kHz or 33.45 MHz - 75 kHz third overtone crystals, which are available fairly cheaply on the surplus market. OTAPLL The modified circuit of the OTA PLL is given in figure 5, and the printed circuit board and component layout are given elektor apnl 1977 -4-49 in figure 6. The following changes for operation at 450 kHz should be noted: Rl , R24 omitted PI, now 2k2 or 2k5. P2, omitted. Can be left in circuit if already built. Cl 1 , now 2n7. 014, now 82 p. C l 5, omitted. Figure 4. Printed circuit board and com- ponent layout for the i.f. converter (EPS 9447 21. Figure 5. Modified circuit of the OTA PLL. { N 4148 1 220 B 1-494 BF494 2*1N4148 33 3? b BF494 C2 LI Y~Y^y ( : I c R ■ y R23 1( Of)n 1 N4'48 BF494 BF494 BC 177 BC 557 o BF494 BF494 R 1 R I Ui t R2 l Qj.' ^ ]QV i BC 107 BC547 ^2G' R2?p"! FfN Ol 3447 hh 4?^ RIB IfiV IC 1 C A 3080 3? 4-50 — elektor april 1977 variometer tune' Ram list for figures 5 and 6. Resistors: R1 ,R24 = omitted R2,R4,R11 H Rl2,R23 = 4k7 R3 H R8 H R 1 5,R 1 7,R 13 r R1 9,R2G - 1 k R5 r R6 = 560 H R7 = 100 n R9,R10,R23 = 10 k R13 P R22 - 220 k R 14 = 47 k R16 = 68 k R21 = 470 k R25.R26 = 2k2 R27 = 100 k PI = 2k2 (2k5) P2 = omitted {wire link) Capacitors: 01,02,06,010- 100 n C3 = 470 6 V C4 P C5- 22 n C7 - 47 m/16 V C8,C13 - 470 n C9,C1 5 = omitted Cl 1 = 2n7 Cl 2 - 47 m/ 10 V 014 - 82 p Cl 6 = 10^/10 V Semiconductors: T1 . * . T7 = BF 494 T8 = BC 107, BC 547 T9 =» 8C 177, BC 557 01 ... D6 - 1N4148 IC1 - CA 3080 Miscellaneous: LI - R.F. choke 470 mH Birdy Filter Established readers of Elektor may recognise the birdy filter of figure 7 a as being the same as that used in the ‘Feedback PLL Receiver 1 featured in Elektor No 9. The only difference is that the function of the input resistor R9 is taken over by the collector resistor of T5, R24 on the i.f. converter board. C2 can be used to improve the stereo separation by adjusting the filter to give zero phase shift at 38 kHz. This can be done using a signal generator and dual beam scope, or if this is not available by tuning in to Aunty BBC's stereo test transmissions and adjusting by ear. Stereo Decoder With the exception of the 7400, 4011 and 741 the MC1310 must surely be one of most overworked IC’s, and here it is again used in the stereo decoder* The circuit has been described so many times that an operational description will not be given here, just the setting up procedure. The best method of setting up, if a frequency counter or calibrated oscilloscope is available, is to set the frequency of the VCO to 76 kHz by monitoring the 19 kHz (!) output at pin 10 and adjusting PI . Otherwise tune to a strong station transmitting in stereo and adjust FI until LED D1 lights (SI must be open). Turn PI left and right until it extinguishes, then set it to about the middle of the range over which it is lit. Printed Circuit Board Figure 8 gives a p.c. board and com- ponent layout for the birdy filter and stereo decoder. To allow for the possi- bility of using this p.c* board with alternative i.f. systems a position for R9 has been included on the board. For use with the double conversion PLL this position should be occupied by a wire link. Power Supply A simple stabilized supply for the mono tuner was given last month. Since the stereo tuner operates from a 12V supply this must be modified as shown in figure 9* The current consumption of the tuner is less than 100 mA, so the stabiliser should be fed from an unregulated 1 7 V (approx) DC supply capable of providing this current. Alternatively an IC voltage regulator such as the TBA625B or similar 12 V type could be used* Another alternative is the power supply used in the ‘local’ radio described in Elektor 22. Pilot Tone Filter The 1 9 kHz pilot tone of the 1310 stereo decoder may not be sufficiently well suppressed for some applications. Those with supersensitive ‘earholes’ may be able to hear this tone, and what is worse, when used with a tape recorder the 38 kHz subcarrier may beat with the bias oscillator, giving some very nasty noises* Fortunately, Toko manufacture a filter designed to remove these offending signals, which has two deep notches at 19 kHz and 38 kHz, combined with a gradual rolloff at high frequencies* It does all this with hardly any detriment to the bandwidth of the audio signal, as can be seen from table 1 (only 1 .2 dB down at 1 5 kHz). Wiring Diagram Figure 10 shows a complete interwiring diagram of the stereo variometer tuner* To avoid any problems this should be adhered to, though the physical disposition of the boards may be varied variometer tuner elektor april 1977 — 4-51 * see text 0 O — ||^hi 470n Figure6. Component layout and p.c. board for the OTA PLL (EPS 6029). Figure 7a. The birdy filter. Figure 7b. Circuit of the stereo decoder. Tablet, Data and pin connections of the BLR31Q7 pilot tone filter. Table 1 BLR 31 07 Parameters Attenuation at: 15 kHz 1 .2 dB max. 19 kHz 26 dB min. 38 kHz 50 dB min. Input impedance 3 k Output impedance 4k7 Insertion loss at 1 kHz IdS mpuL i - — ■ output 1 ground -O pnH -Q input 2 — — - -O Q— output 2 9447 T 1 to suit the chassis used. Connections between the aerial socket and front-end, front-end and iX converter, i.f. converter and OTA PLL should be made using 75 £2 coax cable, preferably miniature for ease of bending round corners etc. All low frequency signal connections, including the connection to the muting level pot., should be made using screeened audio cable. The i.f. converter should preferably be screened from the other units by mounting in a box of steel or tinplate. Tuning Scale Figure !1 shows a suggested tuning scale. If a 45 mm diameter drive drum is used on the Variometer spindle then this icale is full size. For larger or smaller diameter drive drums the drawing must be scaled up or down to suit. Alignment procedure For those who wish to build the stereo :uner ‘from scratch 1 without first building the mono version the complete alignment procedure (including that for :he front-end) is given here. 1 . Connect the tuner input to a good aerial system and the left and right audio outputs to the tuner or auxiliary input of a stereo amplifier. 2. Turn the muting level control fully anti-clockwise so that the interstation noise is unmuted, 3. Set PI In the OTA PLL to mid- position. X Set the core of L6 in the front-end to give maximum noise. 5. Set the tracking of the front-end by adjusting the core of L5 to tune to 87 MHz with the pointer at the left- hand end of the tuning scale, and by adjusting C21 to tune to 104 MHz with the pointer at the right-hand end of the tuning scale. If no stations can be received at these frequencies then find the two stations which are furthest apart on the tuning scale and adjust L5 and C2] to set them to the correct points on the scale. L5 and C21 interact so several successive adjustments may have to be made. 6. Tune to a (preferably weak) trans- mission around 90 MHz*, and adjust the cores of L3 and L4 in the front-end for maximum signal and/or minimum noise, 7. If possible tune to a weak trans- mission around 100 MHz and adjust C5 and C8 for minimum noise. 8. Repeat 5, 6 and 7 until no further improvement can be obtained. Note that a special trimming tool is required to reach L4. If such a tool is not avail- able, the core of L3 can be removed and L4, L5, C8 and C21 are trimmed. Then the core of L3 is inserted and L3 and C5 are adjusted. 9. Tune to a very weak (noisy) trans- mission and adjust the core of L6. in the front end to obtain minimum noise and/or maximum signal. 10, Adjust the free-running frequency of the VCO in the OTA PLL until the capture range of the PLL is symmetrical about the i.f. frequency. To do this turn PI first clockwise, then anti-clockwise until the signal disappears, then set PI to the middle of this range. 11, Tune to a stereo transmission and set PI in the stereo decoder so that the LED D1 lights. Find the range of settings of PI over w T hich D1 will light then set PI to the middle of that range. * If an FM signal generator is available then the alignment procedure can be carried out using it. A strong broadcast transmission can be turned into a 'weak' one by using an aerial attenuator or by substituting a short piece of wire for the good aerial system. 4-52 — elektmr april 1977 12. Using the BBC stereo test trans- missions, of which details are available from the BBC, adjust C2 in the birdy filter to obtain maximum channel separation either by ear or using a millivolt me ter or oscilloscope connected to the amplifier output. When a signal is present only on the right channel C2 should be adjusted for minimum signal on the left channel and vice versa, 13. Tune to the strongest signal in the FM band and adjust P2 in the i.f, converter to give full-scale deflection of the signal strength meter. 14 . Finally, if desired, insert a 10-60 p trimmer in the position marked C x in the front-end and trim for minimum noise on weak stations. variometer tuner Final IMotes The output from the 1310 stereo decoder is nominally 100 mV when connected to the output of the OTA PLL. This output level was chosen to obtain minimum distortion from the 1310, but it may be a little on the low side for some amplifiers. Those who require a higher output and are willing Parts list for figures 7 and a. Resistors: R1/R2 = 470 k R3 - 330 n R4,R5,R6,R 1 0 = 15 k R7,R 1 3,R 1 4 = 4k7 R8 = 2k2 R9 = see text R11,R12 = 1 k R1 5 P R1 6 = 100 k R17 = 10 k R 1 8 = 27 k mi 60 p Capacitors: Cl H C1 3 = 470 n C2 = trimmer 10 . . . f C3= 150 p 04,05,06 = 120 p C7 = 47 p C8 - 100 n C9 - 2q2/16 V 010 - 47 n 011 - 470 p Cl 2,01 4 = 220 n C15,C16 ~ 10 n 017,013= 1 0 ju / 1 6 V Cl 9 = 47 p/1 6 V Semiconductors: T 1 h T3 = BO 107 B, BC 547 B T2 => BC 177, BC 557 01 * LED 1 01 = MC 1310P Miscellaneous: PI = 4k7 1 i n . pot SI = SPST switch Figure 8. Component layout and p.c. board for the birdy filter and stereo decoder (EPS 9447-3) . 7 v t + 1N4148 3Cbb7 BCB47 BC 327 H4iT 9 Figure 9. The power supply used in the mono tuner, modified for use with the stereo tuner. Figure 10. Complete wiring diagram of the stereo tuner. Note that the power supply shown is that used in the 'local' radio (Elektor 20) which is equally suitable for this design. i am m HIM ■ & 'mm I 4 ■ • ^^pia KW 4 i, i H dpi i;:: m i IlSll m v, ::: ' ” 1 m m 1 milfi Hi 1 wmm Ipl: ■ V ?: N:ii wm je Jiilj mm variometer tuner elektor apriJ 1977 — 4-53 12 V RMS 100mA 4-54 — elektor april 1977 variometer tuner albar Figure 11. Suggested tuning scale layout. to tolerate a slight increase in distortion can increase the decoder output by feeding in a larger signal from the PLL, This is achieved by increasing the gain of the PLL output stage T8/T9. To do this R24 and Cl 5 are reinstated in the circuit. CIS should be 100 ju 6 V and R24 can be between 470 £2 and 2k2 3 determined experimentally. Note that the load impedance at the output of the decoder should be not less than 22 k. 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(hi > 3 +J 4-i O I-- > • r-q hM rt ,; 3 VS (-1 0 a pi 05 1 I § s g; > CJj CJJ T3 O 3 3 • r— ■ - — . r— W* fr &S ^ « PS J? qj > > ^ « fll 3 5 -a fe 3 c 3 > n 4.0 1 1 g| ■t! g U-. c/5 c V * u «.§Sy u 4 u H a 8 o “Of d> ^o'o o e 3 3 O 43 T3 3 3 M 0 ) y> 3 ■pj Lh ■j> 05 3 Im 0 c 8 ya ^ MM p Mh 3 [fi U5 ^ U O i) O a> o > 3 33 S -3 3 ^a 3iT3 T3 O a; o a> h» yj P3 U5 S-. o [h. o u 3 Q. 3 O O > O - - 2 — i n > ir 3 13 ^ d w — •3 IH ^ 3 rt 4 j L y M> o 3 l-H y> 3 D A aj CJ cj 05 flj *MM O MM o o (m QJ XI 73 X SJ X U G> 3 T3 bfl ■s £ 2 2. mm cr I A o £ CJ CJ 3 n £ b MM « X >. G o u X K 3 s .a 05 a> ’p 3 X £ 05 MM 3 3 O fjj X &0 « 3 pj >“ h +- x X - t> x > P3 73 “J .42 3 O 05 ca a pa e CJ CJ _ 3 o "S x p ^ £ a> sfl Jn E a> 05 o> OJ x o ^ Mm O CL> 3 market elektor april 1977 -4-55 antiphase signal and, with rise and fall times of less than 0*5 jrsec, switching currents up to 500 mA and operating voltages of between 4,5 volts and 27 volts, these contactless micro switches, manufactured by RAFI, will directly interface with all data logic systems. Designated CM1, 2 and 3, these three sizes of micro switch arc mechanically interchangeable with standard mechanicalversions, have an operating temperature of -25°C to +70 a C, ate available with either push-on or p.c.b, terminations and comply with the application and environmental requirements of classes H$F (DIN 40 040) and IP40 respectively. ESS Ltd., 98 Croydon Road , Penge, SE20 7AB t London 1425 M) new T family of low f current, low voltage. Gallium Phosphide LED indicators which arc fully IC compatible and have fast switching characteristics* Three types of indicators arc available, with the BIM 21 LED being fitted with a choice of two polycarbonate lens styles which are available in red, green, or amber, and give a viewing angle of 55"* The BIM26 and 56 LED indicators utilise 4 and 5 mm dia, light emitting diodes respectively which sit forward from the front of the body thus obviating the need for a lens, and giving a viewing angle of 80°. This new Lumotast 75C range which is manufactured by RAFI, is suitable for both IP4G and DIN 40 040 Class applications, has 0.7 mm by 0.5 mm wire wrappable leads and will operate over a temperature range of -25'eto +70" C. Tire Button tops, which are manufactured in Makralon and available in either neutral, red, yellow, green or blue, can be supplied in either round, square or rectangular design. EES Ltd., 98 Croydon Road, Penge f SE20 7AB, London Prototype Logic Wiring System The Wire Distribution System from Zartronix is a well-proven point-to-point wiring system* Wiring is carried out using a wiring pencil w hich carries 36 SWG solderable synthetic enamelled copper w r ire. The wire is routed and kept in place on the prototype board by special plastic combs. Connection to IC pins is made Availability of polyester capacitors Compstock is now stocking a full range of Erie radial-lead polyester metallised film capacitors, series 51016* gUUfl&g New contactless microswitches Three new ranges of contactless microswitches which utilise Hall Effect IC’s, and thereby provide true solid state, bounce free switching, have recently been introduced by EES Ltd. These long life, high reliability micro switches require a maximum operating force of less than 1 .2 N and, under constant environ- mental conditions have a life in excess of 10* operations and a switching point repro duceabiiity of ± 3 /im. This series is manufactured in four ranges which are designed to handle working voltage ratings of 1 60 volts D.G, 250 volts D.C., 400 volts D.C* and 630 volts B.C They have a capacitance range of 0.010 to 10.0 microfarads and are stocked in ± 10 percent and - 20 percent tolerance. The capacitors will operate over the 'emperature range of -40 to ■^85 B C and have a dissipation factor of less than 0.002 when measured at 1 kHz at 25" C Solderability is to BS9G76 and the capacitors are dipped in a iesin-based solvent resistant coating. The capacitors are available either with standard straight leads or to order with !eads preformed. Compstock Electronics Ltd., -2j44 Bowlers Croft r Basildon, Essex simply by wrapping the wire two or three times around the pin then, when the board is finished, completion of connections is effected using a soldering iron. The application of heat and solder melts the enamel and makes a permanent joint. Zartronix, 115 Lion Lane, Basle mere, Surrey 1434 m All three types have high luminous intensity figures of between 1.2 and 1.8 med at 10 mA, power dissipation of up to 115 mWat 25 X and will operate over a temperature range of -55 to +10G°C Having 23 mm long wire wrappable leads, and being a push-fit into an 8 mm dia. hole, these low cost LED indicators are eminently suitable for both development projects and production equipment. BOSS Industrial Mouldings Ltd., Biggs Industrial Estate, 2 Herne Bill Road , London, SE24 OAU, England Humiliated contactless pushbuttons Recently announced by EES Ltd - ^ new range of illuminated ■ _ hbuttons incorporating a Hall effect chip from which a i itch output via two open : : .lector transistors operating New family of LED indicators Recently introduced by BOSS Industrial Mouldings Limited is a The output is derived from two open coEector transistors which generate a static uniphase or 4-56 — elektor aprjl 1977 market DIN 400 50/F54. Swiss SEV 500 V 1 0 A and Canadian CSA 300 V A*C approvals also apply. Highland Electronics, 8 , Old Steine ; Brighton BNI IEJ , Fast Sussex (424 Ml Lloyds approval for Highland switches The heavy-duty oil- sealed range of switches, (coded series 04), marketed by Highland Electronics Ltd., have received Lloyds Approval under Certificate No. 847 (Rotterdam). This range of switches, introduced in 1975, comprises push buttons, indicator lights, rotary actuators and keylock actuators. Panel mounting is by 22,5 mm diameter circular hole and a modular flex- ible design is employed so that the actuators, lamp block adaptors and snap-on contact blocks, with toggle or butt action mechanisms, are easily interchangeable. The contact blocks, up to four on one actuator, are rated at 10 amp 500 volt A.C. max., and are available in normal toggle action or slow make - slow break versions. Two pairs of normally open contacts, two pairs of normally dosed contacts or one normally open and one normally dosed contacts per block may be specified. In addition to the recent Lloyds approval, making this range of switches eminently suitable for marine use, the actuators are oilproof and waterproof and conform to IEC 144/IP65 and New power transistors Recently introduced by Jermyn Distribution are 3 pairs of NPN silicon power transistors, continu- ously rated at 5,8 and 15 amps. Each pair within this series has a sustaining Vceo of either 300 or 400 volts DC at a case temperature of 25° C and power dissipation figures of 100, 125 and 175 watts. SSOiSSA # ' % § % S: ' 0 WE |! 4 aft # s •m m &. , ** M s®. -4 m 4 ■ iiiSi.lt ' vtf:-. -W v:¥: .• «&■ .•$& m : . $&S- «». ' •••••. » ,. 4 : .m ' ■ppa PF' * ' - Sk ' ' ; U. •/ '% With DC current gains in excess of 60 these 2N6542-7 devices are ideal for high speed power switching in inductive circuits where fall times of less than 800 nsecs are imperative. Capable of operating over a temperature range of — 65 C to +200°C, and housed in a rugged hermetically sealed TO 3 package which gives low thermal resistance and high reliability, all six devices are particularly suitable for 115 and 200 volts line operated switch mode applications such as switching regulators, PWM invertors, motor controls, as well as solenoid and relay drivers. Jermyn Distribution, Sevenoaks, Kent Availability of metal oxide resistors Compstock is now stocking the full range of Erie M04 and MOS miniature metal-oxide resistors. These series, which are available off-the-shelf, have BSI and Post Office Approval and can be released to BS9111-N002 or DQAB 6/49 as appropriate. The resistors are suitable for use in avionic, military and professional industrial applications. The M 04 is rated at % w att and is available in El 2 values from 10 ohms to 270k ohms, the M05 being V 2 watt also available in E12 values from 10 ohms to 330k ohms. Both ranges are stocked in ± 2 percent tolerance, Soldcrabiiity is to BS201 1 and marking to BS2488/BS1852. The M04 is Post Office style 9 IF and the M05 style 9 IE, Compstock Electronics Ltd,, 42/44 Bowlers Croft, Basildon, Essex (432 Ml ;-,xv. My. ® Jm ^ 1 » best commercially available bipolar transistors. The price of the NE 24483 is $85 (100-999 quantity) and is 'considerably cheaper* for higher quantity orders. California Eastern Labs, One Edwards Court , Burlingame , CA 94010 (428 Ml Portable V.C.F. function generator Dana Electronics have announced that their associate company, Exact Electronics, have added a new low-cost v.c.f. portable model to their range of function generators. Model 119P is a very compact instrument that can be either mains operated or run from 12 volts D.C., supplied by either its own rechargeable batteries or a car dashboard socket. Tire in-built batteries will power the 119P for up to 8 hours. The instrument offers a dynamic frequency range from 0.02 Hz to 2.2 MHz, wdth sine, square, triangle and variable time symmetry of all waveforms for ramp and pulse operation. A v.c.f. input is provided to allow the generator to be varied either up or down over a range of 1000 : L Minus 10 volts D.C. will increase the frequency by three decades from a minimum multiplier setting, and plus 10 volts D.C, will decrease the frequency three ■fttsssfr 7 tfwrasj*: A Mtaggsz — -M "■ W W- .... w M. . m. m Hi*, ■00 ...*: " New GaAs FET in low-cost package The availability of a new ? gallium arsenide fleld-effeet-transistor (GaAs FET) in a new low cost 70-mil square metal-ceramic package has been announced by California Eastern Laboratories, exclusive ILS. distributor of Nippon Electric Co., Ltd.’s microwave semiconductors. The new device, called the NE 24483 has been widely expected as a broad based replacement for conventional bipolar transistors in 4-8 GHz small signal applications. The device, with a typical 4 GHz noise figure of 1.7 dB and a 13 dB associated gain is better than and competitively priced with the decades from a maximum multiplier setting. The 600-ohm output of the Model 1 19P delivers 20 V peak- to-peak open circuit, or 10 V into a 600-ohm load. An amplitude control provides a continuou sly- variable control of the output voltage, and a D.C. offset control makes available a maximum of ±10 V of output voltage offset. Pulse and ramp w aveforms can be reversed in polarity by means of an invert switch, A TTL- compatible pulse output capable of sinking 20 TTL loads is available at the front panel. Price £230.00 Dana Electronics Ltd ., Collingdon Street, Luton, Beds (43t IV! 1 1 ! " »■ ml erat ^ior nei P> ejects t< ou/fd. The new Map! in Catalogue is no ordinary catalogue... Catalogue includes a very wide range of components: hundreds of different capacitors; resistors; transistors; I.C.’s; diodes; wires and cables; discotheque equipment; organ components; musical effects units; microphones; turntables; cartridges; styli; test equipment; boxes and instrument cases; knobs, plugs and sockets; audio leads; switches; loudspeakers; books; tools — AND MANY MANY MORE. . — — ^tTiornPYOF OUR CATALOGUE r THIS COUPON FOR voun CO neyndWi 1 new wn*j*to** I Please rush nie ^opv , am comp | e telY receip t. \ bv return of P° at - , send 5 0 p within 1 days withir I worth every P ennV - return the catalogue to Y 1 if | am not satisfied - 1 ma V nt J e rstand that I need not i ° a,ato9ue should ' ch00ie 1 keep it. NAME address I Qur bi-monthly news letter keeps you up to date with latest guaranteed prices - our latest special offers [they save you I pounds] - details of new projects and new lines. Send 30p ■ for the next six issues C5p discount voucher with each copy]. SS 6 SLR I I _ flnx3, Rav'eigh. Essex, Maplin Electronic Supplies*^ " “ DON'T DELAY POST COUPON NOW IT COULD BE THE BEST 50p YOU'VE EVER SPENT. nrapm ELECTRONIC SUPPLIES P.O. BOX 3, RAYLEIGH, ESSEX SSS 8LR Telephone; Southend ( 07 02 ;• 715155 Call 1 at our shop 2B4 London fioad r Westcliff-on- Sea, Esse sc, { Closed all day Monday! Telephone; Southend 10702) 47379