m 8185 918 5 3i8IOI8i , 9131 18 8 5J88 89 1 8 1 118 9 918 5 818 9 188 88138 89 O 8 9 ii 8 5181318 9 ''4 | 21 ' 84 319 818 1 418 / 88 3 a 8 a 3 1 Il8l4f)3 -JJ / »1 / / / /iai / gi3 s 8013 <1 /■ 26p 21p 1.56 £2.80 £2. tO 49p 40p 32p 65p 55p 45p 57p 46p 36p 49p 40p 32p 67p 55p 45p .08 89p 72p 35p 28p 22p 34p 28p 23p 47p 39p 31p 78p 63p 53p 68p 58p 48p .75 £1 .48 86p .00 83p 67p .06 S8p 7 Ip .20 £2.50 £1.90 .35 £1.14 90p .35 £1.14 90p .64 £1.34 99p 555 (8 pin dip) V 55p • 555 CTO-99) T 8lp • 556 (14 pin dip) £1.29 • 703 (RF/IF Amp) 68p 709 (8 pin dip) 38p 709 (TO-99) 45p 709 (1 4 pin dip) 39p 710 (8 pin dip) 39p 710 (T0-9?) 45p 710 (14 pin dip) 44p 711 (TO-99) 51p 711 (14 pin dip) 44p 720 (A.M. Rodio) £1 .76 723 (TO-99) 723 (14 pin dip) 741 (8 pin dip) 741 (TO-99) 741 (1 4 pin dip) 747 (14 pin dip) 748 ( 8 pin dip) 748 (TO-99) 748 (14 pin dip) 753 (F.M. hr. £1.09 74p 36p 4* 36p £1.04 42p 49p 75491 75492 hr. I.F.) £1.08 88p £ 1.10 Regulolors 100 mA 78105WC (TO-92) 60p • 78L12WC (TO-92) 60p • 7811 5WC (TO-92) 60p • Regulators 100mA 78L0SAWC (T8A62SA)90p • 78L12AWC (TBA625B) 90p • 78L15AWC (TBA625Q90p a Regulators 500mA 78M05HC £1.35 a 78M12HC £1.35 • 7QM15HC £1.35 • 78M18HC £1.35 • 78M24HC £1.35 • Regulators IA 7805KC (T0-3) 781 2KC (TO- 3) 781 5KC (TO- 3) 781 8KC (TO- 3) 7824KC (To- 3) £2,09 • £2.09 • £2.09 • £2.09 • £2.09 • St£M£NS LCDs LIQUID CRYSTAL DISPLAY complete with socket and removable reflective booking; Ref AN4132R 13mm character height. Con be directly driven by Notional Semiconductors Alarm Clock chip MM5316. £13.99 a _/ / i • i~ n >— ' D ~ r L/NE-O-LIGHT 70^ Regulators IA 7805UC (T0-220) £1.72 • 7812UC (T0-220) £1.72 • 781 5UC (TO- 220) £1.72 • 781 8UC (TO- 220) £1.72 a 7824UC (T0-220) £1.72* PHOTO -DARLINGTON NEW LED Linear Cursors each device contains 10 light emWting diodes in a 20pin dual- in-line package. Ideal for solid state analogue meters or dials. Type 101 RED £2.26 • SPECIAL PURCHASE ICL6038 AY- 1 -021 2 AY-1 -5051 AY-5- 1224 AY-5-3500 AY-5-3507 AY-5-4007 \ 2N5777 1 Veto; Vcbo 25 v, Vebo 0v // Vceo; Vebo 25v; VEBO 8v " hfe 2500; Ic 250 mA 3Sp. enables £3.52 • £6.93 £1.44 £3.95 • £6.59 a £6.59 a £7.94 a Iftronlx Monsanto 28 36 40 8 • Price 13p I5p 15p 26p 30p 39p 44p 3lp 35p OJ25 ' dio. lens^^^v (TIL209) h 10* 100 * Red 16p I5p I3p • Green 27p 24p 22p • Orange 27p 24p 22p • Yellow 34p 31 p 29p • Low Cost Red GoAiP > s ^-4 Motorola MLED 500 In a T092 package. 15p L.E.D.s Free snap-on plastic retainer 0.16- d>o. tens 0 . 2 - dia. lens (MLED 650) 10- 100 ♦ 1* 10* 100 s 24p 22p • I8p * I6p I4p a 30p 27p « 30p 27p 25p • 30p ?7p • 30p 27p 25p • 32p 29p • 35p 33p 30p • NEW Opto-isolators IL1 (4N25 or T1L1I6) 6 pin industry standard pecLoge. 2. 5KV isolation £1 .00 • Litronix Double Digit Displays 0 .5"; Common Anode * 2 R Ai D.P.'s DL72I gives f 1 .9 DL727 gives 0.0. to 9.9 Suitable for Clocks; Instruments; T.V. Channel Indicator Out Price £4 .75 eoch . e NOTICE Postage & Pocking Charges With the recent incicose in postal charges end a continuing increase in packaging cost* we hove been forced ro review our policy. Henceforward: 1 . Orders valued at £5 or more will be post free. 2. All U.K. ‘small package* orders will go first class mail. 3. Minimum postage & packing charge will increase to 20p. 9.3 9.4 f 9.6 ‘ L INTAR I.C.'s BHA0002 £3.01 MCI 358 (CA3065) £1 .16 SN76S44N £1.81 MCI 375 £1.48 SN76550-2 (TAA55Q) 89p CA2111 £1.19 SN76552-2 81 p CA3045 £1.69 MCI 455 (S55T) 62p « SN76660N (TBA120) 75p CA3046 68p MC1456CG £1.68 SN76666N (CA3065) £^12 CA3053 S9p MCI 45 BC PI 84p CA3065 £1.60 MC1468G £2.18 • TAA263 £1.50 CA3075 £1.64 MCI 4951 £4.24 TAA300 £2.16 CA3078 £1.26 MC1496G 96p TAA31QA £1.87 CA3C80 59p TAA320 £1.44 CA3081 £1.86 MC3302P £1.50 TAA350 £2.43 CA3082 £1.86. MC3401P 74p TAA370 £3.45 CA3039E (T DAI 200) £2. 43 TAA550 75p CA3097E - £1.67 • MFC40008 87p TAA570 £2.74 CA3123E £1.76 MFC406QA 79p • TAA700 £5.03 CA3401E (LM3900) 68p MFC6030A 79p • CA3600E £1.44* MFC6040 96p TBA120S £1.25 MFC6070 £1.66 TBA231 £1.02 CT7001 £5.34 • TBA281 (723) £2.59 MM5314 £4.80 • TBA500Q £3.16 L005TI (TO. 3) £1.46 • MM 53 16 £9.99 • T8A520Q £3.85 L036TI (TO. 3) £1.46 • TBA530Q £3.27 L037TI (T0-3) £1.46 • MVR5V (T0-3) £1.45 • T8A540Q £3.72 LI 29 (SOT-32) 85p • MVR12V (TO-3) £1.45 • T8A550Q £5.29 L130 (SOT-32) 8Sp • MVR15V (TO-3) £1.45 • TBA560CQ £5.27 LI 31 (SOT-32) 85p • NE540L £1.25 TBA625A £1.03 LM301 T (TC-99) 65p NE546A £1.16 TBA625B £1.03 < LM301 S (6. pin dip) 59p NE555V 73p • TBA625C £1.03 LM331A T £1.03 £ 1,12 69p £1.84 80p £2.39 £2.42 £4.13 £4.62 £ 1.12 83p £1.52 64p 88p 68p £1.52 SL415A SL437D SI 440 SL610C SL61IC S1612C S1613C SL620C SL621C SL622C SL623C SL624C SL630C SL640C S1641C SL645C SL650C SN75491N SN75492N £2.75 £7.50 £2.84 • £2.03 £2.03 £2.03 £4.31 £3.06 £3.06 £7.62 £5.57 £2.84 £1.87 £3.75 £3.75 £3.75 £9.85 88p • £ 1.10 • TCA270Q TCA760 TCA800Q TCA830S TCA940 TOA1054 T DA 1200 T DA 1 405 TDA1412 TDA14I5 TDA2010 TDA2020 ULN2111A ZN4I4 £5.24 £2.16 £7.24 £1.04 £2.25 £1.50 £2.43 BQp • 90p • 90p • £3.00 £3.75 £1.52 £1.26 SN76001N (TAA61 1) £1 .82 SN76003N £3.30 SN76013N £1.98 SN76023N £1.98 SN76227N (MC1327) £1.09 SN76532N £1.08 MIN I TR ON Minitron Filament Display bi-directional 0.36" I 3015F 0-9 l/H d.pt. 30I5G-1 £1 .08# SEVEN SEGMENT D IS PL ASS COMMON ANODE R/H Dec. Pt. COMMON ANODE L/H Dec. Pt. COMMON ANODE - 1 COMMON CATHODE R/H Doc. Pt. RED DL707R DL707 DL701 DL704 GREEN RED YELLOW M-‘*N5) MAN7I MANSI MAN52 MAN72 MAN82 MAN53 MAN73 MAN83 MAN54 MAN74 MAN84 ORANGE MAN36I0 GREEN XAN51 RED XAN7I YELLOW XAN8I MAN 3620 XAN52 XAN72 XAN82 MAN 3630 MAN3640 XAN54 XAN74 XAN84 £1.82* £1.82* £1.82* £1.82* £1 .82# £1.49* £1.49* £1.49* GREEN MAN45I0 RED MAN47I0 YELLOW MAN4810 ORANGE MAN46I0 MAN 4520 MAN4720 MAN 4820 MAN4620 MAN 4530 MAN4730 MAN 4330 MAN 4630 MAN 45 40 MAN 4740 MAN4840 MAN 46 40 12.32* £2.32* £2. 32a £2. 32a C.A. l/H Dec. Pt. < - Ur | C.C. l/H Dec Pt. RED DL747 DL746 DL750 DL749 12.42a NOTE: MAN4000 series pinouts are 14 pin dil the same os MAN50;70 & 00 series. VAT INCLUDED I ferns marked with a • Include 8% VAT Itoms unmarked Include VAT at 25% ADVERT. No.1. of Scries B. CALI.ERS WELCOME ■ W • * ■ • • I w I I W elekvor december 1975 - 1201 loin the Digital Revolution reach yourself the atest techniques of Jigital electronics omputers and calculators are only the beginning of the igital revolution in electronics. Telephones, wristwatches, automobile instrumentation — these will be just )me of the application areas in the next few years. re you prepared to cope with these developments? ns four volume course — each volume measuring 1 f " x 8^" and containing 48 pages — guides you ep-by-step with hundreds of diagrams and questions irough number systems. Boolean algebra, truth tables, 3 Morgan's theorem, flipflops, registers, counters and iders. All from first principles The only initial ability turned is simple arithmetic. t the end of the course you will have broadened your Drizons, career prospects and your fundamental under- anding of the changing world around you. Design of Digital Systems Book! AnPvmtoc £5.95 Also available - a more advanced course in 6 volumes: 1. Computer Arithmetic 2. Boolean Logic 3. Arithmetic Circuits 4. Memories & Counters 5. Calculator Design 6. Computer Architecture Offer Order this together with Digital Computer Logic & Electronics for the bargain price of £ 9.25, plus 50 p p & p. plus 50 p p & p. Design of Digital Systems contains over twice as much information in each volume as the simpler course Digital Computer Logic and Electronics. All the information in the simpler course is covered as part of the first volumes of Design of Digital Systems which, as you can see from its contents also covers many more advanced topics esigner anager nthusiast :ientist ngineer tudent These courses were written so that you could teach yourself the theory and application of digital logic. Learning by self-instruction has the advantages of being quicker and more thorough than classroom learning. You work at your own speed and must respond by answering questions on each new piece of information before proceeding to the next. uarantee - no risk to you you are not entirely satisfied with Digital Dmputer Logic and Electronics or Design of Digital /stems, you may return them to us and your oney will be refunded in full, no questions asked. Digital Computer Logic and Electronics A Self -instructional Course C PG.ine MA (Cantab) A W Unwm BA (Cantab) Book Book Book Book 1 3 4 Basic computer logic Logical circuit elements Designing circuits to carry out logical functions Flipflops and registers Digital Computer Logic and Electronics A S*» tni.uitcrut Cant Book *1 li> ~ ■ *•* £3.95 plus 50 p packing and surface post anywhere in the world Quantity discounts available on request. Payment may be made in foreign currencies. VAT zero rated. To: Cambridge Learning Enterprises, FREEPOST, St. Ives, Huntingdon, Cambs PE17 4BR Please send me set(s) of Digital Computer Logic & Electronics at £ 4.45 each, p & p included. * or set(s) of Design of Digital Systems at £ 6.45 each, p & p included. or combined set(s) at £ 9.75 each, p & p included. Name Address L * delete as applicable No need to use a stamp — just print FREEPOST on the envelope. J 1202 — elektor decern ber 1975 publisher's notic Many Elektor circuits are accompanied by designs for printed circuits. For those who do not feel inclined to etch their own printed circuit boards, a number of these designs are also available as ready-etched and predrilled boards. These boards can be ordered from our Canterbury office. Payment, including £ 0.15 p & p, must be in advance. Delivery time is approximately three weeks. Bank account number: A/C No. 1 1014587, sorting code 40-16-1 1 Midland Bank Ltd, Canterbury. circuit number issue price % VAT edwin amplifier 97-536 6 1.20 (25) austereo 3-watt amplifier HB1 1 5 1.10 (25) austereo power supply HB1 2 5 0.55 (25) austereo control amplifier HB1 3 5 1.50 (25) austereo disc preamp HB14 5 0.65 (25) universal frequency reference HD4 5 1.10 ( 8) distortion meter 1437 1 1.65 ( 8) a/d converter 1443 3 0.90 ( 8) tap sensor 1457 1 0.60 ( 8) minidrum gyrator 1465A 2 0.80 (25) minidrum mixer/preamp 1465B 2 0.55 (25) minidrum noise 1465C 2 1.05 (25) miniature amplifier 1486 6 0.55 (25) light dimmer 1487 6 0.45 ( 8) beetle 1492 4 2.20 ( 8) equa amplifier 1499 1 1.20 (25) electronic loudspeaker 1527 2 0.50 (25) mostap 1540 2 1.05 ( 8) car power supply 1563 4 1.25 ( 8) digital rev counter (control p.c.b. only!) 1590 1 0.55 ( 8) car anti-theft alarm 1592 4 1.40 ( 8) mos clock 5314 clock circuit 1607A 1 1.15 ( 8) mos clock 5314 display board 1607B 1 0.85 ( 8) mos clock timebase 1620 4 0.70 ( 8) minidrum tap 1621 A 2 0.70 (25) minidrum ruffle circuit 1621 B 3 1.10 (25) automatic bassdrum 1 621 C 3 0.55 (25) tap preamp front panels: power 1626A 7 1.55 (25) input 1626B 4 1.55 (25) volume 1626C 4 1.55 (25) tone 1626D 4 1.55 (25) width 1626E 4 1.55 (25) microdrum 1661 2 0.95 (25) aerial amplifier 1668 1 0.95 (25) coilless receiver for MW and LW 3166 5 0.80 (25) tap preamp 4003 4 1.80 (25) clamant clock, alarm 4015-13 7 1.30 ( 8) clamant clock, time signal 401 5-1 6 7 0.85 ( 8) twin minitron display 4029-1 2 1.40 ( 8) twin led display 4029-2 2 1.40 ( 8) twin decade counter 4029-3 2 1.40 ( 8) recip-riaa 4039 2 0.50 ( 8) disc preamp 761 31 4040A 3 0.95 (25) maxi display 4409 2 1.50 ( 8) versatile digital clock 441 4B 6 1.10 ( 8) dil-led probe 5027A+B 2 1.85 ( 8) big ben 95 5028 2 1.25 (25) compressor 601 9A 3 1.20 (25) tv sound 6025 2 1.40 (25) ota pll 6029 7 1.10 (25) car clock (2 boards) 7036 6 1.75 ( 8) car clock front panel (transparent red plastic) 7036-3 6 0.90 ( 8) tv tennis, main pcb 9029-1 A* 7 3.80 ( 8) tv-tennis, modulator/oscillator 9029-2* 7 0.90 ( 8) frequency counter 9033* 7 1 .30 ( 8) tap power 9072* 7 1.90 (25) tup/tun tester 9076* 4 1.70 ( 8) tup/tun tester front panel 9076/2A 4 1.90 ( 8) p.c.b. and wiring tester 9106* 5 0.55 ( 8) rhythm generator M 252 9110* 5 0.80 (25) 7400 siren 9119* 5 0.75 (25) CA3090AQ stereo decoder 9126* 5 0.80 (25) kitchen timer 9147* 5 0.75 ( 8) capacitance meter 9183* 5 0.75 ( 8) tv tennis 5-volt supply 921 8A* 7 0.80 ( 8) NEW: circuit number issue price % VAT clamant clock, striking system 401 5-27 8 1.20 ( 8) frequency counter preamp 9031-1* 8 1.15 ( 8) frequency counter, —5 V supply 9031-2* 8 0.75 ( 8) 730/740 (1C control amplifier) 9191* 8 1.10 (25) die 9169* 8 0.70 ( 8) * with solder mask All prices include VAT at the rate shown in brackets. ekeHTor Volume 1 — number 8 Editor Deputy editor Technical editors W. van der Horst P. Holmes J. Barendrecht G.H.K. Dam Art editor Drawing office Subscriptions E. Krempelsauer Fr. Scheel K. S.M. Walraven C. Sinke L. Martin Mrs. A. van Meyel UK. Staff: Editorial : T. Emmens Advertising : P. Appleyard Editorial offices, administration and advertising: 6 Stour Street, Canterbury CT1 2XZ. Tel. Canterbury (0227) — 54430. Telex: 965504. Elektor has been published every two months until August 1975; it now appears monthly. Copies can be ordered from our Canterbury office. The subscription rate for 1975 is £ 3.60 (incl. p & p); the first issue (Nov/Dec 1974) will be included in this at no additional cost. Single copies: £ 0.35 (incl. p & p: £ 0.45). Subscription rates (airmail): Australia/New Zealand European countries outside UK USA All other countries 8 issues £ 7.20 8 issues £ 5.20 8 issues £ 6.25 8 issues £ 6.40 Subscribers are requested to notify a change of address four weeks in advance and to return envelope bearing previous address. Members of the technical staff will be available to answer technical queries (relating to articles published in Elektor) by telephone on Mondays from 14.00 to 16.30. Letters should be addressed to the department concerned: TQ * Technical Queries; ADV = Advertise- ments; SUB = Subscriptions; ADM = Administration; ED = Editorial (articles submitted for publication etc.); EPS = Elektor printed circuit board service. The circuits published are for domestic use only. The submission of designs or articles to Elektor implies permission to the publishers to alter and translate the text and design, and to use the contents in other Elektor publications and activities. The publishers cannot guarantee to return any material submitted to them. All drawings, photographs, printed circuit boards and articles published in Elektor are copyright and may not be reproduced or imitated in whole or part without prior written permission of the publishers. Distribution: Spotlight Magazine Distributors Ltd., Spotlight House, 1, Bentwell road, Holloway, London N7 7AX. Copyright © 1975 Elektor publishers Ltd — Canterbury. Printed in the Netherlands. selektor 1205 calendar — W.G. Paans 1210 Many mechanical and electromechanical clocks and watches are now provided with date indication. Addition o a calendar to an electronic digital clock is a fairly simple matter, and the circuit given here gives the month as well as the date. photofinish — F. Ansoms 1214 automatic barrier control for model railway level-crossings — R. ter Mijtelen 1215 The circuit described here provides control of the 'automatic barrier' type of level crossing (or indeed of the old-fashioned gate type). It gives a realistic simulation of the visual and audible warnings. There is no limit on the train length, and the system will operate when trains are passing in both directions. turning off thyristors — W. Back 1217 When thyristors are carrying A.C., they will turn off at every zero-crossing — which can be a nuisance. When carrying D.C., however, they won't turn off at all — which is worse. This article takes a basic look at how to cope with the latter problem. three-tracer — R. Si ntic 1218 quadi-complimentary 1220 In the new Quad 100 Watt amplifier design, a special negative feedback arrangement enables the output tran- sistors to be zero biased without creating non-linearity problems. elektor services to readers 1 222 fuse indicator — J.W. van Beek 1222 chestnut oven 1223 cd-4 — Victor Company of Japan, Limited 1224 Over five years have elapsed since the CD-4 system was announced as a means of achieving quadraphony from a disc record. However, there seem to be various misunderstandings of the CD-4 system. For this reason, in this paper, we would like to restate our policy on CD-4 and describe the present state of CD-4 technology to give readers a fuller understanding of the CD-4 system. cd4-392 — Victor Company of Japan, Limited 1229 As most quadro-enthusiasts will know, an integrated demodulator for CD-4 has been available in the retail trade for several months now: the CD4-392. In this article, we are pleased to present all relevant information concerning pinning and specifications. A practical circuit using this 1C is included. tut — M. Keul 1234 electronic candle — P. Engelmann 1234 preamp for frequency counter 1235 In the last issue of Elektor the basic circuit of a frequency counter was described. In this issue a preamplifier to increase the input sensivity is discussed. 730/740 1240 Using only two ICs, the TCA730 and TCA740, a complete stereo control amplifier can be built. An excep- tional feature is that the volume, balance, and tone are all DC controlled. die — M.G. Fishel 1244 doorchime driver — T. Mey rick 1245 contents volume 1 1247 clamant clock (2) 1249 In the last issue of Elektor various sound effects which could be added to electronic digital clocks were de- scribed, including a 'tick', alarm systems and a time signal simulator. In the second part of the article various chiming and striking systems are discussed. digital ic's 1253 tup-tun-dug-dus 1255 market 1256 1204 — elektor december 1975 advertise mem Experiments with operational amplifiers by G.B. Clayton This book covers a wide range of practi- cal operational amplifier applications. It provides circuits which include compo- nent values, and suggest measurements that can be made in order to study circuit action. The experiments will be useful for a large variety of measurement andiinstru mentation systems. The way in which performance errors are related to the characteristics of the particular amp- Iffier used in a circuit are treated in an appendix. OOK CORNER Linear integrated circuit applications by G.B. Clayton This book is concerned with the newer circuits now available, important for such things as signal measurement and processing systems. After first looking at how to use oper- ational amplifiers as measurement amplifiers and in active filter circuits, the book then deals with the more recently introduced I inear integrated circuits, monolithic integrated circuit modulators, four quadrant multipliers, timers, waveform generators and phrase locked loops. Paperback £3.30 Hard cover £6.86 Paperback £3.30 Hard cover £6.85 ALLTHESE POPULAR TITLES ARE ALSO AVAILABLE □ LINEAR MICROELECTRONIC SYSTEMS £8.30 hardcover 234 x 156 mm 272pp DELE CJRON'C SYSTEMS FOR RAD!°, A. G. Martin and F. W. Stephenson £4.30 paperback TAEfX?,,-* ^°i EL ^ t ' TR ° N CS , . . . MECHANICS Rhys Lewis compre ensive course - th r ^ rt i ril |oX]ESSENTI AL FORMULAE FOR ELECTRICAL This book covers, in non-mathematical terms. application of amplifier systems with particular reference to those devices readily available in integrated circuit form. £3.75 234 x 1 56 mm 242 pp □ LINEAR ELECTRONIC CIRCUITS AND SYSTEMS G. D. Bishop Macmillan Basis Books in Electronic series (series editor: Noel M. Morris) This text, mainly for Technician Parts II and III, ONC/D, HNC and university students, covers linear electronic circuits from basic a.c. theory to modern integrated-circuit config- urations, with the minimum of mathematics. It takes the 741 1C as a typical example, and illustrates the use of the operational amplifier in many different applications. £2.45 234 x 1 56 mm 147 pp AND ELECTRONIC ENGINEERS Noel M. Morris A handy reference book, A4 size for easy use, containing all the formulae necessary to the student and engineer in a wide variety of dis- ciplines. It is up-to-date, includes a section on SI units as well as resistor colour codes and preferred values. £1.10 32pp □ELECTRONIC EQUIPMENT RELIABILITY J. C. Cluley The principles of assessing the reliability of electronic equipment, including the mathem- the fundamental theory and application of electronic systems, and includes work on fault- finding and system reliability. It meets the requirements of the 'systems' side of the CGLI Course 222, Radio, TV and Electronics Mechanics Part I and II, and gives objective tests similar to those met in the examination. £2.60 234 x 156 mm 240 pp □ AN INTRODUCTION TO ELECTRICAL INSTRUMENTATION B. A. Gregory A comprehensive introduction to the growing number of increasingly sophisticated instru- ments available for use by engineers and tech- nicians. It can be used both as a student text atical background, and methods to improve . , , , _ reliability are described and explained at a level Inc * i u< ^ es suitable for degree and diploma students. IN □ FIELD-EFFECT TRANSISTORS INTEGRATED CIRCUITS J. T. Wallmark and L. G. Carlstedt A concise, fully-illustrated text on the use of FETs in the construction of ICs. The interference, calibration, accuracy and a discussion on the selection criteria advisable when matching instruments to requirements. £4.20 234 x 156 mm 341 pp □ ELECTRONICS MEASUREMENTS Ten contributions from industry showing the Edited bv W F Waller student how to test the working parameters of A practic V a| survey by specja|ist authprs frpm £2.65 216 x 138 mm 192 pp □ ELECTRONIC COMPONENT TESTING Edited by W. F. Waller emphasis lies mainly on their design, manufact- electronic components. Contents include semi- industry pf the methods an d instrumentation ure and use. £4.75 234 x 156 mm 160 pp □ DIGITAL ELECTRONIC CIRCUITS AND SYSTEMS Noel M. Morris Macmillan Basis Books in Electronics series Using the example of the electronic calculator to illustrate many of the systems, this text begins with a description of basic logic functions, including a full coverage of boolean algebra and Karnaugh maps. It then goes on to describe high-speed switching elements, logic conductors and integrated circuits. £1.10 A4 91pp illustrated □transistor audio emplifiers S. J. Hellings The subjects in this book have been specially necessary to obtain useful measurements of electrical quantities. £1.10 A4 82 pp □ electrical installations and REGULATIONS Michael Neidle chosen to be of maximum benefit to designers, Primarily for the CGLI Course 231 Installations users and radio amateurs. The book shows how this large-format textbook covers the special circuits can be designed to suit individ- practical installations part of the course, en- ual requirements using readily available com ponents. £5.60 224 x 150 mm 337 pp gates, calculating functions, asynchronous and □jraNsistORS ,n PULSE CIRCUITS synchronous counters, shift registers and dis- play decoding circuits. £2.45 234 x 1 56 mm 143 pp □electrical circuits and systems Noel M. Morris Macmillan Basis Books in Electronics series (series editor' Noel M. Morris) This, the author's latest in a long line of elec- tronics textbooks, covers the CGLI Parts II and III Technician syllabuses, as well as being couraging safety consciousness, and giving a clear explanation of the applications of the relevant IEE Regulations. £2.45 250 x 225 mm 90 pp □ SEMICONDUCTOR ELECTRONICS BY WORKED EXAMPLE F. Brogan gives a clear picture of the switching transistor Intended for students taking Technician, in its various conditions.The text deals with the Telecommunications and HNC/D courses, this G. Fontaine Intended primarily for students, this book principles of solid state electronic devices as well as with particular applications to pulse circuits — increasingly important in computer and telecommunications systems. The book is lavishly illustrated with colour drawings of numerous circuits. £6.40 216 x-140mm 448 pp large-format, clearly laid-out text teaches the theory and practice of solid-state electronics by answering questions from past examination papers. All circuits, devices and systems discus- sed are related to today's technology, designs and developments. £2.70 254 x 229 mm 136 pp suitable for ONC/D, HNC and HTC. All available from Technical Book Services Ltd., Dept. E8, 25 Court Close, Bray, Maidenhead, Berks. SL6 2DL ORDER FORM Please supply the following (all prices include postage & Packing): copy/copies of Experiments with Operational Amplifiers by G. B. Clayton at £6.85 per copy (hardcover) copy/copies of Experiments with Operational Amplifiers by G. B. Clayton at £3.30 per copy (paperback) copy/copies of Linear Integrated Circuit Applications by G. B. Clayton at £6.85 per copy (hardcover) copy/copies of Linear Integrated Circuit Applications by G. B. Clayton at £3.30 per copy (paperback) Plus any books from the list aboye. (tick appropriate boxes). To: Technical Book Services Ltd. Dept. E8 25 Court Close Bray Maidenhead Berks. SL6 2DL. I enclose £ Name Address Sapphire arc tube increases fficiency of new HPSV lamps i single-crystal sapphire material manu- ictured by Corning for arc tubes in igh pressure sodium vapor lamps in- reases luminous efficacy of such lamps y as much as 5 to 10 percent, he improved efficiency of the new 'orstar sapphire arc tube over conven- lonal polycrystalline arc tubes is a esult of the mono-crystalline material’s xtreme purity and increased trans- iency. Tius, high-pressure sodium vapor lamps aade with Corstar arc tubes can be esigned to produce more lumens per /att from the same energy input, or to iroduce the same amount of light with jss energy. haracterized by the golden color of the ght they produce, high-pressure sodium apor lamps are coming into increased se for street lighting. Coming’s major ustomer for Corstar arc tubes is urrently involved in supplying the imps for the extensive street relighting rogram in New York City, he arc tube is the key component in a igh-pressure sodium vapor lamp ecause it must have the dimensional Lability , corrosion resistance and tem- erature capability to contain a high- emperature discharge of metallic Ddium. he Corstar arc tubes will operate at emperatures in excess of 1200 degrees while maintaining superior resistance o the corrosive attack of high-tempera- ure sodium under pressure. Corstar arc ubes have a rated life of over 20,000 ours. 5BC engineers approve Thermax' apes i small but valuable contribution to the fficiency of the BBC Television service ^ being made by the Somerset firm, 'hermographics Measurements Ltd of lchester. This Company’s temperature- ensitive tapes are used to take tempera- ure readings in the co-axial feeder cables which connect the transmitters to he aerials in the UHF 625-line system throughout the country. It is not possible to take satisfactory measurements by inserting ther- mometers inside the co-axial feeders and, in fact, their presence could upset the operation of the system. What the BBC engineers like particularly about these ’Thermax’ tapes is that they are irreversible and, unlike special paints sometimes used for this purpose in the past, they are not subject to change as the exposure time is extended. ‘You can go back to a site an hour, a week or a year later and the information you want is there.’ These tapes are also used by BBC staff to make temperature checks on transis- tors and on other small components and when new types of equipment are introduced. Available in five temperature ranges be- tween 37°C and 260°C, ‘Thermax’ tapes indicate temperature changes by a simple colour change from silver-white to black, which is easily observed against the temperature scale marked in red. Through a technical innovation in printing on the tape, a clearer print of the temperature scale has been achieved with the further benefit that it cannot be dissolved or erased. Thermographies Measurements Ltd., The Square , Ilchester , Yeovil , Somerset Interplex single-tube colour television camera system A new single-tube colour camera —Interplex — developed by Siemens, incorporates a tube which gives uni- formly high colour rendition of high resolution. The unit consists of a com- pact camera with a tube and a decoder which converts the colour information into standard PAL television signals. The Interplex single-tube colour camera uses a new type of dichroic strip filter. In contrast to a normal three-tube camera, the colour distributor used to break down the image arriving from the lens into red, green and blue channels is integrated in the Interplex picture tube. This has made it possible to reduce the size of the camera considerably by dis- pensing with the accessories for colour coincidence which is so difficult to attain with a three-tube system. The signal information supplied by the television camera tube in the 4.43 MHz range is converted into standard PAL signals in a decoder with comb filter systems and electronic circuitry. Each frequency spectrum of the black-and- white and colour information is sep- arated by the comb filter and here the spectral lines of the video signals are broken down into colour (chrominance) and luminance information. Additional electronic circuits suppress interference from repetitive luminance in the chrominance channel (cross-colour suppression), and are also used to suppress interference in the opposite direction (cross luminance suppression). The individual colour signals are pro- cessed without loss of information or colour rendition and uniformity, and can be passed on to a receiver as a PAL coded colour signal. The decoder can also be used for horizontal and vertical aperture correction and addition/sub- traction of blue, green, red and white colour components (matrixing). A standardized connection has been established within the Interplex system for the coded colour signal supplied by the tube of the camera (multiplex signal). The single-tube colour camera can be fitted with an antimony trisul- phite coated tube (Vidicon with integral filter type XQ 1360) or a silicon tube (Interplex-vidicon type XQ 1365). The silicon tube developed by Siemens is very sensitive to light and has a low intertia and a linear characteristic. A resolution of approx. 6 MHz can be achieved in red-green-blue-operation. Complete resolution is possible in PAL operation. The multiplex signal produced by the camera is compatible and can also be shown on a black-and-white set. Several cameras can be operated in sequence using the standardized connection, the decoder and a selector switch. The multiplex signal can be recorded directly on polychromatic video re- corders so that the decoder is only required for reproduction. Using the new camera it is also possible to set up colour television units that are no larger than black-and-white television units. DIGITAL CLOCK £9.56 MATCHED CHIP & DISPLAY Inc. VAT. Post & Packing FUTABA 5-LT-01. 7 SEGMENT Phosphor Diode. 12.5mm Digit AM/PM and colon CALTEX CT7001. MOS LSI 28/30/31 Day Calendar 24-hour Clock Snooze Alarm Clock Radio Feature Easily Settable Counters DISPLAY Only Price £6.36 Inc. VAT CHIP Only Price £5.50 Inc. VAT IMTECH PRODUCTS LTD. IMP HOUSE, ASHFORD ROAD, ASHFORD, MIDDX. Telephone: Ashford 44211 Telex: 936291 crystals Fast delivery of prototypes and production runs INCLUDING: Statek LF crystals in T05 package Buckman LF, clock and mobile radio crystals Astro Filter crystals Jan General purpose crystals Crystals for all Elektor designs Interface Quartz Devices Limited, 29 Market Street, Crewkerne, Somerset. Tel.: (046031) 2578 Telex: 46283 The Castle 8.RS.DIX A highly sensitive, full range eight inch unit designed for use in the recommended cabinet, or one of similar dimension. Suitable for use with good quality stereo installations, tape recorders, car radios, public address and background music systems, it has a frequency range of 50 to 20,000kHz - the lower limit variable with increases in cabinet volume. Recommended retail price is £9.00 excluding VAT. Aluminium Voice Coil High Flux 14.000 Oersteds Ceramic Magnet Roll Surround Double Diaphragm 8" Die -cast Chassis 8 ohms Impedance 15 Watt DIN Power Handling Acoustics Limited Park Mill, Shortbank Road, Skipton, Yorks. Tel: Skipton 5333. JICMUI leading between the lines elevision viewers throughout Britain re to be offered a remarkable new irvice - stop-press news flashed on to leir screens at the flick of a switch. A nail adaptor on a standard existing set ill offer the choice of up-to-the-minute ews ranging from, say, what won the .30 race to the current price of gold, lready launched by the BBC (British roadcasting Corporation), the Inde- endent Broadcasting Authority (IBA) r ill join in next year, he BBC’s new device, CEEFAX (See acts) has been operating since autumn 974 from the Television Centre, ondon, under editor Colin McIntyre. It as been providing viewers with a video irvice of printed material of up to 00 pages, each page consisting of over 0 lines of electronic type. idex he viewer can punch up an Index page n his screen and then choose which ind of information he wants to have — ews headlines, horse-racing results, oorts news, share or commodity prices, r eather, road condition report, airport formation or the timetable of the ay’s radio and television programmes, e makes his own choice of what he r ants and when he wants it — all the lformation is stored in CEEFAX’s lemory and kept up to date by the ditor and his staff in London. When le viewer punches up the page number squired it will appear within 1 5 sec- nds — less time than it takes to make a dephone connection, he IBA has developed a similar device diich it has named ORACLE (Optional .eception of Announcements by Coded ,ine Electronics) and this will begin a Lval service for viewers soon. Both aim t giving the viewer new and instant ccess to useful and necessary facts rom the price of sugar to the starting une of a play or film. The new services re national but could in time provide Deal neighbourhood information as Yell. Hanking Interval he CEEFAX and ORACLE devices xploit a potential in the television creen which has been known about for early 40 years. There is a section of the ne system which is unused by the lormal television transmission, a group •f 8 lines within the 625 line screen Yhich engineers call the ‘field blanking nterval’. Jy using two of them it is possible to end out a signal which remains invisible m all receivers except those fitted with 1 special decoding device and with the iddition of a small contraption a normal eceiver can be adapted to decode up to 00 pages of printed material sent out through the ordinary television trans- mitters. To provide this material costs the BBC very little. A small staff - only half a dozen — will shortly be assembled and organised into a mini-newsroom, where the various information services will be compiled, ‘subbed’, typed out at fre- quent intervals and stored in a small electronic ‘memory’ from which the viewer will be able to choose whichever sections he wishes. The capital equip- ment required at the end of the oper- ation should not cost more than £ 50,000 in all — about the cost of equipping a very small local radio station. When manufacturers decide to mass produce the equipment, the viewer should be able to adapt his receiver for an extra £ 200-£ 300. A full-size colour receiver with a built-in CEEFAX and ORACLE device will cost £ 600-£ 800 and a black-and-white receiver under £ 200. Prototypes However, for the moment, the only CEEFAX devices available are specially made jobs which cost about £ 800. The total number of subscribers at present is still well under two figures, a select consumer privilege enjoyed by those involved in the current experiment. But even now there is nothing to prevent anyone from building his own decoder, if he is familiar with basic television engineering. Some time in early 1975 the I BA’s ORACLE system should also begin transmitting and ITN (Independent Television News) is actively thinking up an elaborate set of services which it could provide on behalf of the two London programme companies (Thames Television and London Weekend Tele- vision) which will inaugurate the service. ITN sees ORACLE’S services as a suit- able supplement to its present output and could give ITV viewers a feed of the day’s sporting results as well as head- lines, newsflashes, weather reports and information about cinemas, theatres, concerts and television programmes. Sports And City News Racing results and share prices are ob- viously the type of detailed information constantly changing throughout an average day for which the new devices are ideally suited. The racing results on a busy day would require over 1 2 ‘pages’ of transmitted material to bring, say, 300 results. A list of the runners on an average day ( 1 60-200) would need an- other four or five pages. To provide the full range of services which ITV envisages will require in all 50-1 00 pages and that means exploiting almost the whole of the ‘memory’ of the present CEEFAX/ORACLE sys- tems. One area in which the new devices could prove very useful is the City of London. At present London stockbrokers use the four tape services of Extel, Reuters, AP- Dow Jones and the ‘Financial Times’. Reuters supplies a special video service of financial news with 22 channels of information. CEEFAX and ORACLE could provide a similar service very cheaply and conveniently on a single screen instead of the ten which some offices must now have. Maps And Diagrams The actual mode of transmission is a digital one and it is possible to sub- 1208 — elektor december 1975 advertisemei Coming soon in elektor: feedback PLL FM receiver TV-sound front-end digital master oscillator wrist watch MW receiver function generator dynamic noise limiter H.M. ELECTRONICS 275a, Fulwood Road, Broomhill, SHEFFIELD S10 3BD Tel: 0742-669676 BEC CABINETS (Book End Chassis) Standard cabinet GB1 14"x6"x2" GB1 A9''x6"x2'' GB2 14''x7"x3" GB3 14"x9"x4" GB4 14''x9"x6'' Send 1 5p for wallet of leaflets ( Refundable on 1 st purchase) A beautifully designed modern cabinet with simulated black leatherette top (PVC bonded to metal) solid wooden end cheeks, with room at the back for Output Sockets etc. Felt pads are fitted on bottom of cheeks for non-scratch. HARDWARE A comprehensive range of screws, nuts, washers etc. in small quan- tities, and many useful constructors' items. Sheet aluminium to individual requirements, punched, drilled, etc. Fascia panels, dials, nameplates in etched aluminium. Printed circuit boards to personal designs, one-off's or small runs. Machine engraving in metals and plastics, contour milling. Send lOp stamps for catalogue. RAMAR Constructor Services Masons road Stratford on Avon Warwicks. CV37 9NF. elektor back issues are still available Edition c,' / No. 5 is a summer circuit issue. It contains over one hundred cir- cuit designs all proved and mostly original. Price 80p (U.K. including postage). Prices for single copies (1, 2, 3, 4 and 6, including P & P*) U.K. Europe U.S.A. Australia/New Zealand 45p 65p 78p 90p * may be subject to increase in postal rates alektor elektor december 1975 — 1209 1 m ■%■■■■ ■■■■■ .W titute small blocks and shapes for the 3tters of the alphabet and the numerals, ’olin McIntyre and his staff can there- ore construct maps and diagrams to ap- pear on the screen, in various colours, as veil as ordinary writing in upper and Dwer case. With the addition of a com- uter, which is shortly to arrive, the usiness of map-display and diagram resentation will be made much easier nd extremely rapid. 'he computer’s memory will also ex- end the capacity of CEEFAX to make t possible for a 1000-page ‘book’, as it /ere, to be presented on the screen at a ingle viewer selection, each page of the ►ook replacing the previous one at the ate of one per minute. Vnother major use will be to provide )ermanent sub-titles for the deaf. 2EEFAX print can be made to appear t the bottom of the normal television >icture. 'he BBC has worked out that it would ake one man about 40 hours to type >ut and store the dialogue of a full- 3ngth play arranged to appear on the creen at the appropriate moments. By witching to CEEFAX deaf people ould receive the printed ‘soundtrack’ >f plays and documentaries without poiling the programme for ordinary iewers whose picture would be quite lormal. Similarly, viewers can have ‘newsflashes’ uperimposed on their evening’s ordi- lary viewing when major news occurs vithout interfering with the viewing of >thers who prefer to watch a whole mlletin at a stated time. Jew Intersil 24-hour alpha- mmeric readout CMOS watch ntersil recently introduced two ad- litions to its line of CMOS watch and :lock circuits and dropped the prices on wo more. fhe ICM7203 is a single-chip LED digi- al wristwatch circuit with alphanumeric capability, providing hours, minutes, lay, date and seconds readout. It is a 24-hour version of Intersil’s ICM7200 12 nr circuit and is available immedi- itely. rhe ICM7204 is a numeric only version )f the 7203. It interfaces with existing ^-segment LED displays. It is a 24-hour variation of Intersil’s ICM7202. Delivery s also immediate. dicing for the 7203 has been estab- ished at $ 16.70 at the 100-999 quan- ity level. The 7204 is priced at $ 10.80 it the same level. In a related move, Intersil also dropped the prices of its ICM7200 and ICM7202 to comparable levels, 3 16.70 for the 7200 down from $ 29.20, and $ 10.80 for the 7202 down from i 22.00. According to Intersil, no other presently available 24-hour LED watch circuit contains both digit and segment drivers on-chip. Other devices use external tran- sistors for drivers. Intersil LED watch units are totally integrated on one chip, including segment and digit output buffer circuits. This allows OEM watch and clock manufacturers to produce very compact movements without use of multiple-chip assembly techniques. Both circuits are supplied in 24-pin cer- amic leadless packages, 0.335 inches square, designed to be easily soldered onto PC boards. According to Intersil, watches made from these circuits are simple to oper- ate, the only LED timepieces offering the utility of a full calendar feature simi- lar to that used in conventional watches. One button - called the ‘command’ button — when pushed once displays the time, twice the day-date, and three times the seconds. (At second push, the ICM7204 displays date only, while the ICM7203 displays both day and date.) The two new circuits also greatly sim- plify watch setting. A second button, which can be recessed in the case to pre- vent accidental activation, cycles the setting modes: one push for date set; two for hour set, three for day set; four for minutes; and five pushes for seconds. In each set mode, the command button advances the watch display. Hours, for instance, advance by one each time the command button is pressed. Seconds are reset to zero by the command button. All set modes are independent of each other, allowing hours to be advanced past midnight without affecting day or date — important when travelling be- tween time zones. Both circuits operate at 32.768 kHz. Two parts, a 32.768 kHz quartz crystal and one trimming capacitor, complete the oscillator circuit. Current required is 4 microamperes, and the oscillator is de- scribed by Intersil as ultra-stable. The circuits have provisions for light sensors which increase the brightness of the display at high ambient light levels. To conserve battery life, however, only time and day-date can be displayed at low and high brightness. Seconds and set modes are always displayed at low brightness. The ICM7204 interfaces directly with a multiplexed sevent segment/four-digit- plus-colon common cathode LED dis- play, while the ICM7203 requires a nine segment display. The circuits are powered by two silver- oxide batteries and typically require 6 milliamperes per segment at 25% duty cycle with seven segments on. With high- efficiency magnified LEDs, this amount of current gives a very bright display, the manufacturer states. According to Intersil the technology that went into the ICM7200 family was gained directly from their experience developing the ICM7045 single-chip stopwatch microcircuit two years ago. This was the first such device to use ion implanted metal gate CMOS and direct drive of LEDs. It contains 4 modes of user-selected stopwatch functions as well as a full 24-hour clock. In designing it, Intersil had to include a standard mode; a sequential mode to time multi- legged events without restarting the timer at the beginning of each lap; a split mode to clock a complete event while displaying the times of each pro- gressive lap, and an event mode to time a complete event with intermediate interruptions. The circuit building tech- niques included use of computerized cir- cuit simulation to insure worst case operation over full temperature ranges. ‘Once we had mastered the problems in- volved with the ICM7045, development of the ICM7200 family of circuits was relatively straightforward.’ 1210 — elektor december 1975 calenc W.G. Paans Many mechanical and electro- mechanical clocks and watches a now provided with date indicatio Addition of a calendar to an elec tronic digital clock is a fairly simple matter, and the circuit given here gives the month as we as the date. As the calendar is an addition to a digi- tal clock a control signal must be de- rived from the clock to change the date. This can be derived from the changeover from 23.59 to 00.00 with a 24-hour clock, or if used with a 1 2-hour clock the changeover from 11.59 to 12.00 may be used. However, since this occurs every 12 hours a -r 2 flip-flop must be inserted between clock and calendar to give a pulse once every 24 hours. Like all the best calendars, this calendar knows whether the last day of the month falls on the 28th (February), the 30th or the 31st. Those who are worried about the 29th of February can add the optional leap-year correction circuit, in which case the calendar will not need to be reset until the year 2100, when a century correction (omission of leap- year) becomes necessary. The calendar is simply a logical exten- sion of the hours, minutes and seconds counters in the clock, but counting days and months instead. The resetting func- 1 - ( «Ot ' • * S' * 15 14 , 13 . ■ V. i 10 , ,9. r >•; r 4 V. t 0{ i w — < cool «* A 1667 1 1 1 2 l 2_ ‘4 1 5 '6 7 m ty - • & e D , •Y' tions are, of course, considerably me complicated due to the differing nu ber of days in each month. Since t year begins with the first month, ai each month begins with the first day, is not possible to use simple deca counters such as the 7490, which can reset to zero. Instead, presettat counters must be used, so that they c. be preset to one at the beginning of t year or month. A suitable choice is t 74163, which is a four-bit bina counter with synchronous preset ai clear. Two of these counters make i the days and tens of days counter, ai as the capacity of the 74163 is 4 b one of these IC’s will suffice for t ! to day decoder to ten day decoder to month decoder alendar elektor december 1975 — 1211 lonths counter. The pin configuration f the 74163 is given in figure 1. Points 3 watch with this IC are: ) unlike the 7490 it counts on a positive-going edge of the input waveform. ) for resetting purposes a logic ‘0’ is required. i) counting may only take place when there is a ‘1’ at both the enable in- puts P and T. r ) when there is a ‘O’ at the load input the count function is inhibited. The next positive-going transition of the clock input transfers information from the data inputs to the outputs. Counter Circuit he circuit of the counter section of the alendar is given in figure 2. IC7 counts le days, IC8 counts tens of days and 29 counts months. The enable inputs f all three counters are permanently igure 1. Pin configuration of the 74163 used i this design. igure 2. Basic circuit of the counting section f the calendar. igures 3 and 4. Two alternative decoding ircuits for the calendar. connected to positive supply, as are the clear inputs of 1C8 and IC9. The data inputs of the three counters must have the correct presetting data hardwired into them. The day counter is preset to 1 at the beginning of each month, so the A input is connected to positive supply and the B, C and D inputs to ground. The tens of days counter is preset to zero so all the data inputs are grounded. The month counter is preset to 1 , like the day counter. IC7 receives one pulse every 24 hours from the digital clock at pin 2 (clock input). This IC is connected so that it normally counts up to 9 before resetting to zero. When the count reaches 9 (binary 1001) the a and d outputs of the counter are high, so the output goes low, taking the synchronous clear input (pin 1) to ‘O’. On the tenth clock pulse the counter is reset synchronously to zero. This sequence is of course inter- rupted when the counter is preset to 1 at the beginning of each month. While the output of N5 is low this also holds pin 9 of N2 low, so its output is high. The tenth clock pulse which resets IC7 can therefore pass through N1 and N4 to the clock input of IC8. IC8 there- fore counts once every ten clock pulses. As stated earlier IC7 and IC8 must be preset at the beginning of each month, the count that they reach before this occurs depending on the number of days in the preceding month. It is evident from table 1 that with two exceptions the number of days in the month alter- nates between 31 and 30. The excep- tions are February, which has 28 days, August, which has 31 days after July’s 31 and December/ January similarly. It is thus possible to indicate the number of days required in each month with a flip-flop whose state is changed each month, the only corrections necessary being a) additional circuitry to detect when the month is February, and b) cir- cuitry to inhibit the changeover of the flip-flop at the July/August and December/January transitions. The flip- flop is IC6, and the ‘February detection circuit’ is contained in the dotted box. This part of the circuit operates as fol- lows: Assume that the next transition is from a month with 30 days to one with 31 days (say April/May). The Q output of IC6 will initially be high. When the count of IC7 and ICS reaches 30, out- puts Ba and Bb of ICS will go high. This means that all three inputs of N9 are now high so the output is low, taking the load inputs of IC7 and ICS to ‘O’. On the next clock pulse 1C7 and IC8 are thus preset. The output of N9 also holds the input of N3 low. The output of N3 is thus high, so the clock pulse is allowed through N6 to the clock input of IC9. Immediately IC7 and IC8 are preset the output of N9 goes high again. The output of N12 thus goes low. This is connected to the clock input of IC6 so the Oip-flop changes state and the Q output goes low. At the end of the next month, since the Q output of IC6 is low the output of N9 must remain high and the transition can- not take place on day 30. Instead N8 takes over, and on day 3 1 , when out- puts Ba and Bb and output Aa are all ‘1 ’, then the output of N8 goes low and the sequence repeats. Inhibition of the days units days tens tens «- months -> units flip-flop changeover during the July / August transition is accomplished by Nil. As July is the 7th month (binary 0111) outputs Ca, Cb, and Cc are con- nected to the inputs of N 1 1 . When these are all ‘1* (during July) the output of Nil is low. This takes the J and K in- puts of IC 6 low, inhibiting the change of state. N10 performs a similar func- tion during the December/January tran- sition. The ‘February detection circuit’ oper- ates as follows: the rather complicated looking array of gates performs the logic function: . . ‘February transition = Cb • Ca • Cc • Cd • Bb • Ad. Which is to say that the output of N13 goes low when the month is February (binary 0010) and the day is 28 (Ad = 1 , Bb = 1). The only point left to explain in fig- ure 2 is the presetting of the month counter. This is allowed to count up to 12. When the count reaches 12 out- puts Cc and Cd are high so the output of N 7 holds the load input low. On the next clock pulse the counter is syn- chronously preset to 1 . Display Decoding To provide an intelligible display the outputs of the three counters must, of course, be decoded. Two alternative de- coding circuits are given, and the choice is up to the constructor. Since the day counters have BCD outputs these are easily decoded, in both figures 3 and 4, using 7447 BCD/seven-segment decoder- drivers. Although the circuits shown use Minitron displays, LED displays may equally well be used (with appropriate segment series resistors). As the month counter counts to 12 in straight binary decoding is a little more difficult. The month decoding of fig- ure 3 operates as follows: when the month count is less than 10 the flip-flop comprising N 1 /N 2 is reset and the output of N 1 is low. This means that the data from 1C9 (connected to inputs a, b, c, d) is allowed through N4/I5, N5/16 and N 6 /N 8 (data on in- put a is connected direct to decoder) and is decoded into the months (units) display. The inputs of 1 1 , 1^ and I 3 are all con- nected to the output of N2, which is high, so their outputs are low and the ten month display is ‘O’. If a leading zero is not required on the ten month display then these inverters may be omitted. When the month count reaches 10 (binary 1010) the output of N3 goes low, setting flip-flop N1/N2. The out- puts of , I 2 and I 3 are now high, while the output of I 4 is low, so the ten month display is 1 . Month Number of days IC6Q outputs IC6 J and K inputs January 31 0 1 February 28 1 1 March 31 0 1 April 30 1 1 May 31 0 1 June 30 1 1 July 31 0 0 August 31 0 1 September 30 1 1 October 31 0 1 November 30 1 1 December 31 0 0 Table 1. Number of days in each month and the corresponding states of flip-flop IC6. Figure 5. Date setting circuit. Figure 6. Showing the addition of automatic leap-year correction to the calendar. The low output of N2 inhibits the data on the b, c and d inputs from passing through N4, N5 and N 6 . The high out- put of N 1 allows the data on the c input through N7. During months 10 to 12 therefore inputs C and D of the 7447 are low, the B input receives data from the c output of the counter, while the A input continues to receive ‘a’ data. During month 10 (binary 1010) the 7447 receives input code 0000 and thus the display is 0. During month 1 1 (bi- nary 1011) the input of the 7447 is 0001 (display 1 ) and during month 12 (binary 1100 ) the input code is 0010 and the display 2. During this period the ten month display is, of course, always 1 . At the end of the year, when the month counter is preset back to 1 , the d input to the decoder goes low. This transition is differentiated by the 10 k and 100 p on the input of N2, pro- ducing a short negative-going pulse that resets the flip-flop. The month decoding of figure 4 oper- ates on a somewhat different principle Basically, for counts of less than 10 the months units are decoded by IC2. For counts from 10 to 12 the months units decoding is transferred from IC2 to IC3. while IC2 counts the tens of months The circuit operates as follows: foi month counts below 10 flip-flop N2/N? is reset, so the output of N2 is low. T1 and T2 are turned off and IC3 is in hibited by a ‘ 0 ’ on the blanking inpui (pin 4). IC2 thus decodes the data frorr the output of the month counter. When month 10 is reached the output of N 1 goes low, setting the flip-flop an< blanking IC2. The display is now drivei by T 1 and T2, which are turned on causing a 1 to be displayed. The lov state on the blanking input of IC3 is re moved, and this decoder receives daL on its A and B inputs from the a anc endar elektor december 1975 — 1213 outputs of the counter. Thus for :>nths 10, 11 and 12 IC3 receives hi- ts 0000, 0001 and 0010 respectively. the beginning of the new year the p-flop is reset in a similar fashion to it of figure 3. ita Setting is is accomplished by the circuit of ure 5. With SI, S2 and S3 in the pos- m shown the three flip-flops corn- sing N4-N9 are reset. The outputs of , N6 and N8 are thus high. One of i inputs of N3 is held low by N5 so its tput is high, and both inputs of N1 1 low, so its output is high. Two of i inputs of N10 are high so the 24- ur pulses connected to the other in- t can pass through N10 and N12 to 5 day counter. SI is now changed over flip- p N4/N5 is set blocking the 24-hour ses through N10 and allowing a fast se train from the astable multivi- tor N1/N2 through N3 and N 1 2. This i be used for fast setting of the calen- to some value near the required date. SI is now reset to its original position i S2 is changed over N10 is again eked by a ‘0’ on pin 2. Pin 9 of N1 1 now high, so the calendar may be danced slowly to the correct date by gle pulses through Nil, produced by ornately setting and resetting flip- p N8/N9 with S3. Flip-flop IC6 must, course, be set to the correct state for month, according to table 1. ap Year Correction e automatic leap-year correction is remely simple, and consists basically a divide-by-four counter that counts years and gives February an extra r every fourth year. The addition of leap-year correction to the calendar ;uit is shown in figure 6. The counter isists of two JK flip-flops (IC10). ce a year this counter receives a pulse m output d of the month counter K Normally at least one of the Q out- s of IC10 will be low and the base of will be held down via one of the two des connected to these outputs. T1 1 thus be turned off. During the irth year both these outputs are high, loving the constraint on the base of . T1 is now turned on and off on jrnate days by N21, whose input is inected to output a of IC7. On odd 's T1 is off, and on even days T1 is holding pin 12 of N13 low. When 28th of February arrives the ‘Febru- detection circuit’ will try to operate, since it is an even day T1 is turned and the output of N13 remains high, vill not go low until T1 turns off on 29th, and on the next clock pulse day counters are preset. nclusion jse circuits should enable the con- lctor to add a calendar to most digi- clocks. The construction and type of slays used are left to the reader’s ividual preference, and presumably i be chosen to match the existing ck. N G 1214 — elektor december 1975 santatroni< photofinish F. Ansoms When several mini racing cars are driven along a number of parallel tracks it is sometimes difficult to spot the winner. Heated discussions about the results of the race can be avoided by using this simple electronic photofinish system. The circuit is of a quite simple design. The adjustment potentiometer, PI, (figure 1) which together with the LDR controls the base bias of transistor T1 is so adjusted that the transistor is just cut off. When a racing car intercepts the light beam, the resistance of the LDR momentarily increases, so that the base voltage of the transistor also . increases, with the result that the latter J turns on and lamp LI lights up. The resulting current causes a voltage drop of about 1 V across resistor R3 so that T3, too, turns on. The collector voltage of T3 is now about 0.3 V, so that the other thyristor cannot fire, since the cathode voltage is higher than the gate voltage. After the final heat, the circuit is reset by briefly interrupting the supply voltage by pushing button SI. If this is often forgotten, automatic resetting after each round can be achieved by fitting a microswitch under the track some distance before the finish line. i r 1 R <£> 6V ' 100mA LI Thl L_ n 100k T 1 PI R4 6 V 100mA TUN LDR 1 Th 2 LDR 1 3L H 14 T2 R 1 SI reset D1 D2 1 2x -- 1 N4001 100k R5 P2 TUN LDR 2 T3 R2 R3 TUN 6V 200mA 9367 1 NJ ' CN M *♦ 9367 2 LDR 2 If more than two tracks are used, which makes it more difficult to see who was first, the circuit can easily be extended by parallel connection of the section surrounded by the dashed line. And as a last practical hint: the LDRs and the lamp should be mounted in pieces of PVC tubing. The LDRs are then not influenced by ambient light. The supply voltage depends on the types of lamp and can be chosen about 1 volt higher than the nominal lamp voltage. jtomatic barrier-control for model railway level-crossings elektor december 1975 — 1215 t. ter Mijtelen automatic barrier control for model railway level - crossings The circuit described here provides control of the 'automatic barrier' type of level crossing (or indeed of the old-fashioned gate type). It gives a realistic simulation of the visual and audible warnings. There is no limit on the train length, and the system will operate when trains are passing in both directions. 'igure 1 shows the general principle of peration. In the ‘rest’ condition the arriers are, of course, open and no Lghts show. Light dependent resistors LDR1A, IB, 2 A and 2B) are situated •y the track some distance on either ide of the crossing, and these normally eceive light from lamps L4 and L5. Wien a train approaches it will block he light from either LDR1A or IB, .epending on its direction of travel, 'his initiates the following sequence of vents. Firstly amber lights (LI) light or several seconds and a bell sounds, 'hen the amber lamps extinguish and he bell stops, red lights L2 and L3 start o flash alternately and the barriers escend. After the train has passed, (this fact is determined by LDR2A (or 2B) being re-illuminated) then the red lamps are extinguished and the barriers are raised. Circuit The most important part of the circuit is the train detection logic given in figures 2 and 3. This ensures that the barriers are lowered as a train ap- proaches, and are not raised again until tlie train (or trains if there are more than one on the crossing at the same time) have left the crossing. The logic Figure 1. The general layout of lamps, LDRs and barriers. will operate correctly regardless of the length of the train i.e. it makes no dif- ference if the train is either longer or shorter than the distance between LDR1 and LDR2. The circuit of figure 2 is duplicated for the up and the down line, while figure 3 shows how the two circuits of figure 2 are interconnected. The circuit of figure 2 operates in the following manner: when a train approaches it blocks the light from LDR 1 , whose resistance thus increases, causing the input of N1 to go high. The output of N1 thus goes low, setting the Hip-flop N3/N4 and initiating the gate-closing sequence. Once the Hip-Hop is set LDR1 has no further effect. When the train i LDR1A 14 □ LDR2B 1216 — elektor december 1975 automatic barrier-control for model railway level-crossing Figure 2. The train detection logic for om line. Figure 3. For two-way traffic, as in figure 1 two circuits as in figure 2 must be intercon nected. Figure 4. This circuit controls the lights am the opening and closing of the gates. reaches LDR2 flip-flop N6/N7 is set in ; similar manner. The B input of N5 i thus held high by the Q output of thi flip-flop. When the end of the train ha passed LDR2 is re-illiminated so tin input of N2 therefore goes low and thi output goes high. Since both inputs o) N5 are now high the output goes low resetting flip-flop N3/N4, which in turi resets flip-flop N6/N7. Since the gates must be activated by ; train on either the upline or the dowi line the circuit of figure 2 must b< duplicated. Figure 3 shows how the tw< circuits are interconnected. D1 and D. perform an OR function. When a trail approaches on either the up line or tin down line (or both) one of the Q out puts will go low, taking the S input o flip-flop N8/N9 low and setting it, thu initiating the gate closure sequence. I however there are two trains on thi crossing the gates must not open unti both have left, so both inputs of NIC must be high before the output can gc low, thus resetting the flip-flop am initiating the gate opening sequence 05V 4044 4 05V urning off thyristors elektor december 1975 — 1217 W. Back turning off thyristors When thyristors are carrying A.C., they will turn off at every zero- crossing — which can be a nuisance. When carrying D.C., however, they won't turn off at all — which is worse. This article takes a basic look at how to cope with the latter problem. 3ate opening and closing sequence rhe circuit that controls the lights and ipening and closing of the gates is given n figure 4. When a train approaches and he Q output of figure 3 goes low this riggers monostable IC1. While IC1 is in ;he triggered state T1 and T2 are turned m, the amber lamps (LI) light and the :>ell rings. When IC1 resets the Q output *oes low. This transition is differen- tiated by C2 and Rl, producing a short, aegative-going pulse that sets flip-flop \ T 1/N2. Until this flip-flop is set the nputs of the astable comprising N3/N4 ire held low by the output of N 1 , so the Dutputs of N3 and N4 are high and amps L2 and L3 are extinguished. When the flip-flop is set the inputs of N3 and N4 are taken high, so the astable starts to oscillate and lamps L2 and L3 flash alternately. rhe same negative-going pulse also triggers IC2. While IC2 is triggered T3 is turned on pulling in relay RLA, whose :ontacts are used to switch the gate motor (or solenoid). The period of IC2 may be adjusted by P4 until it is just long enough to allow the gates to close, thus avoiding unnecessary dissipation in the relay coil. After the train has passed the Q output of figure 3 goes low, resetting flip-flop N3/N4 so that the lights stop flashing. It also triggers monostable IC3, which acti- vates RLB to open the gates. The period of IC3 may also be adjusted by P5. Setting up procedure It is evident that the LDR’s must be mounted sufficiently far on either side of the crossing that the train does not arrive before the gates close. The LDR’s should be mounted in tubes to screen them, as far as possible, from ex- traneous light. PI and P2 should be adjusted so that, whatever the ambient lighting conditions in the room, N1 and N2 will switch reliably when a train passes. P3 is used to adjust the delay between the approach of a train being detected and the closing of the gates i.e. the time for which the bell rings and the amber lamps are lit. H There is a growing tendency in elec- tronics for electromechanical switches to be replaced by semiconductor devices. In light current applications transistors are now capable of switching currents which a few years ago would have re- quired the use of relays, whilst in power engineering thyristors can switch loads that would normally require fairly hefty contact breakers, without the associated problems of contact wear due to arcing. A.C. current control with thyristors is fairly simple, but this article takes a basic look at some methods of switching D.C. currents with thyristors. Switching of A.C. currents with thy- ristors is relatively easy. As is well known, in its non-conducting state a thyristor will block a potential applied to it in a forward direction (i.e. positive to anode, negative to cathode). How- ever, application of a positive trigger pulse to the gate will cause it to con- duct, and it will remain conducting even after the gate input is removed. The only way of returning the thyristor to its blocking state (unless it is a gate turn-off device) is to reduce the current through it below a critical value (the holding current) for a period of lime de- pending on the device in question (the turn-off time). In A.C. circuits of course, the current through the thyristor attempts to re- verse during the negative half-cycle of the waveform, but since a thyristor will not conduct in the reverse direction it turns off at the zero-crossing point of the waveform. No such convenient trick occurs in D.C. circuits. In D.C. circuits the only two methods of turning off a thyristor are: — break the circuit so that the current is interrupted. — momentarily divert the current from the thyristor so that it will turn off. The first proposition is obviously im- practical as breaking the circuit would require a switch or relay capable of switching the current that the thyristor was carrying, which defeats the object of the exercise. The second proposition brings us to the principle of capacitor commutation. If a capacitor is charged and then connected so as to reverse bias the thyristor, then the load current will see the capacitor as a very low im- pedance into which it will momentarily flow, and the thyristor will turn off. Figure 1 is the most basic example of such a circuit. When current is flowing in the load Rl then Cl will charge with the polarity shown via Rl. When the switch S is closed the capacitor is connected with reverse polarity across the thyristor. The load current sees this as a low impedance and is momentarily diverted into it. The thyristor mean- while is reverse biassed by the voltage across the capacitor and turns off. This circuit is clearly not of much practical use, since it also requires a switch, but it does illustrate the prin- ciple. A more practical variant of the circuit is illustrated in figure 2. This uses an auxiliary thyristor to switch in the capacitor. Rl is chosen such that after Till has turned off and Cl has charged through Th2 with the opposite polarity to its original charge, then the current flowing through Th2 via Rl must be less than the holding current of Th2 so that this thyristor will turn off. This clearly places a lower limit on the value of Rl. The lowest value of Cl is also limited by the time it takes to discharge to zero volts on turning on Th2. This must be greater than the turn-off time of Till as otherwise Cl will have discharged and begun to recharge in the opposite direc- tion before Thl can turn off. The maximum rate at which the circuit may be switched on and off is deter- mined by the time taken to recharge Cl through Rl after Thl has been turned on again. Even with the minimum permissible values for Cl and Rl the switching rate is limited to a few hundred Hz in most instances. A method of increasing the maximum switching rate is to use capacitor turn-off with a ringing choke, and the basic cir- cuit is given in figure 3. If Th 2 is in- itially turned on then Cl will charge through Th2 and Rl, until it has 1218 — elektor december 1975 turning off thyristors three tracer acquired full supply potential, when Th2 will turn off. If Thl is now turned on then a parallel resonant circuit con- sisting of L and Cl is completed, which starts to ring due to the initial charge on Cl. During the first half-cycle current Hows Figure 1. Capacitor commutation using a switch to connect the capacitor across the thyristor. Figure 2. Using an auxiliary thyristor to switch in the commutation capacitor. Figure 3. Using a ringing choke arrangement to increase the maximum switching rate. 3 through Thl, D and L and reverse charges Cl. The diode prevents Cl from attempting to discharge back through L and Thl. Of course, a D.C. current also flows through Thl into the load. If Th2 is now turned on the reverse-charged Cl is connected across Thl, turning it off. With this method switching rates of up to 1 kHz can be achieved. Calculation of commutation ca- pacitor When the auxiliary thyristor is turned on a negative voltage appears across the main thyristor. This reduces to zero as the load current flows into the capaci- tor and, provided the main thyristor actually turns off, the voltage on the ca- pacitor will eventually assume full posi- tive supply voltage, at which point the auxiliary thyristor will turn off. It is evident that the main thyristor must turn off before the voltage on the commutation capacitor assumes a posi- tive value, or it will never turn off. This means that the time taken for the volt- age across the capacitor to reach 0 V must be greater than the turn-off time of the main thyristor. Now this time is determined by two factors, the charging current flowing into the capacitor through the load and the capacitance of the capacitor. Initially current is being driven through the load by a voltage 2 V^. (supply volt- age plus the initial voltage across the ca- pacitor). By the time the voltage across the capacitor has reached zero the cur- rent is being driven by the supply volt- age V b . 2 v . Initially therefore the current is — — , Vu K L and finally it is — . The average current is therefore approxi- 1 5 Vh mately — — This is of course a gross r L approximation as it assumes linear charging, but it is adequate for calculat- ing the commutation capacitor. Now since Q = CV = IAt. where Q is charge on capacitor. C is capacitance V is voltage on capacitor (= V^). I is average charging current r _ 1.5 V b , (= Rl Then At is charging time (= turn-off time of thyristor). 1.5 V b • At = CVb Therefore Rl c = 1 .5 At Rl This is the minimum value of capacitor to turn off current flowing through a load Rl. In practice the value of C should be slightly larger than this to ensure reliable commutation. The commutated turn-off time of the thy- ristor (usually designated tq) can be obtained from the manufacturer’s data sheets, and the load Rl is of course known, so C can easily be calculated. R. Sintic three tracer It is possible to display more than one trace on the screen of a single-beam 'scope, using a fast electronic switch. This design will interleave three traces, which may be of analogue or of digital signals. The practical results appear quite acceptable for such a simple set-up. The circuit The generator for the switching fre qucncy is a discrete-component shift register which is arranged to ‘chasi its tail*. The circuit around Tl, T2 and T2 in figure 1 is the actual shift register A switch (SI) changes the repetitior rate from a low value (200 Hz) to i high one, by applying bias to the diodej D 1 , D2 and D3. The choice of switching frequencies makes it possible to display input frequencies between 20 Hz anc 500 kHz. The pulse-shapers T4, T5 and T6 im prove the rise time of the thre iree tracer elektor december 1975 — 1219 1 bitching waveforms. The same circuit Pitches the DC level of the output ac- >rding to the required position of the iree traces on the screen. Dtentiometers PI to P3 in the collector rcuits of T1 to T3 achieve the DC tting for trace position by varying ie negative level in the three rec- ngular signals. As the signals in turn xome negative, the diodes D4 to D6 iss them to the output RC network, lie signal-switching is done by diodes 7 to D9. They in turn pass the AC miponents of the input signals at Al, 1 and Cl to the output, o achieve a high input impedance id to compensate for the insertion ss of the circuit it is necessary to ovide each input channel with a pre- nplifier. The gain of each of these pre- nplifiers can be preset, for calibration irposes. reamplifiers with input ttenuators simple preamplifier of high input im- ^dance can be made using a JFET. arrent-dependent negative feedback is >plied to improve the linearity. Cali- ation is achieved by presetting the ltput voltage of each channel, ie input attenuator in each channel frequency-compensated, to enable st rise-time waveforms to be repro- iced without distortion. The imponent values given in figure 2 e nearest ‘preferred value’ ap- oximations. They are intended for plications where absolute accuracy Figurel. Circuit diagram of the three-trace switch. A discrete-component shift register, arranged to 'chase its tail', continuously produces three evenly-spaced sequential pulses. Figure 2. Preamplifier with input attenuator. Three of these stages are needed; one to drive each of the switch-inputs Al, B1 and Cl. (see figure 1). is not so important. In other cases it will be necessary to make up the attenu- ator with close-tolerance precision resistors. The input blocking capacitor, shown dotted, will only be needed if small AC voltages superimposed on large DC volt- ages - such as rectifier ripple - are to be observed. 1220 — elektor december 1975 quadi-complimentan quadi- compIimEntary In the new Quad 100 Watt amplifier design, a special negative feedback arrangement enables the output transistors to be zero biased without creating non- linearity problems. The transfer characteristic of the output stage is independent of the (non-linear) characteristics of the active components, provided the values of four impedances in the output stage are suitably chosen. Zero bias for power transistors in the output stage of an amplifier is advan- tageous from the point of view of thermal stability. However, zero bias will result in a ‘dead zone’, which in turn results is crossover distortion. A solution to the problem is to bias the output stage into class AB so that the characteristics of the alternate output transistors overlap, resulting in a more or less linear behaviour around the zero crossings of the signal. For success, this strategy relies on the symmetry of the alternate output stage halves. Further- more, the influence of temperature on the bias setting is of utmost importance. If the base-emitter junctions of the output transistors are part of the bias circuit, the bias will be influenced by the junction temperature of the output transistors. Temperature compensation schemes can never completely eliminate this problem, if only because junction temperature and case temperature are not the same. For this reason, the bias will vary with music program dynamics, resulting in momentary non-linear be- haviour. Thermal processes - including thermal compensation - are compara- tively slow! Things can only become more difficult in the new generation of high power amplifiers rated from 100 Watts to 800 Watts (Crown, Phaselinear, Luxman and others). In order to meet the required voltage and current demands, each ‘output transistor’ has to be composed of several transistors in parallel. Sometimes it may even be necessary to use two transistors in series for each ‘parallel transistor’. It will be obvious that it is very difficult to obtain a constant bias when this is influenced by say eight very hot base-emitter junc- tions. For this reason it is common practice to zero bias all the output tran- sistors. Figure 1 shows the functional block diagram of a zero bias output stage. Block B has a ‘dead zone’, i.e. an area around the crossover point where it will deliver little or no output current. Block A supplies current into the load RL through one of the resistors R, depending on the polarity of the input voltage. If the voltage drop across R exceeds the threshold voltage of block B (Vp>) either the NPN or the PNP power transistor is turned on. Figure 2 shows the characteristics of A, B and the resulting load current II versus the input voltage. In a good design the characteristic of B is quite linear as a result of local current feed- back by emitter resistors. Generally speaking the driver stage A has an optimal class-AB bias. Provided the mutual conductance of A (the slope d(lA)/d(Vi n ), which is usually inversely proportional to R) is in the same order of magnitude as the slope of B, d(lB)/d(Vi n ), the resulting bend of the load current characteristic will be quite small. So, in the high power amplifiers adopting this current dumping strategy, crossover problems are mild and occur at higher output levels. Overall negative feedback can usually straighten things out. However, it would be even better if we could get rid of this bend altogether. Referring now to figure 3: Above the threshold voltage +Vp (and below — Vp>) the contribution of A to the load current is limited. It would be ideal to obtain a zero contribution of A to Ik (/3 = 0), because this would mean that the dissipation of the driver stage A is as low as possible. Unfortunately, this would also mean that the voltage drop across the resistor R is limited to ± Vp>, making it impossible to turn on the transistors within block B any further. So, we will have to look for a solution which gives the smallest possible value for p. The Quad amplifier Figure 4 shows the basic principle of the Quad amplifier, which has been devel- oped by P.J. Walker and M.P. Albinson of the Acoustical Mfg Company. A is a class-A amplifier with a high open loop gain (A 0 ). It is capable of delivering an output power of about Figure 1. Functional block diagram of a ver high power output stage. The transistoi within block B are zero biassed. The inpu voltage V, n is taken relative to the *hot' sid of the load. Figure 2. The currents 1(A), 1(B) and II plo ted as a function of the input voltage. I order to make the bend in the Ip characte istic as small as possible the slope of A shoul be as high as possible. Figure 3. By combining the 1(A) and I (E characteristics a straight Ip curve is obtainei The angle (3 should be minimal in order! keep the dissipation of A as low as possibli Figure 4. The basic principle of the ne’ Quad 100 Watt amplifier. All voltages ai relative to the *hot' side of Zp. ladi-complimentary elektor december 1975 — 1221 Watts. This amplifier supplies current ito the load (Zl) through Z 3 . As soon ; the threshold voltage of either one the zero biassed power transistors T 1 id T2 is exceeded, the transistor in iestion is turned on and supplies a irrent I 4 through Z 4 into the load, oth Zi and Z 2 are large compared to 3 and Z 4 . he negative feedback arrangement is iite unconventional. The voltage at the verting input of A with respect to the ot’ side of Zl is the sum of the iltage drop I4Z4 across Z 4 and a action Z 1 /(Z j + Z 2 ) of the base-emit- r voltage of Ti — T 2 . This means that jgative feedback is not only derived om the output of the amplifier, as in mventional circuits, but also from the put voltage of the (non-linear) output age. It will be shown that this feed- ick arrangement may lead to a load irrent II that is independent of the characteristics of Tl and T2. Re-arranged: Calculations Referring to figure 4, the current II through the load Zl is calculated as a function of the input voltage Vi n . All voltages are with respect to the ‘hot’ side of Zl. For amplifier A, with an open loop gain of A 0 , the following holds: V+ = V in 0) V- = I 4 Z 4 + (Va - I 4 Z 4 ) (2) Z.J + l 2 Va = A 0 (V + -V_) (3) (1) and (2) substituted into (3): VA = AoVi n — AoVa - — ^ — Z 2 -A 0 l4Z 4 7-^V ( 4) L\ t L 2 V A (1 + Zi Z, + Z 2 — AoVj n — A0I4Z4 Ao) - Z 2 Zl + z 2 (5) If A 0 » 1 then: so: v _Zi +Z 2 w Z 2 f „ * A ? # Vi n — ^ ~ * I4Z4 (6) U = _ v A _ Zi + z 2 Z 2 Z 4 Vin ~ I 4 Z,Z 3 Z1Z3 (7) !l = h + I4 (7) combined with ( 8 ) results in: (8) , L = Vin ^ + l4(.-^)(9) z,z 1^3 ZjZ : 1222 — elektor december 1975 fuse indicato Something wonderful happens when: Z,Z 3 =Z 2 Z 4 (10) (Just like the Wheatstone-bridge!) In that case: lL= V in Zi + Z 2 Z,Z 3 = V in (= L + 7 L )!!!! (ID In other words: The current through the load is independent of any active para- meter. The dead zone of Tj - T 2 doesn’t appear in the load current. What happens is that this feedback arrangement introduces non-linear feed- back in such a way that the relationship between load current and input voltage becomes linear. From formula (6) it follows that the drive voltage Va to the output stage Tj -T 2 depends on the value of I4Z4. In the Quad design, the impedance Z4 is a 0.3 /ill inductor; Z3 is a 100 £2 re- sistor, Zi a 3.3 k resistor and Z 2 a 10 pF capacitor. Condition (10) is met. In the emitter follower configuration a resistor between the inverting input of A and ground is added. The output impedance of the amplifier is Z3 and Z4 in parallel. Summing up, a lot of typical class-B problems are solved. The design has no bias stability problems, no bias adjust- ments, no bias at all. A reasonable imbalance in the values of Z1-Z4 due to component tolerances in claimed to have only a minor effect on the distor- tion. However, we wonder what the influence of the DC-resistance of the Z4 inductor might be - it causes a departure from condition (10), which can, of course, be solved by placing a resistor in parallel with the Z 2 capacitor. Another possibility would be to use resistors for the impedances Z1-Z4. Z4 could be say a few tenths of an ohm, composed of non-inductive carbon resistors. Amplifier A could be preceded by an amplifier, and an overall feedback of say 20 dB would result in a virtually zero output impedance and minimise the effect of imbalance in the resistors R1-R4. Nevertheless, referring to formula (11), one can say: Quad erat demonstrandum! H Literature: ‘ Current dumping amplifier * 1 2 3 4 * * * * 9 by P.J. Walker and M.P. Albinson. (Lecture during the 50th A.E.S. Con- vention , London 1975.) elektor services to readers With reference to the column that appeared under this heading in elektor no. 3 (April 1975), we should like to amplify the following points. eps service Publication of a p.c. layout does not automatically imply that we supply a board for that design. We can only sup- ply boards which appear in the eps list in the current issue of elektor. To avoid errors please quote board name (as in eps list) and part number when ordering. technical queries 1 . Telephone queries can be accepted only on Monday afternoons between 14.00 and 16.30. At other times the editorial staff are busy writing your next magazine and are not available. 2. The service is for genuine technical problems only. Many queries are about sources of supply for com- ponents and we can answer these only by asking readers to contact advertisers in the magazine. Remem- ber, most advertisers do not advertise complete stocks, so please contact them before overloading the tq ser- vice. If you still have difficulty, then contact us. 3. When writing to the tq service please enclose a stamped, addressed envel- ope, otherwise we cannot guarantee a reply. 4. When writing to several different de- partments please enclose separate letters to each department, as other- wise delays may result while your letter is processed by each depart- ment in turn. N J.W. van Beek fuse indicator In this circuit, the neon indicator lam] shows whether or not the power is 01 and whether or not the fuse is blown. As long as the power is on and the fus- is intact, the neon lamp will drav current through the fuse, D2 and th built-in series resistor. It will bur: brightly to indicate that all is well. If the fuse is blown, however, curren can only flow through D1 and Rl. Thi current will charge Cl until the ignitio voltage of the neon lamp is reached. Th lamp will light up. It will now dra\ enough current to discharge Cl until th extinction voltage is reached, when upon the lamp will go out agair Cl recharges through Rl, and the cycl repeats. The result is that the neo lamp will flash continuously as long a the power is on. The only critical points in this circuj are the resistors. The value of Rl mus be so large that current flowing througl this resistor into the neon lamp i insufficient to keep it ignited. On th* other hand, the built-in resistor shouh be small enough to discharge Cl fairl; rapidly but not so small that the lamj will ‘burn out’ when fed directk • through D2 (actually, a neon lamj doesn’t burn out - it can progressivel; darken as the electrode materia ‘migrates’ to the inside of the glas envelope). h atronics elektor december 1975 — 1223 t ■•WVsi / fit; 0. ©■ o o. i&mm v .• • • 9366 As the days were once again growing shorter, a designer had a nostalgic dream — about the good old days with the whole family gathered around the open coal fire. Something, he felt sharply, was missing from his thermostatically- controlled centrally-heated home ... Something that would roast chestnuts ... The oven is constructed in an old tea- or biscuit-tin, with lid. An inner compart- ment, made from aluminium or copper sheet, is fitted inside the tin - well insulated and adequately supported by means of a blanket of glass-wool. A heating element is mounted under- neath (or on top of) the floor of the insulated inner compartment. This element may conveniently consist of a few wirewound resistors — but the thermostatically-controlled centrally- heated version would use the dissipation from an LM 395 regulator. B40C2200 -018V -©15V 2A LM 395 9366 1 0 - ©- -0- R. J 9366 2 chestnut oven The maximum power required for an oven of 50 cubic inches will be about 50 watts. The circuit for the simple ver- sion is given in figure 2. A tapped trans- former (to enable the heating power to be adjusted) directly feeds the power- resistors. The transformer must be rated for at least 50 voltamperes (in the example) and the required resistor value V 2 follows from R = where V is the highest available secondary voltage. Since 10-watt resistors are readily avail- able, and also to spread the heat pro- duction over a larger area, it is rec- ommended that five resistors be used in parallel. The individual resistors should then each of course have five times the value R determined above. The automated version of the chestnut oven (see figure 1) uses an LM 395 integrated regulator. This device is in fact a voltage stabiliser, provided with a current-limiting circuit and a thermal shut-down. When the device operates into a dead -short it will work as a con- stant-current sink (at the limit-current of 2 A), dissipating Vrms x 2 A watts — at any rate until the temperature reaches 170°C, the shut-down tempera- ture. The supply voltage determines the ‘on’ dissipation, while the device deter- mines its own duty cycle as required to maintain 170°C. There are two ways to destroy an LM 395: one can connect a supply of the wrong polarity, or one can apply a (peak) voltage in excess of 36 volts. The circuit requirements are therefore simple: the transformer secondary voltage must first be passed through a full-wave rectifier (in this case a bridge type), and the applied peak voltage must remain below 36 V under the highest mains supply value that can possibly be encountered. A safe voltage rating for the transformer secondary would be 22 V. At 20% mains over- voltage (what is the chance of that occurring?) the peak voltage would be 37.2 — which means 36 V after the rectifier. The chestnuts pop after a half- to one hour. smUdtSHS 1224 — elektor december 1975 cd-4 Victor Company of Japan, Limited (JVC) Over five years have elapsed since the CD-4 system was announced as a means of achieving quadra- phony from a disc record. The worldwide popularization of the CD-4 system is now making rapid progress, thanks to various factors such as the wider variety of soft- ware — about 1000 CD-4 albums have been released — improved, more compact cutting equipment, the development of high perform- ance PU cartridges, the high level of integration used in the demodu- lation circuit, etc. However, there seem to be various misunder- standings of the CD-4 system be- cause of some unfortunate occur- rences at the initial stage: played back sound was unsatisfactory be- cause of inferior service to end users during the course of the development, parties with differ- ing interests issued misleading publicity, etc. For this reason, in this paper, we would like to restate our policy on CD-4 and describe the present state of CD-4 technology to give readers a fuller understanding of the CD-4 system. The CD-4 system was undertaken to make possible a disc record which would transmit accurately the 4-channel musical information that the people who make the record — musicians, directors and mixing engineers — want the music listeners to hear. Figure 1 shows a process through which the four separate channel signals can be transmitted using the disc record as the medium. The 4-channel program source is achieved by collecting sound from the sound field where the live performance is going on, converting these sounds to electrical signals and then composing the 4-channel signals with the mixing console. The various steps in this chain of events are carefully controlled so that the end result satisfies the producers. At this stage, the producer’s sole inten- tion is to achieve the desired artistic effect. The end result of this step is a combination of the musical techniques and artistic expression of the performers and the technical abilities of the mixing engineers; it is monitored always paying careful consideration to the listening conditions in which the average user will hear it. The output is the 4-channel master tape (A) which represents the sum total of these peoples’ efforts, which they wish to present to the public. Therefore, it is essential that the record- ing system transfers all the information to the master tape and that the perform- ance of the mixing console — in the artistic stage described above — is such that the output is loss-free with respect to the input and that the physical prop- erties of the sound such as phase and amplitude of the input signal can be controlled to match the intentions of the producer exactly. The 4-channel master tape is not the end product which the user buys; a suit- able means of duplicating it must be found. This is the CD-4 disc record (B), which has the same channel capacity as tape (A). The fact that the channel capacity of (A) is equivalent to that of (B), means Editorial note After publication of our article on quadrophony ('Quadro 1 -2-3-4 . . Elektor 1, p. 33) we received a request from JVC/Nivico to give them an oppor- tunity to comment on it. We agreed to this, subject to the proviso that the proponents of the other three systems (SQ, QS and UD4) were given an equal opportunity. To this end, we sent copies of this article and of the second article ('Quadro in practice', Elektor 4, p. 646) to all parties concerned, explaining the situation and asking for their comments. However, to date the only comment we have received for publication comes from JVC — in spite of repeated written and personal requests to the other parties. We now feel that it is only fair to JVC to print their reaction in full, even though we cannot present a parallel dis- cussion of any of the other systems. that the same quality of sound can be played back. In this way the producer’s intentions are transmitted exactly as they are to the listener. While every effort has been made to achieve this (A) = (B) concept in the CD-4 system, another consideration which was not neglected in any way was sufficient compatibility with conven- tional stereo and mono playback equip- ment. Therefore : 1 . It must guarantee sufficient channel separation when the CD-4 record i< played back in stereo and a natural sound image must be obtained. 2. There must be no loss of musical information when the CD-4 record is played back in mono. If sufficient consideration of these factors is not given when the signals foi the 4-channel disc record are being com- posed, channel separation in stereo play- back will deteriorate and phase differ- ences will occur, thus narrowing the sound field and creating out-of-focus sound images resulting in music which is fatiguing to listen to. If this happened, the (A) = (B) concept would not achieve its full potential. Another basic consideration in the de- sign of the CD-4 system is that the cost to the buyer must be minimized as far as possible, without detriment to the technical and artistic considerations out- lined above. Keeping this principle in mind, the composition of the record sig- nals in the CD-4 system were simplified as much as possible. Manufacturers have taken various steps in the past to lessen the load on the buyer; compensation for distortion and losses, integration of the detector cir- cuit, improvement of styli (especially the SHIBATA stylus) and the develop- ment of low-cost PU cartridges for CD-4 using this Shibata stylus. The will of the developers to achieve the two basic concepts ‘discreteness’ and ‘compatibility’ are reflected in the fact that the new system was named CD-4; C for compatibility between the different playback 'modes and D for the discrete- ness of the signals, inherent in the cd-4 elektor december 1975 — 1225 recording studio 4ch master tape record factory ‘/////////////////////////////////A m oooo 0 console user's listening room record groove direct signal CHI +CH2 7V^ CH3 + CH4 modulated carrier 19dB CHI — CH2 CH3 — CH4 30 45 f (kHz) - 9384 2 SHIBATA STYLUS record having equivalent channel ca- pacity. Details of the CD-4 record Since the CD^t record is required to transmit four signals perfectly, it is necessary to double the channel ca- pacity when compared with conven- tional stereo records. However, changing the physical shape of the disc record would make it incompat- ible with stereo and mono records and would greatly increase cost to the user because a new, complicated transducer would be needed. This is the reason why the frequency superimposition technique — ‘base band signal’ + ‘carrier’ - was introduced for the CD-4 system. At this stage, the points under consider- ation were that the modulation and de- modulation of the signal to be super- imposed had to be relatively straightfor- ward and the medium to be used for the recording was the standard disc record. Through careful investigation and the comparison of various modulation sys- tems, angular modulation was adopted as being most suitable for the CD-4 sys- tem. Before entering into the problems involved in modulation, it is first necessary to devote some time to de- scribing the composite signals. If full compatibility is to be maintained, the following requirements must be met by the four separate signals, Lp, Lg, Rp and Rg which corresponds to (A). 1. In stereo playback, none of the four signals must be lost. The left signals, Lp and Lg, must be reproduced from the left channel speakers and 1226 — elektor december 1975 cd-4 cji ±coO PLL the right signals, Rp and Rg, must be reproduced from the right channel speakers. There must be no crosstalk or phase difference between the left and right channel signals, as this would cause the expansion of the sound field and the focus of the sound images to deteriorate. There- fore, the left and right signals must be transmitted discretely, that is in- dependently of each other, and with the same phase. 2. In mono playback, none of the four signals should be lost. The simplest signal composition which satisfies these requirements is: Base band Carrier band Left channels Lp + Lb Lf — Lb Right channels Rp + Rb Rf — Rb The simplicity inherent in this signal composition guarantees that the cost of the recording and playback equipment will be minimized. With angular modulation, the variation in amplitude is very small when the carrier is recorded on the disc record, so that this system has advantages with regard to the cutting operation and the record’s resistance to wear, if the most appropriate cutting level is determined. Also, as the demodulated output is dependent only on the angular devi- ation, the playback sensitivity of the pickup cartridge has no influence. The next thing to be decided was the carrier frequency to be used. This had to be determined taking into account the upper limit of the frequencies to be transmitted. We thought it reasonable to regard 15 kHz as the upper limit of the audio signal because of the frequency response of the human ear. After making this judgement, it became apparent that the frequency of the carrier to be subject to angular modulation would have to be 30 kHz or more. On the other hand, as the carrier fre- quency was increased, cutting would be- come more difficult and there would be increased interference from the base band signals. After considering these practical factors, we determined 30 kHz to be the most appropriate carrier fre- quency. Therefore, the composition of the CD^l signal is: Base band 0-1 5 kHz Carrier 30 kHz ± 1 5 kHz The required bandwidth is up to 45 kHz, but of this, the section between 15 kHz and 20 kHz is necessary for the filter ir. the playback system which separates the carrier signal from the composite signal This section should not be regarded as part of the signal bandwidth. However, this does not mean that fre- quencies between 15 kHz and 20 kHz are eliminated in recording. If the cut- off characteristic of the filter is good 6 9384 6 CD-4 DEMODULATOR BLOCK DIAGRAM d-4 elektor december 1975 — 1227 enough, components with these fre- juencies can be utilized, t was determined that the carrier level hould be 19 dB lower than the base >and signal. Figure 2 shows the band- vidth structure of the CD^l record. 2D-4 system technology 'D-4 records which cover a wide band- vidth of 45 kHz require correspond- ngly broadly based technology. Stating hese in the order in which they are re- tired: I . Required : Signal superimposition techniques Solutions: - the various parameters of the angular modulated signal were established - wide range modulator was de- veloped !. Required: Cutting techniques Solutions: - the use of 1/2 speed cutting mode was adopted - the cutting stylus was improved 1. Required: Techniques to improve playback sound quality a. pickup of superimposed signals Solutions: - carrier level control recording system - Shibata stylus b. prevention of tracing distortion Solutions: — Neutrex I - Neutrex II c. improvement of S/N and carrier crosstalk distortion Solution: - ANRS d. stabilization of carrier demodu- lation Solution: - PLL demodulation L Required: Related techniques a. reduced cost of playback equip- ment Solution: — Development of IC b. control of phase characteristics of record/playback transducers Solution: - pulse train measuring method c. absolute measurement of record cutting amplitude and crosstalk Solution: - double-beam interference fringe observation These techniques are fully described in Lhe literature given in the bibliography it the end of this paper. Here, we would like to select from these techniques 'hose which affect the tone quality and :ost. The pickup cartridge The pickup cartridge is a very important link in the chain, and is a key item be- cause of its influence on playback sound. It must cover all frequencies up to 45 kHz; apart from this, it must : 1 . not change the playback character- istic from the edge of the record to the center of the record, 2. maintain the same playback charac- teristic when changes occur in the ambient temperature, 3. not damage the record groove. Also, as the carrier must be picked up with as little loss as possible, the radius of the stylus tip must be reduced to about 7 microns. This was why the whole subject of stylus design was re- thought, resulting in the invention of the Shibata stylus. Figure 3 is an en- larged view of this stylus. Almost all the problems which we had considered to be the bottleneck im- posed by the stylus were solved by in- creasing the area of contact of the stylus tip and the walls of the record groove. In 1974, a new, bonded Shibata stylus was developed. In this stylus, titanium is bonded to the diamond tip. This stylus has exactly the same performance as the stylus made of diamond alone; the advantage is that only 1 /20th the amount of diamond is used in the bonded stylus when compared with the diamond stylus. The result is that low cost Shibata styli are now being mass produced. As well as the Shibata stylus, several other kinds of stylus have been developed with the same increased area of contact with the record groove. By putting these into practical use, the world’s leading manufacturers of pickup cartridges have released many cartridges with CD-4 applicability (see ‘Market’ — Ed.). The group delay characteristics of the pickup cartridge and cutter head are physical factors which, unless they are understood, make the handling of FM signals correctly impossible. How- ever, they were unknown until the pulse train measurement method was de- veloped. The fact that the group delay characteristics can now be controlled and optimized when this measurement method is used has greatly contributed to the improvement in CD-4 sound. Neutrex Apart from those kinds of distortion which can be eliminated by improving the performance of the pickup car- tridge, there is a kind of distortion which cannot be eliminated as it is in- herent in the system. For example, tracing distortion resulting from the tracing of the base band inter- feres with the tracing of the carrier, de- grading the sound quality and channel separation however good the perform- ance of the cartridge. To cope with this, Neutrex was developed for use in the CD-4 system. Neutrex I modifies the shapes of the cutting waveform to be complementary to the tracing distortion waveforms. Neutrex II performs reciprocal modu- lation of the component of the carrier which would be modulated by the base band signal because of tracing. The optimum combination of these two Neutrex systems effectively suppresses distortion. ANRS As well as these kinds of distortion there is also crosstalk distortion and tri- angular noise which is a result of the superimposition of the angular modu- lated carrier. The former occurs because of inter- ference between the two carriers. The frequencies at which this distortion occurs are almost pre-determined and the amount of this distortion is directly related to the amount of crosstalk. The latter increases in frequency ranges where the demodulator output is high. To effectively eliminate these kinds of distortion, ANRS (Automatic Noise Re- duction System) was adopted; this functions so as to be frequency selec- tive. Its operation is shown in figure 4. In this context, the adoption of the angular modulation system for the CD-4 system was helpful. Since the demodu- lated output depends on the angular de- viation alone and not the carrier level, ANRS functions correctly regardless of the pickup cartridge used as long as the cartridge picks up the carrier. As ANRS does not modify the base band signal, it has no effect on compatibility. Recent progress in pickup cartridge technology has made possible a separ- ation of 25 dB in the carrier band. Be- cause of this and the effects of ANRS technology, crosstalk distortion has ceased to present a problem. With regard to noise, the improvement of the plating process, the record material and cartridge tracing ability and the use of PLL demodulation has made the noise reduction effect of ANRS more stable and more reliable. For all these reasons, the noise characteristics of CD -4 are now very close to those in 2- channel stereo. PLL The next subject to be introduced is PLL. This is an abbreviation for Phase Locked Loop, as is widely known. It is a feedback system, as shown in figure 5, which produces an output voltage which corresponds to the momentary fre- quency of the input signal. In this sys- tem, frequency trackability is not de- graded by a decrease in the level of the input signal. The PLL circuit used in the demodulator allows the angular modu- lated carrier to be demodulated while maintaining a good S/N ratio. By adopting the PLL demodulation sys- tem, it has become possible to play 1228 — elektor december 1975 cd-4 CD-4 records with no instability, even when the record is worn and the car- tridge has inferior sensitivity. Further- more, the PLL has simplified the de- modulator circuit by making many L and C elements unnecessary. Demodulator ICs As was shown in figure 2, the structure of the signals recorded on the CD^l record is very simple; the fact that the demodulator can also be greatly simpli- fied can be seen from the block diagram in figure 6. Demodulator ICs were developed jointly by Signetics Inc. and JVC and inde- pendently by QSI Inc. in the U.S.A. following this block diagram. Figure 7 is a photograph of the CD4-392 demodu- lator IC developed jointly by Signetics and JVC and figure 8 is a photograph of the CD4 demodulator using this IC. After the results of ICs developed by these companies had been announced, Hitachi completed its development of a demodulator IC; now several manufac- turers are developing IC demodulators. The result of integration of the demodu- lator (including PLL, ANRS and even disc preamp!) into a single IC is that the production cost and size of the CD^t de- modulator component have been greatly reduced. Now, after a great deal of effort, the CD4 system has become much more easily available to users. Conclusion In this paper, the policy and technology of the CD-4 system have been pre- sented. To summarize: 1 . In the CD-4 system the record is of the same quality as the master tape; (A) = (B). 2. The CD^l record is compatible when played back with stereo and mono playback equipment. 3. The CD^l system combines maxi- mum simplicity with minimum cost. Every possible technique was used, from other fields of engineering technology wherever and whenever necessary, to satisfy these basic requirements. As was mentioned at the beginning of this paper, as far as the musicians and record producers are concerned, the 4-channel tape is the result of their art. When software is exchanged between any of the world’s record manufac- turers, it is done using this master tape. The relationship (A) = (B) shows the goal, perfect fidelity in recording and playback. The development of the CD-4 system and CD-4 records has been a quest for this ideal. The fact that this has been combined with items 2 and 3 in one system is one of the strong points of CD-4. Summing it up as briefly as possible: CD-4 has one aim. This is a frontal challenge to achieve a perfect disc record system incorporating the re- cording and playback equipment, which will meet the three requirements stated above. Our basic philosophy, (A) = (B) will, we feel sure, have maximum appeal to both lovers and producers of music. Literature T. Inoue, N. Takahashi, I. Owaki: ‘A Discrete Four-Channel Disc and its Reproducing System (CD-4 System)*; JAES vol. 19, pp. 576-583 (1971) I. Owaki, T. Muraoka, T. Inoue: Further Improvements in the Dis- crete Four-Channel Disc System CD-4 V JAES vol. 20, pp. 361-369 (1972) T. Inoue , N. Shibata, K. Goh: ‘ Techni- cal Requirements and Analysis of Phono Cartridges for Proper Play- back of CD -4 Discrete Four-Channel Records *; JAES vol. 21, pp. 166-171 (1973) T. Inoue, I. Owaki, Y. Ishigaki, K. Goh: ‘ Half-Speed Cutting of the CD-4 Discrete Four-Channel Records and Other Improvements of the System*; JAES vol. 21, pp. 625-629 (1973) W.H. Hoe ft, G. Kelson, N. Takahashi: A Monolythic CD-4 Quadraphonic Demodulator *; Preprint of the Con- vention ofISSCC 1974 Y. Kokubun , S Muromoto, J.M. Eargle: 1 The CD-4 Mark II Modulation Sys- tem*; JAES vol. 22, pp. 416-421 (1974) T. Muraoka, H. Onoye, A. Takayanagi : Measurement of Phonograph Car- tridges by the Pulse-Train Method*; JAES vol. 22, pp. 502-510 (1974) T. Muraoka, N. Takahashi, T. Inoue: ‘ANRS in the CD-4 System*; To be published as the paper of the \ Journal of Audio Engineering So- ciety * k d4-392 elektor december 1975 — 1229 /ictor Company of Japan, Limited (JVC) \s most quadro-enthusiasts will ;now, an integrated demodulator or CD-4 has been available in the etail trade for several months iow: the CD4-392. n this article, we are pleased to tresent all relevant information oncerning pinning and specifi- cations. A practical circuit using his 1C is included. Ve received this article one day lefore going to press, so we are •resenting it as received from JVC - with their original drawings. -Ed. NEW CD-4 MODULATION SYSTEM MARK-11 At present in Japan 8 record companies are releasing CD-4 discs on 3 1 labels and 750 albums are available. People throughout Japan are enjoying a wide variety of music from CD-4 records. In America, 350 CD-4 albums are avail- able on 13 labels including those of the RCA and WEA groups. Recently the A & M label with its excellent repu- tation in the field of popular music has joined the CD-4 family. We expect that the number of CD-4 releases will grow in the future. Supporting this, CD-4 playback equipment has been greatly improved by the enhancement of the fidelity of pickup cartridges and de- modulators. Initially CD^l playback equipment was designed for incorpor- ation in big console-type stereo systems; now the ground-work for its incorpor- ation in component systems has been completed. The introduction of a low- priced demodulator will make avail- able high performance playback equip- ment in a price range which will make it more popular. This has been achieved after a year of cooperation between JVC, the inventor of the CD-4 system, and Signetics, one of the world’s leaders in linear IC technology. The CD4-392 single monolithic IC has two basic functions, a carrier recovery system and an audio processing system. This monolothic IC’s parameters are sufficient to allow the user flexibility in the design of either high performance or minimum cost demodulators. Features of the CD4-392 IC chip The CD4-392 IC chip developed for CD-4 playback offers the following advantages. 1. Versatility and wide range of appli- cations. The CD4-392 IC can be used in a number of design configurations, either low-priced for consumer use or high-priced for professional appli- cations. This IC is designed to be adaptable to newer methods of carrier recovery which may be introduced in the future. 2. Automatic compensation for carrier dropout. CD-4 demodulators using Matrix & audio amplifiers THD (V 0 = 1 .5 V RMS) Equiv. input noise voltage Gain balance Output swing : < .05% : <2 /UV : <0.2 dB : >3 V RMS Carrier recovery system Sensitivity (30 dB quieting 3% deviation) :<200/iV Distortion (± 20% deviation) :<0.2% S/N (V carrier = 20 mV, ±20% dev) : > 70 dB PLL freq. drift with temp :<200 ppm/°C Table 1. Typical CD-4 Demodulator Per- formance (Vcc = 12 V). CD4-392 ICs can be designed to compensate automatically for unex- pected carrier dropout and for other undesirable input conditions. The carrier level is also automatically adjusted for cartridge output levels from 1 to 7 millivolts. 3. High performance combined with economy. The performance of a typical CD -4 demodulator is shown in table 1 . Each block on the integrated circuit is designed for lower noise and distortion and wider dynamic range than previous circuits; this was con- sidered essential if the degree of demodulator performance demanded today was to be realized. Inputs and outputs of the circuit blocks are de- signed to have high and low impedances respectively. Gain, balance and signal level at all block terminals have been established so that external components such as filters and equalizing networks can be simplified. The CD4-392 IC is of the standard 16-pin configuration and two are required for a complete demodulator. The pin configuration is shown in figure 1 . Description of the IC Figure 2 is a block diagram. The single monolithic chip contains two basic systems, a carrier recovery system and an audio processing system. The carrier recovery system consists of a limiter amplifier, a PLL and a synchronous detector. The PLL includes a phase detector, lock range tracer, VCO (volt- age controlled oscillator) and audio amplifier. The audio processing system consists of an automatic noise reduction system (ANRS), its driving amplifier and a matrix circuit. A regulator pro- vides stabilized power for the IC. Lock range characteristics of the PLL are shown in figure 3. These are extremely wide, the DC lock range characteristic being some 20 dB wider than the AC characteristic. Such a wide range is un- usual in PLL systems and allows correct synchronization with any input con- 1230 — elektor december 1975 cd4-392 (TOP VIEW) 1. ANRS-2 INPUT 2. LOCK RANGE CONTROL 3. LOCK RANGE CONTROL 4. LIMITER INPUT 5. LIMITER OUTPUT 6. +VCC 7. SYNC. OET. OUTPUT 8. VCO CAPACITOR 9. GROUND K). FRONT OUTPUT 11. MAIN INPUT 12. REAR OUTPUT 13. ANRS CONTROL U. ANRS-I INPUT 15. ANRS-2 OUTPUT 16. DEMODULATED OUTPUT eo ex ex 15 20 25 30 FREQUENCY (KHz) 35 40 45 LAMP ditions which could occur in practice. Even if a CD-4 disc is inadvertently played back at 45 rpm, the resultant transposed carrier (40.5 kHz) will be within the locking range of the system. Figure 4 shows the quieting curve of the carrier recovery system. There is an S/N of 72 dB from 6 kHz deviation and ANRS improves this ratio by 13 dB for a total S/N of 85 dB, sufficient for professional use. To prevent interference between the carrier recovery system and audio pro- cessing system which are integrated in the single CD4-392 chip, a VCO circuit which oscillates with low current and voltage and which is highly stable was adopted. Figure 5 is a block diagram of this circuit. The VCO consists of a pair of transistors, Q2 and Q3, the emitters of which are common. The transistors switch ON and OFF alternately and repeatedly. The drive circuit connected to a common point between the emit- ters of these transistors operates in response to a control signal controlling current ij , thereby driving the transis- tors. The output is taken from the base potential of transistor Q3. This base potential varies between two values in accordance with the alternating ON and OFF states of the pair of transistors. The charging and discharging currents :d4-392 elektor december 1975 — 1231 Figure 1. Pin configuration of CD4-392 1C. Figure 2. Block diagram of demodulator using the CD4-392 1C. Figure 3. AC/DC lock range characteristics of ’LL. Figure 4. Quieting curve of carrier recovery iystem. Figure 5. Schematic diagram of voltage con- trolled oscillator. -vhich are of the same value are applied o a capacitor connected to the base of ransistor Q2. The pair of transistors are twitched ON and OFF in response to lie charging and discharging of the ;apacitor via the current mirror circuits ;onnecting the pair of transistors. The ;ontrol signal controls the oscillation requency of the linear voltage con- rolled oscillator. Circuits external to the 1C riie 1C has been designed so that a lumber of approaches to ancillary cir- ;uitry can be accommodated. 1. Band-pass and low-pass filtering. Either passive LCR or active RC devices can be used. 2. RIAA Equalization Normally, the RIAA circuit is div- ided into two sections, LF boost and HF roll-of, with the feed to the carrier recovery system taken before the HF roll-off. However, as the carrier can be limited by up to 50 dB in this IC, it is possible to take the feed to the carrier recovery system after the RIAA equalizer. 3. ANRS System ANRS is an encode-decode noise reduction system and the parameters in playback must match those used in the recording system. Splitting the ANRS into two blocks facilitates a number of approaches to the tailor- ing of the dynamic characteristics of the ANRS to those desired. 1. Carrier loss compensation Carrier loss can result from self- erasure of the HF signal during the cutting process as well as from the CD4-392 DEMODULATOR IC CHIP PATTERN 14 0 516 910 1232 — elektor december 1975 cd4-392 :d4-392 elektor december 1975 — 1233 presence of significant harmonic interference from the base band during playback. The synchronous detector compares the phases of the VCO and the input carrier signal, producing an output signal which accords with the condition of the input signal. This output signal is fed to the lock range controller which controls the PLL so that it is set to the condition most suitable to the input signal, giving a demodulated output which is compensated for carrier loss. Figure 6 is a block diagram of a standard circuit using the CD4-392. History of the CD-4 demodulator Figure 7 is a photograph showing the progress made in CD-4 demodulators. Figure 6. Block diagram of CD4 demodulator. Figure 7. From left to right: the CD4-1, the 4DD-5 and the TDM-18A. Figure 8. A complete CD-4 demodulator using the CD4-392 (one channel shown). The CD4-1 is the first generation demodulator used when the CD-4 system was first released, the 4DD-5 is the second generation demodulator using the PLL IC and the TDM-18A is the third generation demodulator which incorporates the CD4-392 IC. Figure 8 is a schematic diagram of the standard TDM-1 9A. The TDM-1 8A and the TDM-1 9A represent a significant step forward in the design and evolution of CD-4 playback hardware with their performance characteristics having been substantially improved. |< 1234 — elektor december 1975 santatronic M. Keul SI R 1 Ci R? I in R3 LED •— © N — ' 9105 The circuit shows a transistorised uni- versal thyristor, or TUT. It operates as follows: When SI is open, the LED will light when the supply voltage is turned on, because T1 and T2 are both turned off. If SI is now closed, T1 receives a base current, so that this transistor turns on and T2 is driven into saturation. The voltage drop across the emitter- collector junction of T2 and across the base-emitter junction of 11 will be lower than the voltage drop across the LED, so that the LED will extinguish. The thyristor is now on, and re-opening of S 1 makes no difference. Only a very brief interruption of the supply voltage can extinguish the thyristor causing the LED to light again. This TUT circuit can be useful, for instance, as a mains failure indicator for a digital clock. O-* 0105 P. Engelmann Electronic candle & i 4.5V =4^ This candle is a simple electronic toy. It can also be used for conjuring tricks. The circuit is designed around an incan- descent lamp which can be ignited with a match and can be blown out. Figure 1 shows the circuit and figure 2 the mechanical construction. The LDR is mounted in the side wall at lamp level. The side wall is covered with a translucent material to hide the LDR. If a burning match is held close to the lamp, the LDR is illuminated and the lamp lights. From now on the LDR is illuminated by the lamp, so the lamp continues to burn. If we blow against the lamp, so that it swings away from the LDR, the lamp is extinguished. The sensitivity can be adjusted with the 100 k preset potentiometer. For the prototype the darlington transistor MPS A 14 was used. Owing to the high current gain of this transistor the circuit is very simple. Of course, a darlington made from discrete transistors will also do. The box below the ‘LDR-wall’ accom- modates the electronics as well as the battery. The bottom of this box is fitted with a pin which operates a spring con- tact when the box is placed on the table. This spring contact is the battery switch. preamp for counter elektor december 1975 — 1235 preamp for counter In the last issue of Elektor the basic circuit of a frequency counter was described. In this issue a preamplifier to increase the input sensitivity is discussed. The input sensitivity of the basic fre- quency meter is inadequate for use in most audio and r.f. circuits where signal levels of at the most a few hundred millivolts are likely to be encountered. It was felt that an input sensitivity of around 10 mV RMS would prove suit- able for most applications, and it was with this in mind that the parameters of the preamplifier were determined. The circuit must meet the following specifications: 1. Bandwidth: Greater than the band- width of the frequency counter (D.C. - 18 MHz) 2. Gain: 40 dB 3. High input impedance and low input capacitance. In order to keep the circuit reasonably simple and stable the first criterion was relaxed slightly and the gain rolls off at high frequencies. The input resistance must be kept high to avoid loading the circuit to which the frequency counter is connected, and the input shunt capacitance must be kept low. In practice an input resistance of 1 M£2 should be adequate for most applications. The input capacitance depends to a large extent on the capaci- tance of the input leads and the wiring within the instrument. The complete circuit of the preamplifier is given in figure 1. It consists of three differential stages and an output emit- ter-follower. The input stage is a differ- ential source-follower. This configur- ation was chosen because the input FET offers a high input resistance in this mode, and in fact the input resistance of the preamp is almost entirely deter- mined by the 1 M resistor R4. The E420 dual FET has a very low gate-source capacitance and, since both FET’s are grown on a single chip, exceptionally good matching and temperature tracking. A further advantage of this type of input stage is that its overload margin is virtually full supply voltage. However the maximum input voltage is clamped to 0.7 V by D1 and D2, whilst R5 limits the current through these diodes. Cl provides some compensation »*•*•*••• • • • • ••••••••• • »• •••••• • • • • * • • ••••••• • * * » • • • • • • • • • • • • • • »•••••* •••• • ••••• ••••■- I • • • • !•••** for high-frequency roll-off by shunting R5 at high frequencies. As the pre- amplifier must operate down to D.C. it is D.C. coupled throughout. However, so that A.C. signals with a large D.C. component superimposed can be measured without the D.C. blocking the amplifier Cx is included in series with the input. For voltages in excess of 1.5 V p-p, especially from high impedance sources, it is recommended that an input attenu- ator be used to avoid D1 and D2 loading the signal source when they conduct. D1 and D2 could also be replaced by zeners to increase the maximum input voltage, but if an attenuator is fitted this should not be necessary. The input stage is followed by two differential stages using bipolar transis- tors (T2/T3 and T5/T6). The output is taken from the emitter-follower T8. As the frequency counter will not operate correctly with insufficient signal an indication of the output level of the amplifier is desirable. This is provided by T7 and D5. As the signal approaches a level sufficient to drive the counter the signal at the emitter of T7 will also increase and D5 will start to light up. By the time the ampli- fier is limiting D5 will be glowing quite brightly. The frequency response of the amplifier is compensated by C3. Up to about 5 MHz an input signal of 50 mV will cause limiting. Above this the gain rolls off until at 18 MHz 100 mV is required for limiting, although 60 mV. will produce sufficient output to drive the frequency counter (see figure 2). When the amplifier is not connected to the frequency counter the output will swing between positive and negative supply (less base-emitter voltage of T8) in the limiting condition. However, as soon as the frequency counter is con- nected to the output of the preamp the clamping diodes (D2 and D3) in the TTL input circuitry of the counter will limit the maximum negative excursion to about -0.6 V. This output con- 1236 — elektor december 1975 preamp for count* Figure 1. Complete circuit of the input pre- amplifier. Figure 2. The frequency response of the preamplifier is flat up to about 5 MHz, and above this frequency it rolls off. The lower curve shows the input voltage required for reliable operation of the counter, whilst the upper curve shows input voltage for full limiting of the amplifier. Figure 3. Voltage doubler and stabilizer to provide the negative supply required for the preamplifier. This is derived from the existing transformer that provides the +5 V supply for the counter. Figure 4. Printed circuit board and compo- nent layout for the preamplifier. Photo 1. The completed preamp board. figuration has the advantage that the logic ‘O’ level is well defined. Power Supply The frequency counter requires a single +5 V power supply. The input pre- amplifier, however, needs a negative supply also. Fortunately, since the current drawn from the negative supply is small it can be derived by a voltage doubler type of arrangement with a simple stabilizer as shown in figure 3. This derives its A.C. input from the existing transformer for the positive supply to the counter. Construction Printed circuit boards for the pre- amplifier and the associated power supply are given in figures 4 and 5 respectively. The only point to note when building the boards is the connec- Parts list for figure 1. resistors: R1,R4 = 1 M R2,R3,R5,R14,R16 = 1 k R6 = 4k7 R7,R 1 2,R13 = 150 12 R8,R9 = 10 12 RIO = 39 12 R11 = 100 12 R1 5 = 22 12 R1 7 = 470 12 PI = 47 12, preset capacitors: Cl = 33 p C2= 10 n C3 = 1 n5 C4.C5 = 82 n semiconductors: T1 = E 420 T2,T3,T4 = BF 199 T5,T6 = BC 557 B T7,T8 = BC 547 B D1 to D4 = DUS (e.g. 1N4148) D5 = LED Parts list for figure 3. R1 = 1 k Cl ,C2 = 1000 n, 16 V C3 = 220 fl, 10 V T1 = BC 161 D1,D2 = 1N4002 D3 = zener 5,6 V, 400 mW ireamp for counter elektor december 1975 — 1237 3 1 BC161 I I 1 9031-3 ion to the E420 dual-FET as this device nay be unfamiliar. The pin configur- ition of this device is given adjacent to he circuit diagram in figure 1. Because of the high frequencies involved dose attention must be given to the ayout of the complete frequency counter. A suitable layout is given in 1 igure 6 and it is recommended that this .hould be adhered to. In particular it .hould be noted that all earth con- lections go directly to chassis and that eparate +5 V supply leads are taken rom the power supply board to the pre- amplifier, the control logic and to the display/counter assembly. The input and output connections to the input preamplifier must be made using screened cable but, in contrast to normal a.f. amplifier practice the braiding of these cables should be earthed at both ends. Preliminary Adjustments The only adjustment necessary to the input preamplifier is made with the 1238 — elektor december 1975 preamp for counter offset control PI. This adjusts the D.C. balance of the second differential stage, and hence the quiescent D.C. output level. In a normal instrumentation amplifier this would be set at 0 V so that clipping would be symmetrical (i.e. limiting would occur at the same level for both the positive and negative half cycles of an A.C. input waveform). In this case, however, the amplifier is providing an asymmetric output wave- form (i.e. TTL logic ‘0’ to logic ‘1’). The optimum setting for the quiescent out- put level is just below the TTL guaranteed logic ‘O’ level of 0.8 V. The amplifier is then in its most sensitive state as a small positive swing will take the output above the TTL ‘T threshold. In practice, to be on the safe side and allow a small noise margin it is best to set the quiescent output level at about 0.5 V. This should, of course, be done with the amplifier input grounded. Applications of the frequency counter With the addition of the input pre- amplifier the uses to which the fre- quency counter may be put are ex- tended considerably. A few of the appli- cations are: a.f. circuits — monitoring of oscillator frequency when doing frequency re- sponse measurements on amplifiers, filters etc. r.f. circuits — monitoring output fre- quency of r.f. oscillators, frequency to which receivers are tuned etc. PLL circuits — setting up of VCO free- running frequency, particularly useful for stereo decoders. time measurements — in the period mode the counter can be used to measure time intervals, and may thus be used as a stopwatch. Of course, the frequency counter does have its limitations. At low frequencies the input resistance of 1 M determines the type of circuits to which it can be connected. It should not, for example be connected to points in a circuit where the output impedance is greater than a few tens of kilohms, as it may load the circuit under test. At high fre- quencies the capacitance of the (screened) input leads will have an effect, and if connected to the fre- quency-determining section of an oscil- lator, for instance, it can easily alter the frequency, thus giving a false reading. Due care should therefore be taken to connect the counter to the lowest impedance point in the circuit, and away from frequency-determining cir- cuits such as tuned LC or RC circuits, RC networks in filters etc. The second limitation concerns the timebase facilities of the counter. As the timebase is derived from the mains this limits the accuracy, but this is adequate for most applications. The three gate periods of 10 ms, 100 ms and 1 s give maximum full-scale readings of 99.9999 MHz (not attainable because of upper frequency limit of TTL 18 MHz), 9.99999 MHz and 999.999 kHz. With Figure 5. Printed circuit board for the negative supply. As this is the same length as the preamp board the two can easily be mounted side-by-side. Figure 6. Wiring diagram for the complete frequency counter including the input pre- amp. It is advisable to adhere strictly to this layout for trouble-free operation. Photo 2. The —5 V supply can be mounted on top of the +5 V supply to save space. If, however, space is not a problem side-by-side mounting makes for easier servicing. the gate period set to 1 second the resolution at frequencies below 100 Hz is worse than 1%. At very low fre- quencies it is probably best to use the counter in the period mode and measure the period of the signal rather than the frequency. In the period mode the full- scale reading is 9,999.99 seconds so a 1 Hz input can have its period measured with a resolution of 1%. This still leaves a ‘hole in the middle’ for frequencies between 1 Hz and 100 Hz where the resolution in either the fre- quency or period mode is worse than 1%. If this is felt to be a problem then the addition of a crystal timebase is worth considering, and a suitable choice is the universal frequency reference published in Elektor No. 5. This has outputs at frequencies from 1 MHz to 1 Hz, and in addition to providing greater accuracy it enables the resol- ution in the period mode to be greatly increased. For instance, if the 1 MHz output is used to time a 1 Hz signal, then the period can be measured with a resolution of 0.0001% (in fact the counter will just overrange). Addition of a crystal timebase to the frequency counter will be discussed in a future issue. M reamp for counter elektor december 1975 — 1239 1240 — elektor december 1975 730-740 Using only two ICs, the TCA730 and TCA740, a complete stereo control amplifier can be built. An exceptional feature is that the volume, balance, and tone are all DC controlled. Specifications: Frequency response (±1 dB): 20-20 ,000 Hz Signal-to-noise ratio: 57 dB Channel separation: 60 dB Distortion: 0.1% Input overload level : 1 V Input impedance: 250 k Max. output level: 1 V Output impedance: 4.7 k Volume control range: +20 - -70 dB Max. bass lift/cut: 15 dB Max. treble lift/cut: 15 dB An alternative to the conventional potentiometer control of volume, bal- ance and tone should be more than wel- come. The problems connected with running audio signals over potentio- meters are well-known: the long cables almost inevitably lead to hum and cross- talk, and ‘crackly’ potentiometers are notorious . . . A solution is offered in the circuit de- scribed in this article. The Philips inte- grated circuits TCA730 and TCA740 have built-in ‘potentiometers’ that can be DC-controlled. The TCA730 contains the electronic volume and balance con- trols, whereas the TCA740 can be used for bass and treble control. The necessary control voltages can be derived from a simple voltage-divider circuit incorporating several potentio- meters. Since these potentiometers don’t have to carry the audio signal, they can be connected to the circuit via virtually any length of cable: the hum pick-up would have to be very severe be- fore it could cause audible modulation of the audio signal. Another advantage of these ICs is that one (mono) potentiometer can be used to control several audio channels simul- taneously, with a minimum of imbal- ance between the channels. The circuit described here can be used in combination with almost any pre- amplifier, and the performance is definitely ‘Hi-Fi’. TCA730: volume and balance The complete stereo control amplifier is shown in figure 1. For convenience, the circuit has been cut in two: fig- ure la shows the TCA730 with associ- ated components and figure 1 b shows the TCA740. The output of la (con- nections A and B) is connected to the input of 1 b. The first section (TCA730) is the vol- ume and balance control. In the maxi- mum setting of the volume control the gain is x 10: 100 mV in gives 1 V out. This input sensitivity, in combination with the input impedance (250 k), means that the amplifier can be driven direct from practically any receiver, tape recorder or crystal pickup. Micro- phones or dynamic pickup cartridges will need a separate preamplifier, of course. The circuit shows the complete stereo version, so that most of the components come in pairs. For instance, R1 (270 k) is the input resistor for the left channel and R 1 ' is its twin in the right channel. The (DC) control voltages for volume and balance control are connected to pins 13 and 12, respectively; both voltages should be linearly adjustabh from IV to 9 V. Figure 2 gives tht circuit of the control potentiometers: four linear potentiometers (10 k) are connected in parallel in a voltage divider circuit that gives the correct control range. Two of these potentiometers are used for volume and balance control. Needless to say, the supply voltage for this control circuit must be well stabil- ised and smoothed. If the audio input voltage is 100 mV or less, the volume control has a range from +20 to —70 dB. The balance con- trol range depends in part on the vol- ume setting: when the volume is set at —20 dB or less, the balance control range is +10 to -10 dB, but this range is reduced at higher volume settings. An interesting option is offered at pin 4. If this pin is simply connected to supply common through an electrolytic (Cl 1 , 470 n ), the volume control works as normal. However, if a 1 k resistor is connected in parallel with this electro- lytic (R15, dotted in figure la) a physiological volume control is ob- tained. It seems unncessary to go into this in any further detail — the effect, also known as ‘contour control’, is well- known by now. Suffice it to say that in the maximum setting of the volume control (control voltage: 9 V) the fre- quency response is flat, whereas at a much lower setting (control voltage: 3.2 V) the bass is only 40 dB down and the frequency range from 200 to 7000 Hz is already 70 dB down. 30-740 elektor december 1975 — 1241 TCA 740 12 4 PI. . . P4 =10K- T 22n R8 01 03 R7 02 R6 I D4 12 ■o o-*a — L I 8 11 R5 ABC BD| N |q 2 a in 7490 r 0I1)(2) r 9I1M2) © GND 10 _ _ J 9169 M.G. Fishel Table. throw 7490 count 9=1001 0=0000 1=0001 2=0010 3=0011 4=0100 D 1 on off on off on off D4,D5 off on on on on on D2.D3 off on on on • • D6,D7 off on display In this simple electronic die only two ICs are used for the oscillator, 6-counter, decoder and display-drivers. Four basic display patterns are used to create the six displays required for a die (see table). The LEDs D1-D7 are driven from a simple decoder circuit consisting of 4 inverters (2/3 IC1), which in turn is driven by a 7490 (IC2) wired as 6-counter. A slightly different circuit from the usual is used to achieve the count of six: the A and C outputs are connected to the reset-9 inputs, so that as soon as a count of five (0101) is reached the 7490 is reset to 9. It now counts 9-0-1 -2 -3 -4-9-0-..., a total of six I i •V-rV m " m mjp i HHr ■u-'i'Z'.'kW., V , HP mm )ik wMfMj, wmtii m wi s f jNHHV -P ^0Stwmw>' V, 'i,, ;.; / , : v '' L .'- V • VAy , - , 7,*Y ( v • , i /, ■ / yj ' v.'.v > 1 - - santatronics elektor december 1975 — 1245 counts per cycle. The 7405 contains 6 inverters, of which only four are used for the decoding circuit. The remaining two can therefore be used to construct a simple multivi- brator for driving the counter. The multivibrator is free-running; it can be connected to the counter via push- button S2 for each throw of the die. The LEDs ( D 1 - D7) are arranged on the printed circuit board in the correct pattern. They can be mounted either on the component side or on the copper side; this second alternative will prob- ably prove the most practical when mounting the die in a small box. Parts list: Resistors: R 1 - R4 = 1 k R5 - R7 = 120 £2 R8 = 220 £2 Capacitors: C1\C3,C4 = 22 n C2* = 22 /i/6 V Semiconductors: IC1 = 7405 IC2 = 7490 D1 - D7 = LED • — = see text Capacitor Cl (dotted in the circuit) is not mounted on the p.c.b., because it did not prove essential in practice. However, a good power supply (prefer- ably stabilized) is advisable. If batteries are used Cl should be mounted on the board in place of C2, the leads from the battery must be short, and a larger electrolytic (220 /i - 470 ju) must be mounted across the supply connections to the board. { Pl 6 106 & —cm ■ n i T si 9 169 A M I T. Meyrick door- chime driver The bell-wiring-system in many blocks of flats consists of a set of pushes in the communal entrance hallway, a centrally-located (and inaccessible) transformer and a cheap-and-nasty trembler bell in each flat. One's own front door is then usually fitted with a kind of king-size bicycle-bell that requires considerable force to be applied to the push. Since the whole set-up is a minimum-price job, the avail- able current is invariably too low to operate a full-length door- chime, particularly when the flat concerned is located on a higher floor. It is also not normally possible to connect a bell-push on the flat-door into the system. The circuit described here is the result of one engineer's taking up the gauntlet . . . Figure 1, The original situation. The trans- former and the bell-pushes (a, b, c, d . . .) are installed in the communal entrance hall. The existing bell-system is typically wired as in figure 1. The transformer and the bell-pushes are installed in the main hall (or at the street-door), with only the ringing-lines and the common return being brought upstairs. In par- ticular the ‘hot side’ of the transformer secondary is therefore inaccessible, except at the actual pushes. The sol- ution is obvious: shunt a rectifier diode across the bell-push contacts, so that santa tf<2S 1246 — elektor december 1975 door-chime driver ©- 0 - T I Figure 2. Circuit diagram of the complete door-chime driver. In the entrance-hall one only needs to mount D1 across the contacts of the bell-push correspond- ing to one's own flat; the remainder of the elec- tronics is installed upstairs near the chime. S2 is an extra push that can be in- stalled at the flat entrance, to replace the usual mech- anical 'bicycle bell'. one always has at least half-periods of the AC to play with upstairs. (Figure 2). The upstairs installation starts out with a bridge rectifier (D2 ... D5) and a large electrolytic capacitor (Cl). This reser- voir is charged, under standby con- ditions, almost to the peak value of the unloaded AC — typically to about 12 volts. The actual driver circuit is a monostable trigger that responds to the arrival of positive half-periods. These come in either through D6, when the downstairs push shorts Dl, or through D7 and S2 (installed at the flat door). If Dl is connected in the opposite sense across the push SI (it may not always be obvious which wire is which), the positive half-waves will come in continu- ously through D6. This is no cause for alarm (or for another trip downstairs) - one simply interchanges the incoming wires, so that D6 is connected to the common return and D7 to the ringing- line from SI . The zenerdiode provides a high thres- hold at the input, so that interference pulses or the voltage drop in the common return (when someone elses bell is ringing) do not give a false signal. The positive wave peaks pass through R2 to charge C2 and C3. When T1 is driven into conduction it will also turn on T2. Positive feedback through C4 and R3 will now turn T1 on further, so that the circuit quickly saturates. This applies power to the chime solenoid . . . ‘Ding’. C4 will now charge up through R3, pro- viding a base current for T1 that will keep the circuit temporarily in satu- ration. The current through R2 on its own is not sufficient to do this, even when the push is held down. As C4 builds up a steadily higher voltage, the current through R3 will drop, until the point is reached at which T1 and T2 start to come out of saturation. The voltage at T2 collector now shifts slightly negative, causing a negative- going drive to be applied to T1 through R3 and C4. This further reduces the drive to T2 ... so that the circuit rapidly turns off. ‘Dong’. The base of T1 has now been driven far negative, so that the circuit is blocked for several seconds - until C4 can dis- charge sufficiently through R3, R2 and Rl. If one of the bell-pushes is held down the switch will re-trigger after this interval, so that the chime will play slowly but continuously: ‘Ding . . . dong . . . ding . . . dong’. The chime solenoid should be wound for 12-volt operation. If a 6-volt type is used the current surge will cause a far too violent ‘ding’ (the reservoir voltage being a given condition) - and T2 may be destroyed. The prototype circuit actually did use a 6-volt chime (Fried- land) which happened to be on hand - but it was rewound for 12-volt oper- ation. Inspection showed that the solenoid was fully wound with 0.3 mm diameter enamelled-copper wire (SWG 32) to a total resistance of 6 £2. This winding was stripped and replaced by a full winding of wire-diameter 0.22 mm (SWG 36). The new solenoid had 30 £2 resistance and about the right number of ‘ampere-turns’. A second choice solution would be to use a 6 £2 wirewound resistor (about 5 W) in series with the original solenoid - but this would call for a 25000 (jlF reservoir capacitor! Note that D9, across the solenoid winding, prevents the rise of a back- voltage at turn-off which could (would) destroy T2. There is, incidentally, no reason why sc rap -box ‘germanium’ transistors (such as AC 127/AC 128) should not be used - the circuit is fairly uncritical. ;ontents volume 1 elektor december 1975 — 1247 AF page amplifier for 1 .5 volt supply 717 amplifier, 6.5 watt 1C 749 austereo control amplifier 729 austereo disc preamplifier 730 austereo power supply 732 austereo 3 watt amplifier 730 austereo 15-30 watt amplifier 733 CD-4 1224 CD4-392 1229 compressor 440 disc preamp 444 edwin amplifier 910 electronic loudspeaker 50 elektor 'sonant' loudspeaker system 230 equa amplifier 16 from din to equa-standards 8 loudspeaker diagnosis 53 miniature amplifier 917 mixer, FET- 713 motional feedback 220 output power nomogram 22 philips sonant 216 quadi-complimentary 1220 quadro 1 - 2 - 3 - 4 ... or nothing? 33 quadro in practice 646 sonant 215 tap preamp (part 1 ) 410 tap preamp (part 2) 624 tap-power 1130 volume control, fingertip 750 volume, voltage-controlled 750 730/740 (control amplifier) 1240 Amusement, games beetle 638 bicycle trafficator 944 big ben 95 239 brake lights for model cars 1143 chestnut oven 1223 die 1244 direction pointer 746 electronic candle 23 electronic candle 1234 fido 418 flickering flame 49 humming kettle 1129 lep 942 level crossing protection 1215 lie detector 1143 marine diesel 427 the moth 644 photofinish 1214 roll out the bandit 934 siren, 7400 723 sniff race control 660 steam train 56 steam whistle 57 three-eyed bandit 238 tv tennis 1111 volume 1 Cars page afterburner 715 alarm circuit 714 car lights failure detector 716 digital rev counter 12 interference suppression in cars 616 service meter, auto 718 supplies for cars 632 thief suppression in cars 618 traffic news indicator 724 Digital clock suppression 721 cos/mos digital ic's 241 decimal to bed converter 417 divide by 1 to 10 23 divider, programmable frequency- 748 division by 2, 5 and 10 740 division by 3 740 division by 4 740 division by 6 740 division by 7 740 division by 8 740 division by 9 741 EXOR 711 EXOR 724 interference rejector 723 mos tap 242 NAND-NOR gate 728 NAND-OR gate 746 NOR-ANDgate 713 pulse multiplier 725 reset for TTL, automatic 722 tap sensor 43 trigger, 555 as a 716 tri-stable 933 the 74121 742 use of 7493 instead of 7490 741 30 MHz amplifier 945 Displays contrast adapter 752 decimal to 7-segment converter 710 display input selector 734 LED display dimmer 736 led displays 451 improved 7 segment display 258 improving readability of displays 221 maxi display 251 seven segment abc 718 universal display 223 General elektor services to readers 422, 1222 elektor shorthand 660 how to gyrate - and why 255 1C lists 753 ota 927 tup - tun - dug - dus 9 tup-tun list 459,615,756,1255 1248 — elektor december 1975 contents volume page turning off thyristors 1217 what is cybernetics? 637 with a pencil point 1153 HF FM detector with ceramic filter 714 modulation systems 246 modulation systems (part 2) 454 mono/stereo indicator 720 pll systems 412 receiver, sensitive coil-less 719 selective amplifier 720 stereo decoder, CA 3090 AQ 747 traffic news indicator 724 tunable aerial amplifier 38 tv sound 234 tv-sound channel, 1C 743 ultrasonic detector 739 universal ota pll 1144 Lights disco lights 924 economic flashlight 739 'find me' flashlight attachment 745 flasher, triac 712 fluorescent lamp dimmer 738 psychedelic lights 259 running light 717 Measuring a/d converter 448 capacitance meter 751 conductance tester, mini siren 745 curve tracer 722 dil led probe 227 distortion meter 29 dual slope dvm 936 FET 12 adapter 737 four-channel switch 738 frequency counter 1121 frequency counter, preamp for 1235 frequency meter, analogue 737 frequency reference, universal 715 generator, diac sawtooth- 750 generator, staircase 713 h-l logic tester 623 led-level 460 micro-squeaker 645 noise generator 427 oscillator, 1 kHz 744 overmodulation indicator 725 recip-riaa 252 signal squirt, mini 751 tester 716 tester, p.c. board and wiring 735 three-tracer 1218 tup -tun -tester 608 tv test pattern generator 1149 volume 1 Musical instruments page key contacts for organs, gated 734 microdrum 222 minidrum 208,428 mostronome 731 phasing 922 rhythm generator M 252 726 rhythm generator M 253 727 rhythm generator power supplies 728 string sound 622 tremolo, LED controlled 748 Power supplies austereo power supply 732 battery stand-by circuit 736 heatsink shrinker 746 rhythm generator power supplies 728 slow-rise power supply 745 supplies for cars 632 voltage regulator, wide range 712 voltage stabilizer as current source 717 +/0/— from one winding 724 Sundries active flash slave 1129 arithmetic teacher 736 cuff links 728 dark room exposure timer 749 diode, ideal 719 doorchime driver 1245 economic relay 737 filter, active low-pass 752 fish feeder 931 flasher, LED 744 flasher, LED, for higher voltages 744 frequency dependent resistor 712 fuse destroyer 747 hum receiver 735 inductance, variable 743 light dimmer 935 opto-coupler for mains sync 728 oven, mini- 720 speed controller, triac 735 swinging inductor 12 temperature monitor 742 thermometer 721 time machine 423 tut 1234 Timekeeping calculator doubles as stopwatch 732 calendar 1210 car clock 946 clamant clock (1) 1134 clamant clock (2) 1249 mos clock 5314 24,634 seven segment for MOS-clocks, improved . . 748 time signal simulator 741 timer, kitchen 711 versatile digital clock 918 ilamant clock (2) elektor december 1975 — 1249 Missing link Clamant clock part 1 There is an error in the circuit shown in figure 12 (p. 1 139). The 'O' position of S3 should not go to supply common. Instead, it is connected through a 330 £2 resistor to supply common and through two diodes to the A8 and A9 outputs of the 7473. The anodes of the diodes are connected to A8 and A9 and the cathodes of both are connected to the switch contact. A8 A9 o— @ 4Q15A [Z] nf clock In the last issue of Elektor various sound effects which could be added to electronic digital clocks were described, including a 'tick', alarm systems and a time signal simulator. In the second part of the article various chiming and striking systems are discussed. In a conventional chiming clock there are two systems. A chime, which plays a tune just before the hour, and a striking system, which sounds a bell a number of times equal to the number of hours. In more sophisticated clocks the chime may also play a portion of its tune at the quarter-, half- and three- quarter-hour marks. In simpler clocks the chime may be absent altogether. It is difficult to convincingly imitate bells and chimes electronically, so in this article two types of system are described, a fully electronic system driving a loudspeaker, and a hybrid electromechanical system suitable for driving a normal electric door chime. The circuit of a simple electronic chime is given in figure 17. It operates as follows: Figure 17. Circuit of a simple electronic chime. every hour the tens of minutes counter in the clock produces a negative-going pulse that changes the state of the hours counter, and hence the hour display. In the circuit of figure 17 this is used to trigger a monostable with a period of about 4 seconds. The Q output of this monostable is connected to one of the reset inputs of a 7493 divide-by- 16 counter, so that when the Q output of the monostable goes low the counter is enabled and counts pulses from the clock seconds counter, which are fed into the A input. T1 and T3 form a voltage-controlled oscillator, and as the output states of the 7493 change so does the voltage applied to the base of Tl, thus altering the frequency of the oscillator. The oscillator will thus produce a se- quence of notes until the monostable resets, and when the seconds pulse input (which is also connected to the other reset input of the 7493) goes high then the counter will reset. T2, which is driven by the Q outputs of the mono- stable, switches the power supply to the 1250 — elektor december 1975 clamant clock (2) is output C from 10 min. counter 0 ° 0 ° hours reset see text A i 14 BDin a A, | (C1 = B 7492 R om c r 0(2) D 12 II B 14 BDim A a in |C2 = B 7492 R om C r 0(2) D 12 11 A1 A2 IC3 = 74121 c R/c 10 11 9 cr ni_ 100 n 6 V 1.2 s pi 47k li )£> -| N6 N7 11 4 ^. N5 7486 7401 13 12 Q Q V 2 IC4 = clear clock 1/2 7473 10 13 I 4015 18 5V -© C2 from _ 3 Sout from S'out o — - I — Dl — i 100/J 10 6 V 11 14 •- PI R1 47k A1 A2 R/C + V b R 74121 GND Dl 1 10/J 16V_ ♦ — (+: ? DUS I 'tun -DUS D2 4 — 4015- 5V I «8V — • — 100o 1 Gong J Figure 18. A striking system suitable for driving an electric bell or chime. PI must be adjusted to give a monostable period time greater than 1 second but less then 2 seconds. Figure 19. By gating the 3 output of the ten- minute counter and using it to drive the chime the clock will strike on the half-hours as well as the hours. Figure 20. A suitable drive circuit to switch the chime. There are two inputs, one from the striking circuit and one from the half hour gating of figure 19. Figure 21. If the drive circuit is used with a D.C. bell then the relay may be omitted and an additional transistor connected as shown will switch the bell. Figure 22. Circuit of a complete striking sys- tem. PI must be adjusted to give a mono- stable period time greater than 2 but less than 4 seconds. VCO, thus disabling it when the chime has finished. PI can be used to vary the length of the monostable pulse and hence the number of notes in the chime sequence. Altering C2 will change the frequency range of the VCO — the larger C2 the lower the frequency. As a final point, if a faster chime rate is required then the 7493 may be driven by 10 Hz pulses instead of 1 Hz pulses. Striking the hours A circuit for striking the hours is shown in figure 18. The basic idea is that the output of the hours counter is com- pared with the output of a second coun- ter which is driven by 1 Hz pulses. Every hour on the hour 1 Hz pulses are gated into this counter until its count equals the output count of the hours counter. The number of 1 Hz pulses required to achieve this state is thus equal to the number of hours and these pulses may be used to drive a chime or bell. The circuit operates in the following manner: instead of using the hours counter in the clock to provide the re- "19 from 10 min. counter ao b out bo 4015-19 R 1 1 f 05V bell T 1 1N4001 3 ) TUN T2 omit D2 139 4015-21 quired information an auxiliary divide- by-1 2 counter (IC1) is used. This has the advantage that (coded) outputs from 0 - 1 1 are available directly from a single counter, whereas deriving these outputs from the hour and ten-hour counters in the clock would require additional gating. It should be noted that, whereas the clock counts hours 1-12 the counter counts 0-1 1. This is no disadvantage as there are still 12 output states for the striking system. Every hour on the hour monostable IC3 is triggered by the output of the ten- minute counter. The Q output of this mcnostable is used to clear flip-flop IC4, thus allowing 1 Hz pulses from the 10-second counter through Nl. IC2 now counts the 1 Hz pulses. Exclusive-OR gates N2 - N5 are used to compare each bit of the hours count with each bit of the output of IC2. When each bit is equal the outputs of N2 - N5 are all low so the commoned outputs of N6 - N9 go high. On the next pulse to IC2 the outputs of the two counters become different and the commoned outputs of N6 - N9 go low again, thus clocking IC4. :lamant clock (2) elektor december 1975 — 1251 Output C from 10 min counter 05V 0«8V depending on relay fromlOminute counter STOP Bo 1 ^. 1/3 IC1 =7410 C R/C ■ A1 IC8 = A2 7412 Gl k R Q 1 VID Q 7 4015 22 Hie Q output of IC4 goes high, resetting C2, while the Q output goes low, docking N1 so that no more 1 Hz Dulses can be counted. The number of lulses allowed through N1 is thus equal :o the count of 1C1 plus 1, which is of ;ourse the number of hours since IC1 is ilways one digit behind the counters in he clock. The pulses can therefore be taken from the output of N 1 and be jsed to drive the chime or bell, ro ensure that counter IC1 is in synchronism with the clock hours coun- :ers, and thus prevent the wrong hour from being struck, it is necessary to r eset 1C1 to zero at the change from 12 :o 1 o' clock in a 1 2 hour system, or at he transition from 12 to 13 hours or 00 :o 01 hours when used with a 24-hour dock. Striking the half-hour As a small refinement it is possible to make the clock strike once on the half- hour. As the half-hour corresponds to an output of 3 or binary 0011 on the tens of minutes counter this can easily 3e derived by NANDing together the A 1252 — elektor december 1975 clamant clock (; and B outputs of the ten-minute coun- ter in the clock, as in figure 19. The out- put of this NAND-gate can then be used to trigger the bell, which will then strike every half-hour. A suitable drive circuit for the bell is given in figure 20. It consists of a mono- stable multivibrator driving a transistor which switches a relay. This enables the circuit to be wired into the household A.C. doorbell circuit. If a separate (D.C.) bell or chime is used it is possible to drive it directly with a transistor and dispense with the relay, as in figure 21. PI adjusts the pulse length of the mono- stable and hence the time for which the bell coil is energised. It should be ad- justed so that the bell will just strike reliably, to minimise the energised time and hence the dissipation in the coil. If a normal ding-dong type of door chime is used it may be a good idea to remove one of the tubular or bar res- onators so that the chime produces only a single stroke. In the larger tubular type of chime the tubes are usually suspended on cord or wire and are easily removed. The smaller types of chime usually employ metal bar res- onators which are suspended from rubber mounts. These can also be removed quite easily. The circuit of a complete striking system is given in figure 22. It embodies the ideas of figure 18 together with the half-hour striking circuit of figure 19. The only difference is that the spare half of IC6 is utilised and the strikin; occurs at a Vi Hz rate. If this is though to be too leisurely then the second input can be connected direct to pin *• of IC1. A printed circuit board am component layout for this circuit an given in figure 23. & Figure 23. Printed circuit board and compo- nent layout for the striking system of figure 22. sQ - 3 DI X S© © * — GZ1 — • M •il I e? .nnnnnnn f C3 O oaotjuua i annnnnn CO u UUUTJUUXJ l nnnnnnn oaaaoaa i nnnnnnn n U TJOOUUUTJ * i t nnnnnnn CM u uuuouoc l oonnmn u DUUUUUU jti > I I nnnnnnn CD O OUUUUUTJ nnnnnnn O auuoauu rs kO 6 cq 6 <6 J 6 too u 6 a> cc 6 l-ics elektor december 1975 — 1253 *CC <-0 <•* *v 36 3A 3* A 6Y Sa Sy 4a 4Y ■vj 43k 00 CD 46 4 A 4Y 36 3A 3Y >xj 43 k Inputs V Output Inputs 43k CD O Input A NC *xj CD Q 43 k 2 4^ 80 R 0(1) Rq(2) NC V CC R 9(|) R 9( j, Input >1 2 k D O + 3 k o X) 30 3A 26 GND "xl 4^ CD ro Inpul NC A 6 OC C 0 SN?4 144 N £k 43 k 7416*) *-xJ 43 k CD CD InJ^T T "' <>**• inputs *xj 4^ 2 + Input Rq(|) Rq(2) vj 4^ O vj ■xj 4^ ■xj CD + *D 4* ID Clock Clear K V£C Clock Cleor J S* >4 tic N NC At A2 6 0 GND A O 00 *xj 4^ *xj 43 k + 2 2 2 vrr Clcof 20 Clock Preset 20 ?0 IV 2A 20 ■D 43 k N3 ro Cleor ID Clock Preset 10 1 1 1 A A 03 43> Outputs T 5 r*-a f — sr E Clear Clod* °H °G o F q e Cleor a 8 bit shift Clock 6 0 A q b Qc o D I x 2 ]* u A2 61 62 ,Cltor v~ Dolo Inputs Senol Inputs Outputs «xj 4^ O + Cj 4^ ro * V CC 1C tv 3C 38 3A IV iReit/ 1 t A 10 2A 20 2C 2V GND 'xj 43k CD CD + -x / ■ I, load Dolo Doto C - 2 R«it/ C ejt C*> t A Clcor Borrow Corry Lood C B Count Count 0 °A Oowrn UP 0(- ±SL \2_ 11 1 4 . n s n 6 r 7 r °B \ 0 A Count Count ()£ °0 * uP J Outputs Inputs Outputs 20 2 C NC 28 10 l6 GND ?K 20 20 2 j rajTTUmJi ■xj 43 k — A CD ro 4 A 4V 38 3A 3v 'xj 43k CD CD Doto Inputs A V CC Cleor Qo 14 Cleor 0q D 0 Coynt/ Lood Clock Clock Qc C A 2 Clock Preset Cleor 1 J 1 1 1 V CC Clod Preset Ckor 2 2 2 tA tB IV 2A 26 2Y GND Count/ 0c Lood Dolo Inputs * open collector + Pin-compatible CMOS equivalents available from Teledyne Semiconductor and National Semiconductor 1254 — elektor december 1975 mos-i VqqHGMLFE O ro 4 * o CO 4 * o 4 * o HimimiHiHzP A B J K C D Vss Vdd H G M L F E A B J K C D V 5S vqd k h g f e nc r - 4 ^ 4>lReRR'°l--R--R— ' 4^ o J A B C D NC V ss Lli V DD Q2 02^ d a. o ui r- RR»M»M^R4>1-R Q2 Q2 CL2 R2 D2 52 DUAL DATA FLIP/FLOP % CLI Rl 01 SI l^trmznzmte ^ § s 1 s Vss OUTPUT J a K U < UJ _ _ u h to m m tn < o — » w_ _ < uj Jr ™ ^ JJ, V DD oco crco O O O a u< ^^3=0=1”^^ < Vss (£> < ^ n oj 2 uj 2 O^OO O o" S< < < -i UJ Q vss OUTPUTS (■ UJ to .UJ fld I/P _ UJ< I/P uJ -« r r; *.* FT err 03 v dd a d 05 *5* Q4 gZ Cl uj 4 R Cl 05 "5" 02 P. 5 STAGE COUNTER D r - ”2” 02 Oi 03 E. ••4“ 3" ill lli lit - llJ - (±| '.f J l ll 1 8 i ‘ 5 up up °* 01 J P vss o O ,0 V DD G H I L J C u i — R4^14^14°U»U^- 4^ O NJ 'vl 4* O ro o CO CJI A B D E F K Vss 4» O ro 4> o V DD NC Ql Q2 NC Q3 NC — r*i — f^i — i^i — i^i — i^°i — fi — f*i — i — LMil'lil - liriiJ~^l “lil H F- 07 Q6 Q5 Q4 Vss O UJ O to _j uj u a: OUTPUT I O uj INPUT a 0 lh A VDD 01 5l U d Kl Jl SET 01 on c LI Rl Kl Jl SI DUAL J/K FLIP/FLOP 02 03 CL2 R2 K2 J2 S2 im-iininininiru 02 02 , ^ h K 2 J2, SET Vss u uj v ' lO UJ a; OUTPUT 2 9 uj INPUT —* nr u BCD INPUTS oo vdd 03 oi / b c d /T qb Q3 01 B C D A 04 OBH 02 QO Q7 09 05 06 _ Q4 02 QO 07 09 Q5 Q6 VSS — — v DECIMAL OUTPUTS j — OUTPUT — | O IO IO PARALLEL inputs rv M ^ / / ’ v ■. V DD 0 O O Pi -4 Pl-3 Pi-2 PM r~R R R RR3~R R FI— i 02/(52 3 3 Pl-4 Pl-3 PI -2 — Ici/S’ g | Pll|— I T/C V J R CL P/S "tmininininnr t- ^ ►- * ujTTLj vqc wa SERIAL in tfi Q og S§ S d |«g <8 VSS U /C OUTPUTS /s CLOCK X IN V DD°II Q IO °8 °9 R 0 Q l 43=0=0=1”^^ q I 2 °6 °5 q 7 q 4 q 3 Q 2 v SS N / SA OUTPUTS 4* O 05 V DD uj S*§0 a EXTERNAL wQ. Q: RESISTORS aO na=B=B=B=B=B R2 Rl &0>Z PLL R2 Rl SF V; VCO vo i cm c(2> nx.x .i 8 a E mi Vss a?°ao8 o ,9 1 CAPACITOR a U u > ~ 4^ o 45^ CO (1C L F NC K E J D ~ ck 0 h rCE OIL 4^ O CD 00 TD J H G F E N/C MJRiHzlRHiMzrtLM N/C A B C D N/C VSS 4^ O o V DD r -4RR^4RR-R-R— , Si" K v -R lil iil-liHil-lzHil- Vss 45» O v DO J £ l£l_ vss o CO vdd r — R-R-Rh-R-R-R-R Vss 4^ O v dd oo r -R-4R-R-R-R-H-R— Vss 4* O 00 ro Vdd nc r — R-R-R-R-R-R-R — i tRiFlMHilizIiir NC V SS o v od h g m l f E S r-RRRRRRR-RR 7 !- IJiiHiJizHiHzJiiR A B J K C D VSS O o EXTERNAL TRIGGER CO MPONENTS f~. INPUTS OUTPUTS 4^ fei rv rv N ui / 2 Sc^ yv O V^rv x x x uj h h >n V DD u U a. OUTPUTS a. Note: A prefix to the type number denotes the manufacturer, e.g. CD 4001 ( RCA), MC 14001 (Motorola), N 4001 (Signetics), SCL 4001 (Solid State Scientific), SIL 4001 (Siltek). tup-tun-dug-dus elektor december 1975 — 1255 TUP Wherever possible in Elektor circuits, transis- tors and diodes are simply marked 'TUP' (Transistor, Universal PNP),'TUN' (Transistor, Universal NPN), 'DUG' (Diode, Universal Ger- manium) or 'DUS' (Diode, Universal Silicon). This indicates that a large group of similar devices can be used, provided they meet the minimum specifications listed above. For further information, see the article 'TUP- TUN-DUG-DUS' in Elektor 1, p. 9. type u ce 0 max • c max hfe min. Ptot max fT min. TUN TUP NPN PNP 20 V 20 V 100 mA 100 mA 100 100 100 mW 100 mW 100 MHz 100 MHz Table la. Minimum specifications for TUP and TUN. Table 1b. Minimum specifications for DUS and DUG. type UR max IF max IR max Ptot max CD max DUS DUG Si Ge 25 V 20 V 100 mA 35 mA 1 (JA 100 jLiA 250 mW 250 mW 5 pF 10 pF L Table 2. Various transistor types that meet the TUN specifications. TUN BC 107 BC 208 BC 384 BC 108 BC 209 BC 407 BC 109 BC 237 BC 408 BC 147 BC 238 BC 409 BC 148 BC 239 BC 41 3 BC 149 BC 317 BC 414 BC 171 BC 318 BC 547 BC 172 BC 319 BC 548 BC 173 BC 347 BC 549 BC 182 BC 348 BC 582 BC 183 BC 349 BC 583 BC 184 BC 382 BC 584 BC 207 BC 383 Table 3. Various transistor types that meet the TUP specifications. TUP BC 157 BC 253 BC 352 BC 158 BC 261 BC 415 BC 177 BC 262 BC 416 BC 178 BC 263 BC 417 BC 204 BC 307 BC 418 BC 205 BC 308 BC 419 BC 206 BC 309 BC 512 BC 212 BC 320 BC 513 BC 213 BC 321 BC 514 BC 214 BC 322 BC 557 BC 251 BC 350 BC 558 BC 252 BC 351 BC 559 Table 4. Various diodes that meet the DUS or DUG specifications. DUS DUG BA 127 BA 318 OA 85 BA 217 BAX 1 3 OA 91 BA 218 BAY 61 OA 95 BA 221 1N914 AA 116 BA 222 1 N4148 BA 317 Table 5. Minimum specifications for the BC107, -108, -109 and BC177, -178, -179 families (according to the Pro-Electron standard). Note that the BC179 does not necessarily meet the TUP specification 0c, max ~ 50 mA). NPN PNP BC 107 BC 177 BC 108 BC 178 BC 109 BC 179 V ce 0 45 V 45 V max 20 V 25 V 20 V 20 V v eb 0 6 V 5 V max 5 V 5 V 5 V 5 V 'c 1 00 mA 100 mA max 100 mA 1 00 mA 1 00 mA 50 mA p tot. 300 mW 300 mW max , 300 mW 300 mW 300 mW 300 mW fT 150 MHz 130 MHz min. 150 MHz 130 MHz 150 MHz 130 MHz F 10 dB 1 0 dB max 10dB 10 dB 4 dB 4 dB The letters after the type number denote the current gain: A: a' (j3, h fe ) = 125-260 B: a' = 240-500 C: a' = 450-900. Table 6. Various equivalents for the BC107, -108, . . . families. The data are those given by the Pro-Electron standard; individual manu- facturers will sometimes give better specifi- cations for their own products. NPN PNP Case Remarks BC 107 BC 108 BC 109 BC 177 BC 178 BC 179 •Q E BC 147 BC 148 BC 149 BC 157 BC 158 BC 159 • r- O z 0 H 1 m □ O H H m O i“ m CANTERBURY CT1 2BR 3. FOLD HERE AND TUCK IN 1 I would like to receive a □ copy of ELEKTOR no □ subscription to ELEKTOR, starting with no. . □ trial subscription to ELEKTOR. □ dark green ELEKTOR binder. I am not enclosing money, but await your invoice Name Add ress Post code Cut out address and glue it on envelope; cut out order slip and enclose it in envelope. or: Remove page then fold and tuck in as indi- cated. No stamp is needed. ELEKTOR BACK ISSUES. 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